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NASA-CR-193013 






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NASA 




third International Symposium on 
Space terahertz technology 



JPL 




Th« University 
of Michigan 



Symposium Proceedings 




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March 24-26, 1992 

University of Michigan 
Ann Arbor, Michigan 



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Sponsored by: 
NASA Office of Aeronautics and Space Technology (OAST), University Space Engineering Research Centers 
Program, with cooperative sponsorship by the Microwave Theory and Techniques Society of IEEE. 

Organized Jointly by: 
The University of Michigan's NASA Center for Space Terahertz Technology and JPL's Center for Space 
Microelectronics Technology. 



Proceedings of the 

Third International Symposium on 

Space Terahertz Technology 



March 24-26, 1992 



University of Michigan 
Ann Arbor, Michigan 



Symposium Co-chairs: 
Technical Co-chairs: 



Organizing Committee 

Fawwaz T. Ulaby, University of Michigan 
Carl A. Kukkonen, Jet Propulsion Laboratory 

Gabriel M. Rebeiz, University of Michigan 
Margaret A. Frerking, Jet Propulsion Laboratory 



Local Arrangements: Valerie Kabat, University of Michigan 

Symposium Proceedings: Valerie Kabat, University of Michigan 



Group Photo 
A group photo of some of the 1992 Symposium participants. 




ORIGINAL PAGE 
BLACK AND WHITE PHOTOGRAPH 



Preface 

The Third International Symposium on Space Terahertz Technology was held at the 
University of Michigan in Ann Arbor, Michigan, on March 24-26, 1992. The Symposium, 
which was attended by approximately 165 scientists and engineers from the U.S., Europe, 
and Japan, featured papers relevant to the generation, detection, and use of the terahertz 
spectral region for space astronomy and remote sensing of the Earth's upper atmosphere. 
The program included thirteen sessions covering a wide variety of topics including solid- 
state oscillators, power-combining techniques, mixers, harmonic multipliers, antennas and 
antenna arrays, submillimeter receivers, and measurement techniques. 

The Symposium was sponsored by the University Space Engineering Research Centers 
Program of NASA's Office of Aeronautics and Space Technology (OAST), and organized 
jointly by the University of Michigan's NASA Center for Space Terahertz Technology and 
JPL's Center for Space Microelectronics Technology. The Microwave Theory and 
Techniques Society of IEEE served as a cooperative sponsor of the Symposium, as well as a 
medium for publication of some of the papers that were presented at the Symposium in 
the form of a mini-special issue (April 1993) of the IEEE-MIT Transactions. 

The Fourth International Symposium on Space Terahertz Technology will be held at the 
University of California, Los Angeles, on March 30-April 1, 1993. 



Fawwaz T. Ulaby 
Carl A. Kukkonen 



Contents 



Opening Session Chain Fawwaz Ulaby 

NASA's OAET Sensors Program * 

M. Sokoloski 

NASA's Astrophysics Program in High-Resoluton THz Spectroscopy * 

L. Caroff 

Applications of Terahertz Technology to Astronomy * 

T. G. Phillips 

Coherent Systems in the Terahertz Range: Elements, Operation & Examples 1 — / 

P. F. Goldsmith 



Session L Sources I Chain CarlKukkonen 

Broadband Millimeter-Wave GaAs Transmitters and Receivers Using 

Planar Bow-Tie Antennas 24~2^ 

Y. Konishi, M. Kamegawa, M. Case, R. Yu, M. J.W. Rodwell, 

R. A. York, D. B. Rutledge 

Active CPW-Fed Slot Antennas for Power Combining Applications 32— >2> 

B. K. Kormanyos, G. M. Rebeiz 

2x2 Quasi-Optical Power Combiner Array at 20 GHz 37-^/ 

S. Kawasaki, T. Itoh 

Monolithic Millimeter-Wave Diode Array Beam Controllers: 

Theory and Experiment 45 ~~> 

L. B. Sjogren; H-X. L. Liu; F. Wang; T. Liu; W. Wu; X-H. Qin; E. Chung; 

C. W. Domier; N. C. Luhmann, Jr.; J. Maserjian; M. Kim; J. Hacker; 

D. B. Rutledge; L. Florez; J. Harbison 

A Study of Subterahertz HEMT Monolithic Oscillators 58""'^ 

Y. Kwon and D. Pavlidis 



Session 2: Varactors I Chair: Tatsuo Itoh 

Varactor Diodes for Millimeter and Submillimeter Wavelengths 73 ""V 

B. J. Rizzi, J. L. Hesler, H. Dossal, T. W. Crowe 

A Schottky/2-DEG Varactor Diode for Millimeter and _ 

Submillimeter Wave Multiplier Applications 93 ^ O 

W. C. B. Peatman, T. W. Crowe, M. Shur, B. Gelmont 

Thermionic Emission Current in a Single Barrier Varactor 110 ~" / 

H. Hjelmgren, J. R. East, E. Kollberg 



*Presentation only. 



Progress on Single Barrier Varactors for Submillimeter Wave 

Power Generation 115 "10 

S. M. Nilsen, H. Gronqvist, H. Hjelmgren, A. Rydberg, E. Kollberg 

Effect of Cooling on the Efficiency of Schottky Varactor 

Frequency Multipliers at Millimeter Waves 134-7/ 

J. Louhi, A. Raisanen, N. Erickson 

Session 3: Varactors n Chair RayBlundell 

Superlattice Barrier Varactors 146 "/ *■ — 

C. Raman, J. P. Sun, W. L. Chen, G. Munns, J. East, G. Haddad 

A New Fabrication Technique for Back-to-Back Varactor Diodes 158 W "^ 

R. P. Smith, D. Choudhury, S. Martin, M. A. Frerking, J. K. Liu, 
F. A. Grunthaner 

A 200 GHz Tripler Using Single Barrier Varactor 164 -7 7 

D. Choudhury, M. A. Frerking, P. D. Batelaan 

A Submillimeter Tripler Using a Quasi-Waveguide Structure 181 " * 

N. Erickson, G. Cortes-Medellin 



Session 4: SIS Receivers I Chain Neville Luhmann 

A 380 GHz SIS Receiver Using Nb/A10 x /Nb Junctions for a 

RadioAstronomical Balloon-Borne Experiment: PRONAOS 189 -/& 

P. Febvre, P. Feautrier, C. Robert, J. C. Pernot, A. Germont, M. Hanus, 

R. Maoli, M. Gheudin, G. Beaudin, P. Encrenaz 

A Low Noise 410-495 Heterodyne Two Tuner Mixer, Using Submicron 

Nb/Al203/Nb Tunneljunctions 210 "pi 

G. de Lange, C. E. Honingh, M. M. T. M. Dierichs, R. A. Panhuyzen, 

H. H. A. Schaeffer, T. M. Klapwijk, H. van de Stadt, M. W. M. de Graauw 

Double Dipole Antenna SIS Receivers at 100 and 400 GHz 222 — / % 

A. Skalare, H. van de Stadt, Th. de Graauw, R. A. Panhuyzen, 
M. M. T. M. Dierichs 

Slot Antenna SIS Mixers for Submillimeter Wavelengths** 234 ~ * I 

J. Zmuidzinas, H. G. LeDuc, J. A. Stern 

Session 5: SIS Receivers II Chain David Rutledge 

A Planar Quasi-Optical SIS Receiver for Array Applications 235^ 

P. A. Stimson, R. J. Dengler, P. H. Siegel, H. G. LeDuc 

A Fixed Tuned Broadband Matching Structure for Submillimeter . 

SIS Receivers** 243 ~-0.\ 

T. H. Buttgenbach, H. G. LeDuc, P. D. Maker, T. G. Phillips 



* Abstract only. 



Modeling and Performance of Nb SIS Mixers in the 1.3 mm 

and 0.8 mm Bands 244 ~v?2_^ 

A. Karpov, M. Carter, B. Lazareff, D. Billon-Pierron, K.H. Gundlach 

Comparison of Measured and Predicted Performance 

of a SIS Waveguide Mixer at 345 GHz 251-^3 

C. E. Honingh, G. deLange, M. M. T. M. Dierichs, H. H. A. Schaeffer, 

J. Wezelman, J. v. d. Kurr, Th. de Graauw, T. M. Klapwijk 

A Low Noise 492 GHz SIS Waveguide Receiver 266 ^~^¥ 

C. K. Walker, J. W. Kooi, M. Chan, H. G. LeDuc, P. L. Schaffer, / 

J. E. Carlstrom, T. G. Phillips 

Session 6: Antennas & Networks I Chair: Gabriel Rebeiz 

Slot-Line End-Fire Antennas for THz Frequencies 280~^-D 

H. Ekstrom, S. Gearhart, P. R. Acharya, H. Dave, G. Rebeiz, S. Jacobsson, 
E. Kollberg, G. Chin 

Quasi-Optical Antenna-Mixer-Array Design for Terahertz Frequencies 291 "W & 

Y. Guo, K. A. Potter, D. B. Rutledge 

Analysis of a Novel Non-Contacting Waveguide Backshort 298 ~~i? / 

T. M. Weller, L. P. B. Katehi, W. R. McGrath 

Silicon Micromachined Waveguides for Millimeter 

and Submillimeter Wavelengths 316 ~b2^P 

M. Yap, Y.C. Tai, W.R. McGrath, C. Walker 

Session 7: Antennas & Networks II Chair: Linda Katehi 

Progress in Integrated-Circuit Horn Antennas for Receiver Applications: 

Parts I & II 324 -^ ^3^ 

G. V. Eleftheriades, W. Y. Ali-Ahmad, G. M. Rebeiz 



Zone Plate Lens Antennas for Millimeter and Submillimeter Wavelengths 345 

P. F. Goldsmith 



-31 



Onset of Dispersion in Nb Microstrip Transmission Lines 

at Submillimeter Wave Frequencies 362 ^j5 2-» 

H. H. S. Javadi, W. R. McGrath, B. Bumble, H. G. LeDuc 

Double-Slot Antennas on Extended Hemispherical Dielectric Lenses 382 "~ * ^ 

D. F. Filipovic, S. J. Gearhart, B. K. Kormanyos, G. M. Rebeiz 

Session 8: SIS Theory & Fabrication I Chain MarcFeldman 

Embedding Impedance Approximations in the Analysis of SIS Mixers 394 ""^2 j/ 

A. R. Kerr, S. K. Pan, S. Withington / 



Submicron Area Nb/ AlOx/Nb Tunnel Junctions / ~ 

for Submillimeter Mixer Applications 408 ^ ' pb 

H. G. LeDuc, B. Bumble, S. R. Cypher, A. J. Judas, J. A. Stern 

Noise in Josephson Effect Mixers and the RSJ Model** .419 - 3"? 

R. Schoelkopf, T. Phillips, J. Zmuidzinas 

Fabrication and Characterization of High Current-Density, ^ 

Submicron NbN/MgO/NbN Tunnel Junctions 420 "3 L 

J. A. Stern, H. G. LeDuc, A. J. Judas 

Session 9: Sources II Chain James Mink 

A Quasioptical Resonant-Tunneling-Diode /> 

Oscillator Operating Above 200 GHz** 439 ^3^ 

E. R. Brown, C D. Parker, K. M. Molvar, A. R. Calawa, M. J. Manfra 

Transit-Time Devices as Local Oscillators for Frequencies - 

above 100 GHz 440-3/ 

H. Eisele, C. Kidner, G. I. Haddad 

Negative Differential Resistance (NDR) Frequency , n 

Conversion with Gain 457^ H 

R. J. Hwu, R. W. Aim, S. C. Lee 

Modeling, Design, Fabrication and Testing of 

InP Gunn Devices in the D-Band (110 GHz-170 GHz) 477 ^i\ / 

R. Kamoua, H. Eisele, J. R. East, G. I. Haddad, G. Munns, M. Sherwin 

Session 10: SIS Theory & Fabrication II Chair: Anthony Kerr 

Recent Advances in Superconducting-Mixer Simulations 494 **h ^ 

S. Withington, P. R. Kennedy 

Submillimeter Wave Detection with Superconducting . . . « 

Tunnel Diodes 502 ~*j£ 

M. J. Wengler 

Evaluation of Integrated Tuning Elements with SIS Devices 522 

M. M. T. M. Dierichs, C. E. Honingh, R. A. Panhuyzen, B. J. Feenstra, 
A. Skalare, J. J. Wijnbergen, H. v. d. Stadt, Th. de Graauw 

Source Conductance Scaling for High Frequency ^\Jh 

Superconducting Quasiparticle Receivers 538 "-' 

Q. Ke, M. J. Feldman 



** Abstract only. 



Session 11: Sources m Chair: George Haddad 

Resonant Tunneling Diodes as Sources for 

Millimeter and Submillimeter Wavelengths 548— y^ 

O. Vanbesien, R. Bouregba, P. Mounaix, D. Lippens, L. Palmateer, J. C. Pernot, 
G. Beaudin, P. Encrenaz, E. Bockenhoff, J. Nagle, P. Bois, F. Chevoir, B. Vinter 

Simulation of Electron Transport in Quantum Well Devices 560 — V ' 

D. R. Miller, K. K. Gullapalli, V. R. Reddy, D. P. Neikirk 

Parallel Arrays of Josephson Junctions for Submillimeter Local Oscillators 575 ~ V O 

A. Pance, M. J. Wengler 

Monolithic Millimeter- Wave Diode Grid Frequency Multipler Arrays 595 "77 

H. X. L. Liu; X. H. Qin; L. B. Sjogren; W. Wu, E. Chung; 
C. W. Domier; N. C. Luhmann, Jr. 

Session 12: Mixers and Detectors I Chairs: Margaret Frerking 

Planar GaAs Diodes for THz Frequency Mixing Applications 600 "**i?Z5 

W. L. Bishop, T. W. Crowe, R. J. Mattauch, H. Dossal 

Planar Doped Barrier Subharmonic Mixers 616 "^ ' 

T. H. Lee, J. R. East, G. I. Haddad. 

New Approach to the Design of Schottky Barrier Diodes for THz Mixers 631 , ~*<3 ^*» 

A. Jelenski, A. Grub, V. Krozer, H. L. Hartnagel 

Electrical and Infrared Properties of Thin Niobium Microbolometers Near T c .643 ~>S3 

E. N. Grossman, J. E. Sauvageau, D. G. McDonald 

Session 13: Mixers & Detectors II Chair: Thomas Crowe 

Measurements of the Single Sideband Suppression for a 

650 GHz Heterodyne Receiver 654 — <£5^/ 

S. Crewell, H. Nett / 

InGaAs/InP Heteroepitaxial Schottky Barrier Diodes for Terahertz _ 

Applications 661 "--0-3 

U. V. Bhapkar, Y. Li, R. J. Mattauch 

A Broadband THz Receiver for Low Background Space Applications 678 -~ Q1& 

C. Hagmann, D. J. Benford, A. C. Clapp, P. L. Richards, P. Timbie 

AlGaAs/GaAs Quasi-Bulk Effect Mixers: Analysis and Experiments 688 ^£? / 

K. S. Yngvesson, J.-X. Yang, F. Agahi, D. Dai, C. Musante, W. Grammer, K. M. Lau 

All-Solid-State Radiometers for Environmental Studies to 700 GHz 706 —S3 

R. Riidiger, P. Zimmermann 

A 492 GHz Cooled Schottky Receiver for Radio Astronomy 724 

J. Hernichel, R. Schieder, J. Stutzki, B. Vowinkel, G. Winnewisser, 
P. Zimmermann 



Third International Symposium on Space Terahertz Technology Page 1 

N93-27727 

COHERENT SYSTEMS IN THE TERAHERTZ 

FREQUENCY RANGE : ^ oSJ £ 

ELEMENTS C. - 

OPERATION 
& EXAMPLES 



PAUL F. GOLDSMITH 

MILLITECH CORPORATION 
South Deerfield MA, 01373 

and 

FIVE COLLEGE RADIO ASTRONOMY OBSERVATORY 

Department of Physics and Astronomy 

University of Massachusetts, Amherst MA 01003 



8 e 2 Third International Symposium on Space Terahertz Technology 

TERAHERTZ COHERENT SYSTEMS APPLICATIONS 

RADIOMETRY / SPECTROSCOPY 

ASTRONOMY 

ATMOSPHERIC REMOTE SENSING 
ALL-WEATHER SYNTHETIC VISION SYSTEMS 
CONTRABAND DETECTION 

HIGH POWER 

PLASMA HEATING 

HIGH ENERGY ACCELERATORS 

PLASMA DIAGNOSTICS 

THERMAL IMAGING 
DENSITY PROBING 
BACKSCATTER MEASUREMENTS 

COMMUNICATIONS 

PERSONAL & VEHICULAR 

DIGITAL DATA LINKS 

TV REMOTE / STUDIO LINKS 

MATERIALS MEASUREMENT AND COMMERCIAL PROCESS CONTROL 

PAPER MAKING 

HV CABLE MANUFACTURING 

RADAR SYSTEMS 

MILITARY - SEEKERS, INSTRUMENTATION, AND MODELING 

AUTOMOTIVE COLLISION AVOIDANCE 

ATMOSPHERE, METEOROLOGY, GROUND, ICE, AND FOLIAGE 



Third International Symposium on Space Terahertz Technology 



Page 3 



COMPONENTS OF COHERENT SYSTEMS 
AT MILLIMETER & SUBMILUMETER WAVELENGTHS 



/ 
/ / 



INPUT OPTICS 



SIGNAL 

PROCESSING 

ELEMENTS 



COLUMAT1NG MIRRORS 
AND LENSES 



POLARIZING GRIDS 
WAVEPLATES 



@— XN 




COHERENT SOURCE 



LOCAL OSCILLATOR; 
TRANSMITTER 



< 



DIPLEXER 



%-rv— 1 



COMBINATION OF 
LOCAL OSCILLATOR 
AND SIGNAL 



1L 




ANTENNA/FEED 
ELEMENTS 



MIXER 



IF SYSTEM 



EFFICIENCY, BEAMWIDTH, 
BANDWIDTH, CONSTRUCTION 
INTEGRABIUTY 



CONVERSION LOSS; NOISE; 
LO. POWER; BANDWIDTH 



NOISE; BANDWIDTH 



•S-KHD 



^o 



S-KHD 



DETECTION/SIGNAL 
PROCESSING 



SPECTROMETERS: 
FREQUENCY COVERAGE; 
RESOLUTION; FLEXIBILITY; 
POWER CONSUMPTION 



Page 4 Third International Symposium on Space Terahertz Technology 

BRIEF OVERVIEW OF SELECTED COMPONENTS 



EMPHASIZE AREAS THAT I FEEL DESERVE MORE ATTENTION 
THAN THEY ARE RECEIVING AT PRESENT 



[1] MATERIALS MEASUREMENT 

FUNDAMENTAL FOR MANY ASPECTS OF SYSTEMS DESIGN 
NEED MORE DATA, BETTER DATA, AND BETTER ACCESS 



REXOLITE DATA FROM G. J. SIMONIS, J. P. SATTLER, T. L. 
WORCHESKY, AND R. P. LEAVITT, INT. J. INFRARED 
AND MILLIMETER WAVES, VOL. 5, 57 - 72, 1984. 



BORON DATA FROM. A. J. GATESMAN, R. H. GILES, AND J. 
NITRIDE WALDMAN PROC. MATERIALS RESEARCH SOCIETY 

SYMPOSIUM ON WIDE BANDGAP SEMICONDUCTORS, 1991 

FALL MEETING, BOSTON 



INTERCOMPARISON OF TECHNIQUES FOR DETERMINATION OF 
NEAR MILLIMETER DLELECTRIC PROPERTIES 

JAMES BIRCH ET AL. - NATIONAL PHYSICAL LABORATORY 

TEDDINGTON, MIDDLESEX 
U.K. TW110LW 

REPORT DES 115, OCTOBER 1991 



Third International Symposium on Space Terahertz Technology p a g e 5 

[2] QUASIOPTICAL COMPONENTS 

HOW CAN THEY BE FABRICATED IN SUBMILLIMETER REGION? 



TRADITIONAL MACHINING METHODS BECOME VERY DIFFICULT 
AND EXPENSIVE- 



NEED TO FIND CONSTRUCTIVE COMBINATIONS OF METAL- 
WORKING AND SEMICONDUCTOR PROCESSING APPROACHES SUCH 
AS SELECTIVE ETCHING 



EXAMPLES: 



PROCESSING SILICON TO FABRICATE TWO DIMENSIONAL 
IMAGING HORN ANTENNA ARRAYS (REBEIZ ET AL. IEEE 
MTT 38, 1473 (1990)) 



ETCHING AND PLATING SILICON TO MAKE DICHROIC PLATE 

HIGH PASS FILTERS IN 1000 GHZ RANGE 

(SIEGEL AND LICHTENBERGER 1990 MTT-S SYMP. DIGEST, 

1341) 



Pige 6 Third International Symposium on Space Terahertz Technology 

RADIOMETRY AND SPECTROSCOPY : ASTRONOMY 

OBSERVING LOCATION DEPENDS PRIMARILY ON FREQUENCY: 

GROUND - BASED 

AIRPLANE AND BALLOONS: KAO; SOFIA 

SPACE: SWAS; SMIM; FIRST 

[1] SENSITIVITY 

HIGHEST SENSITIVITY ALWAYS REQUIRED 

CRYOGENIC COOLING IS ACCEPTABLE 

BROADBAND SYSTEMS WILL BE REQUIRED FOR FUTURE SYSTEMS 

[2] IMAGING SYSTEMS 

FOCAL PLANE ARRAYS DEVELOPED FOR MILLIMETER RANGE: 

FCRAO 15 -ELEMENT QUARRY ARRAY 85-115 GHZ 
NRAO 8-ELEMENT ARRAY IN 230 GHZ RANGE 

CANNOT SACRIFICE FEED EFFICIENCY SIGNIFICANTLY JUST TO 
OBTAIN LARGER NUMBER OF ELEMENTS DUE TO COST AND 
COMPLEXITY OF ASSOCIATED SIGNAL PROCESSING. 



[3] OTHER COMPONENT DEVELOPMENT 

RAPID PROGRESS IN FREQUENCY MULTIPLIER SOURCES, BUT 
FURTHER DEVELOPMENT REQUIRED FOR GREATER BANDWIDTH 
AND REACHING HIGHER FREQUENCIES 



Third International Symposium on Space Terahertz Technology 



Page 7 



PLANAR HETERODYNE ARRAY USING A DIELECTRIC-FILLED PARABOLA 



Receive 
Element 
(Schottky 
Dlod« or SIS 
Tunnel Junction) 



Focal Point of 
Dielectric— Riled 
Parabola 



Coplanar 
Lines for 
IF it DC 



Signal 
Input 




EXPLODED VIEW (SIDE) 



1.0 INCH 
I I | I I I I | 



TOP VIEW 
(COVER REMOVED) 



P.H. Siegel 



California Institute of Technology Jet Propulsion Laboratory 



p <*ge 8 Third International Symposium on Space Terahertz Technology 

RADIOMETRY: 
AIRCRAFT ALL WEATHER LANDING SYSTEM 

APPROACH 

FOCAL PLANE IMAGING SYSTEM AT 94 GHZ TO PROVIDE 
SYNTHETIC VISION CAPABILITY FOR AIRCRAFT LANDING IN 
ALMOST ALL WEATHER CONDITIONS 

MILLIMETER - WAVE IMAGING ALLOWS GOOD VISIBILITY OF 
RUNWAY BOUNDARIES AND POSSIBLY DANGEROUS OBSTACLES 
FROM APPROPRIATE DISTANCE 

FOCAL PLANE RADIOMETRIC IMAGING PERMITS REAL-TIME (30/ 
SECOND) UPDATE RATE 

IMAGES READILY INTERPRETABLE WITHOUT EXTENSIVE 
PROCESSING 

HEADS-UP DISPLAY STRAIGHTFORWARD TO IMPLEMENT 

TECHNOLOGY: 

FOCAL PLANE ARRAY OF 256 (TO DATE) PIXELS UTILIZING 
CONSTANT - WIDTH SLOT ANTENNAS 

SINGLE -ENDED HARMONIC MIXERS WITH QUASIOPTICAL LOCAL 
OSCILLATOR INJECTION 

DICKE-TYPE LOAD COMPARISON ESSENTIAL 

MECHANICAL OR ELECTRONIC (QUASIOPTICAL HYBRID OR 

MONOLITHIC ) REALIZATIONS POSSIBLE 

COMPACT OPTICS 



Third International Symposium on Space Terahertz Technology 



Page 9 




Page 10 Third International Symposium on Space Terahertz Technology 



DETECTION OF CONCEALED 
WEAPONS AND CONTRABAND MATERIAL 

E RQ BI ^M : 

• DETECTION OF PLASTIC WEAPONS AND 
EXPLOSIVES CONCEALED BENEATH CLOTHING OF 
AIRLINE PASSENGERS. 

CONSTRAINTS : 

• EFFECTIVE PERFORMANCE 

• NON-INVASIVE OPERATION 

• RAPID PROCESSING 

TECHNICAL APPROACH : 

• ACTIVE (REFLECTING) AND PASSIVE (RADIOMETRIC) 
MILLIMETER-WAVELENGTH IMAGING SYSTEMS 

• RADIOMETRIC SYSTEM LEAST INVASIVE AND 
OFFERS GOOD FIDELITY 

• CLOSE FOCUSED OPTICS AND FOCAL PLANE ARRAY 




millitech 



Third International Symposium on Space Terahertz Technology 



Page 11 




Scan 86: 10/81x91 14:39 
rr-rt vla» 
la talst 



x (crt) : -38.88 38.88 8.25 t 241 1 
*, (err) : -28.88 VS. 88 8.25 < 381 i 



Passive Line Scan 94 GHz Millimeter Wave Image 



ORIGINAL PAQE IS 
OF POOR QUALITY 



/\ millitacfr 



Page 12 



Third International Symposium on Space Terahertz Technology 



RADIOMETRY: ATMOSPHERIC REMOTE SENSING 



[1] ISSUES: 

MEASUREMENT OF TRACE CONSTITUENTS INCLUDING: H 2 

3 
CIO 
N 2 

PHYSICAL CONDITION (TEMPERATURE) PROFILING 

DELAY MEASUREMENTS FOR RADAR ALTIMETERS 

MESOSPHERIC WIND VELOCITY DETERMINATIONS 

TRACE EMISSIONS FROM LOCALIZED SOURCES 



[2] OBSERVING LOCATIONS 



GROUND - BASED: 3 AND CIO MONITORING NETWORK 



ANTARCTIC AND POLAR REGIONS 



AIRPLANE: 



USEFUL AS TEST PLATFORM AND FOR 
STUDY OF LOCALIZED PHENOMENA 



SPACE 



UARS - SUCCESSFULLY OPERATING ! 



MAS (SHUTTLE LIMB - SOUNDER) 



EOS (EARTH OBSERVING SYSTEM) 



AMSU-B / METEOSAT 



Third International Symposium on Space Terahertz Technology 



Page 13 



. LOW ELEVATION 
• BEAM 

A. \ ~ A 



■JJ ' \ 



HIGH ELEVATION 
BEAM 

A . *A * 






BEAM SWITCHING 
CHOPPER 




PARTIAL ABSORBER FOR TOTAL POWEH BALANCE 



SPECTROMETER 



CONTROL 
AND 

ANALYSIS 
COMPUTER. 



CONFIGURATION FOR GROUND-BASED RADIOMETER TO STUDY ATMOSPHERIC TRACE GASES 



094 ca COLLIMATING 
LENS 



SIGNAL 



DOUBLE DIELECTRIC 
SLAB FILTER 



TRANSMISSION 




MILLIMETER 

FEEOHORN 




LOCAL 
OSCILLATOR 



SIGNAL 
BANO 



IMAGE 
BANO 



FREQUENCY -— 



ABSORBING 
LOAO 



DIELECTRIC SLAB SINGLE-SIDEBAND FILTER FOR 279 GHZ CIO RADIOMETER 



Page 14 



Third International Symposium on Space Terahertz Technology 




Third International Symposium on Space Terahertz Technology 



Page 15 



r SCANNING 
ANTENNA 
SYSTEM 

THERMAL RADIATION 



PRIMARY 



FROM ATMOSPHERIC LIMB / 




SPACE VIEW 



FOR CALIBRATION 



SWITCHING 
MIRROR 



OICHROIC 
PLATE 



H 



63 GHz 
RAOIOMETER 



FILTER BANK 



POLARIZATION 
GRIO 



CAL 
TARGET 



205 GHz 
RAOIOMETER 



183 GHz 
RAOIOMETER 



CIO 
FILTER BANK 



H 2 Oj 
FILTER BANK 



FILTER BANK 



H 2 
FILTER BANK 



I FILTER BANK 



H 



COMMAND 

AND 

DATA 
HANOLING 



TO 
UARS 



UARS MICROWAVE LIMB SOUNDER INSTRUMENT SIGNAL FLOW PATH 



h/lcm 
80 



70 



O s 



H,0 



60 



50 



40 



30 



20 



10 - 



,L 



CIO 



UARS- MLS TARGETS AND ALTITUDE RANGES 



Page 16 Third International Symposium on Space Terahertz Technology 

PLASMA DIAGNOSTICS 

THERMAL IMAGING - RADIO METRY WITH HIGH TIME RESOLUTION 
EXTREMELY BROADBAND AND/OR SWEPT - FREQUENCY 



DENSITY PROFILING - MEASUREMENT OF ELECTRON COLUMN 

DENSITY THROUGH PLASMA 

INTERFEROMETERS - EITHER RADIO OR OPTICAL TYPES 
DEPENDING ON WAVELENGTH 



SCATTERING EXPERIMENTS- PROBE TURBULENCE AND SCALE 

OF FLUCTUATIONS IN PLASMA 



EXAMPLE OF PLASMA DIAGNOSTIC SYSTEM 

2 - MM WAVELENGTH 180 DEGREE BACKSCATTER IMAGING 
SYSTEM DEVELOPED BY DR. P. EFTHIMION (PRINCETON 
PLASMA LABORATORY) AND E.L. MOORE ET. AL. (MILLITECH 
CORPORATION) 

INCLUDES PHASELOCKED TRANSMITTER AND 64 ELEMENT 
FOCAL PLANE IMAGING ARRAY 



Third International Symposium on Space Terahertz Technology Page 17 



COMMUNICATIONS 



APPLICATIONS: 



PERSONAL 



VEHICULAR - CAR TRAIN AND PLANE 



DIGITAL DATA LINKS - SATELLITE AND GROUND 



MILITARY COMMUNICATIONS (MILSTAR) 



TV REMOTE - STUDIO LINKS 



DEVELOPMENTS IN FIELD HAVE BEEN REVIEWED BY H. MEINEL IN 
PROC. 18*k EUROPEAN MICROWAVE CONFERENCE, STOCKHOLM, 
pp. 1203-1216, 1988 



Page 18 Third International Symposium on Space Terahertz Technology 

RADAR SYSTEMS 

■ MILITARY RADAR SYSTEMS 

INSTRUMENTATION RADARS 

SEARCH RADARS 

SEEKERS 

HELICOPTER OBSTACLE AVOIDANCE SYSTEMS 

■ AUTOMOTIVE RADAR 

PRESENTLY VERY ACTIVE FIELD 

GOALS ARE COLLISION AVOIDANCE AND ULTIMATELY 
AUTOMATIC CONTROL OF VEHICLE 



ATMOSPHERE CLOUD STRUCTURE (ICE & WATER) 

METEOROLOGY 



REMOTE SENSING OCEANS 

VEGETATION 
ICE 



MODELING 

MILLIMETER / SUBMILLIMETER MODELING OF LOWER 
FREQUENCY RADAR SYSTEMS AND TARGETS 



Third International Symposium on Space Terahertz Technology 



Page 19 



GUNN OSCILLATOR 



CZD- 



POLARIZATION 2 
IF OUTPUTS 



POWER 
DIVIDER 



ODD 



POLARIZATION- 
DIPLEXING GRID 



POLARIZATION 1 
IF OUTPUTS 



LENS- 




T 



LINEAR/CIRCULAR 
TRANSFORMER 



DUAL POLARIZATION MONOPULSE LENS ANTENNA 



GS07 



SPECIFICATIONS SUMMARY 

TYPE: PULSE 

TRANSMITTER FREQUENCY: 77GHz 

PULSE WIDTH: 40ns 

RISE/FALL TIME: 4ns 

ANTENNA: THREE BEAM SCANNING 

BEAM WIDTH: 2' ELEVATION AND AZIMUTH 

IF BANDWIDTH: lKHz TO 200MHz 



145mm 
DIA 



OSCILLATOR 




PIN SWITCH 
MATRIX 



FEED 
NETWORK 



INTERFACE - 
CONNECTOR 



MILLITECH AUTOMOBILE RADAR FRONT END 



_ / \ millitech 



I 

o 



3 






S 

in 
£' 

3 
a 

? 

a- 
3 



3 

n 
a- 
a 

Q 



88 



Third International Symposium on Space Terahertz Technology Page 21 

MATERIALS MEASUREMENT AND 
MANUFACTURING PROCESS CONTROL 



MAJOR CONSIDERATIONS 

■ DEMANDS EXTREMELY RUGGED SYSTEMS 

■ COST IS A CRITICAL FACTOR 

■ MOST INDUSTRIES ARE CONSERVATIVE AND NEED TO BE 
CONVINCED OF VALUE OF NEW SYSTEM 

■ WHAT ARE THE UNIQUE CAPABILITIES OF TERAHERTZ 
RANGE? 



APPLICATIONS: 

HIGH VOLTAGE CABLE INSPECTION 
PAPER MAKING 



Page 22 



Third International Symposium on Space Terahertz Technology 



PAPER MEASUREMENTS AT SUBMILLIMETER WAVE LENGTHS 



BEAM FROM 
FIR U>SER 



BEAM SPLITTER 




PAPER SAMPLE 



D1 



FPI 



D2 



1.0 



0.8 - 



[A 


0.6 


U) 




2 




W 




-z. 
< 


0.4 


a: 




\— 






0.2 








\ 




_ x 

X 


70 fi 


\ 


96 \i 




V 1.18^ 


- 


\ 570^ 




\ 

** 


\ 


X 




X 

\ 


\\ 


, X 


X. 


\ 


\ N 


\ 


\ 






\ ^^ .. 


1 


— I ~i r- i ~ "" i i i i 



20 



40 



60 



80 



100 



% MOISTURE 

TRANSMITTANCE OF 80 fiM NEWSPRINT 
AS A FUNCTION OF MOISTURE CONTENT 



FROM BOULAY; ET AL, IR & MM WAVES, VOL. 5, PP 1221-1234, 1984 



Third International Symposium on Space Terahertz Technology Page 23 

CONCLUSIONS 



APPLICATIONS OF COHERENT SYSTEMS IN TERAHERTZ RANGE ARE 
EXTREMELY DIVERSE AND ARE EXPANDING 



RAPID TECHNICAL PROGRESS IS TAKING PLACE ON MANY FRONTS 



TRANS - MILLIMETER REGION IS NOW SIMILAR TO MILLIMETER RANGE 
JUST A FEW YEARS AGO AND A < 3 MM RANGE IS COMPARABLE TO 
MICROWAVE REGION IN RECENT PAST 



REAL SUBMILLIMETER REGION STILL HAS MANY CHALLENGES 
INCLUDING BASIC QUASIOPTICAL COMPONENTS, FREQUENCY SOURCES, 
ANTENNAS (INCLUDING ARRAYS) AND HIGH EFFICIENCY AND RUGGED 
MIXERS AND DETECTORS 



AN IMPORTANT CONSIDERATION: DD7FERENT APPLICATIONS HAVE 
ENORMOUSLY DIVERSE REQUIREMENTS 



THE SINGLE GREATEST OBSTACLE TO BROADER COMMERCIAL AND 
INDUSTRIAL UTILIZATION OF TERAHERTZ REGION IS COST 



WE NEED TO MAKE IT CHEAP AS WELL AS GOOD ! 



I WOULD LIKE TO ACKNOWLEDGE CONSIDERABLE ASSISTANCE FROM 
J. BIRCH, P. EFTHIMION, R. GILES, D. KEAVENEY, R. MCINTOSH, E. 
MOORE, A. PARRISH, P. SIEGEL, J. WATERS AND OTHER CO-WORKERS 
AT MILLITECH AND AT F. C. R. A. 0. 



Page 24 Third International Symposium on Space Terahertz Technology 

3z-**~ N93-27728 



Is 



Broadband Millimeter- Wave GaAs Transmitters and Receivers 
D Using Planar Bow-Tie Antennas 



Y. Konishi*, M. Kamegawa*, M. Case, R. Yu, M. J. W. Rodwell, R. A. York, 

and D. B. Rutledget 

Department of Electrical and Computer Engineering. 

University of California, Santa Barbara 

*On leave from Shimadzu Corp. Kyoto, Japan. 

^Division of Engineering and Applied Science. California Institute of Technology 



Abstract 

We report broadband monolithic transmitters and receivers ICs for mm-wave 
electromagnetic measurements. The ICs use non-linear transmission lines (NLTL) and 
sampling circuits as picosecond pulse generators and detectors. The pulses are radiated and 
received by planar monolithic bow-tie antennas, collimated with silicon substrate lenses and 
off-axis parabolic reflectors. Through Fourier transformation of the received pulse, 30-250 
GHz free space gain-frequency measurements are demonstrated with = 0.17 dB accuracy, 
RMS. 



Introduction 

For mm-wave and sub-mm wave gain-frequency measurements, convenient, 
broadband power sources and detectors have been required for some time. Measurement 
systems based upon waveguide components (harmonic mixers, frequency multipliers, and 
horn antennas)[l] have played a dominant role, but each component has narrowband 
frequency coverage (1.5:1). To measure over a broad bandwidth, many waveguide 
systems must be used, which is both inconvenient and very expensive. In addition, above 
100 GHz it is difficult and expensive to machine the small waveguides and difficult to 
attain efficient device-waveguide coupling. Broadband monolithic mm-wave ICs address 
these difficulties. 

Several groups have reported superconductor devices such as SIS (Superconductor- 
Insulator-Superconductor) detectors[2-4] or oscillators [5] for mm-wave measurement or 



Third International Symposium on Space Terahertz Technology Page 25 

for radio astronomy. Popular devices based on niobium technology (e.g., Nb/A10x/Nb 
junctions) must be cooled to liquid helium temperature, so a large and expensive cooling 
system is required. Additionally, due to the very low impedance of superconducting 
devices (» 0.1 Q), impedance matching to a 50 Q system is difficult 

Antenna-coupled picosecond photoconductors have also been used to generate and 
detect picosecond radiated electromagnetic pulses. Though Fourier analysis of the received 
signals, several groups have recently demonstrated broadband spectroscopy (~ 50 GHz- 
1.5 THz) [6-8]. Such systems require expensive and complex mode-locked lasers (= 
$150,000) to excite the photoconductors, and the radiated power is extremely small. 

As with the photoconductive systems, our system for mm-wave measurements 
radiates and detects picosecond pulses and obtains frequency information through Fourier 
transformation. Our system uses solid-state monolithic devices, NLTLs and sampling 
circuits for pulse generation and detection[9-ll]. With the NLTLs, we have several 
advantages. First, the system has fewer components and is very compact without the laser 
or its optics. Second, there is substantially more radiated power than the photoconductive 
system. Third, since the NLTL is driven by a microwave synthesizer and the NLTL input 
frequency can be varied by as much as one octave, the system can easily be tuned to any 
desired mm-wave harmonic frequency. Finally, the transmitters and receivers are 
inexpensive components fabricated on GaAs with a 5 mask process at 3 |im device 
geometries. No cooling system is required for GaAs ICs as with the superconducting 
devices. 

Here we will describe the system, especially the broadband bow-tie antenna and its 
optics. We have demonstrated the system performance by spectroscopic measurement of a 
thin alumina substrate with accuracy of 0.17 dB RMS and reproducibility better than 0.3 
dB from 30 to 250 GHz. 

NLTLs & sampling circuits 

The NLTL is a ladder network of high impedance transmission line sections 
periodically loaded with reversed biased monolithic Schottky diodes serving as voltage- 
variable capacitors[9]. The resulting voltage-variation in wave propagation velocity results 
in the compression of negative-going wavefronts and the formation of picosecond shock- 
waves. The NLTL converts an input 7-14 GHz sine wave to a sawtooth waveform. In on- 
wafer measurements, ~ 1.5 ps falltime and « 5 V peak to peak voltage swing has been 
attained. NLTL-gated sampling circuits attained similar risetime. Such devices allow 



Page 26 



Third International Symposium on Space Terahertz Technology 



generation and detection of transient signal with » 250 GHz bandwidth. The transmitter 
NLTL is typically driven by a 10 GHz + 100 Hz sinusoidal wave from a microwave 
synthesizer. This NLTL drives an on-wafer bow-tie antenna. The receiver consists of an 
NLTL-gated sampling circuit integrated with a bow-tie antenna. The NLTL which 
generates the sampler's strobe pulse is typically driven by a 10 GHz sinusoidal waveform 
from a second synthesizer. The resulting sampled 100 Hz IF signal is observed on a 
standard oscilloscope. 

Antenna and quasi-optical system 

In the case of a planar antenna on a dielectric substrate, most of the power is radiated 
into the substrate, and is trapped. This causes standing waves and resulting resonances 
within the GaAs substrate (er= 13). To avoid this, hyper-hemispherical substrate lenses are 
used with the bow-tie antennasf 12, 13]. 



NLTL 




hyper- 
hemispherical 
lens 

attenuators 



NLTL-gated 

sampling 

circuit 



// 



^ 



/ 



off-axis 

paraboloidal mirror material or array 

under test 




Figure 1: Measurement system schematic diagram (left-' transmitter, right: receiver) 

The output of the transmitter NLTL is connected by a coplanar waveguide (CPW) feed 
line to the feedpoint of the bow-tie antenna. This structure also serves as a balun. 
Sawtooth waves generated by the NLTL are radiated from the antenna. The bow-tie 
antenna is scale-invariant and has frequency-independent radiation impedance and 
frequency-independent far-field radiation patterns as long as its linear dimensions are larger 



Third International Symposium on Space Terahertz Technology Page 27 

than a free space wavelength. The antenna thus acts as a high pass filter, with the 2 mm 
length resulting in a ~ 35 GHz low-frequency cut-off[14]. The 55 |im total width of the 
CPW feedline defines a ~ 1.3 THz upper frequency limit for the antenna. 

The radiation is extracted through a silicon (er= 11.8, 16 mm diameter) hyper- 
hemispherical substrate lens on the back side of the IC. Matching of the IC and lens 
dielectric constants is very important. For example, a sapphire lens (er= 9.9) causes 
standing waves in GaAs substrate due to the discrepancy in er. This results in substantial 
resonances at 60 GHz, 120 GHz and 180 GHz. Compared to hemispherical lenses, hyper- 
hemispherical lenses improve the poor numerical aperture of the bow-tie antennas, and 
provide defocusing of the parasitic reflections arising at the lens-air interface. In contrast, 
hemispherical lenses exhibit strong spherical-mode resonances. The radiated beam is 
collimated with off-axis parabolic mirrors, and is focused on the receiver through similar 
optics. The antenna system loss, including substrate lenses absorption, coupling loss 
between the antenna and the lens etc., is = -20 dB as determined by 10 MHz - 40 GHz 
network analysis! 14]. 

Metal surfaces surrounding the experimental apparatus are covered with microwave 
absorber (Emerson & Cuming, FGM-40) to suppress reflections. Additionally, imaging the 
transmitter antenna onto the receiver produces a resonant cavity because of reflections at the 
air-lens and lens-antenna interfaces. To obtain accurate gain-frequency measurements, 
these resonances are suppressed by placing =» 5 dB thin-film metal attenuators on both sides 
of the sample under test. 

Device Fabrication 

The circuits were fabricated on GaAs semi-insulating substrates with a five mask 
process at 3 jtm design rules. Schottky diodes are formed on GaAs with a 425-nm-thick 
exponentially graded N" active layer with a 2x10*7 cm"3 surface doping and 225 nm 
exponential grading constant. Beneath the N" layer, a buried 1 Jim-thick N + layer (6x10*8 
cm"3) provides the diode cathode connection. Ohmic contacts to the N + later (the diode 
cathode connections) are formed by a 0.5 \un recess etch to the N + layer, a self-aligned 
AuGe/Ni/Au liftoff, and subsequent alloying. Proton implantation (masked by 1.6 nm gold 
on 1.1 ^im polyimide) provides isolation between diodes and defines Schottky contact 
areas. The transmission line sections are implemented in CPW, formed with a 1.1 |J.m 
Ti/Pt/Au liftoff; Schottky contacts result where this liftoff intersects unimplanted regions. 
With two additional mask steps, air-bridge crossovers are formed. 



Page 28 



Third International Symposium on Space Terahertz Technology 



Results 

The received signal (Fig. 2) shows that the sawtooth waveform has changed to a pulse 
train with initial fast rise and a decay time set by the antenna system's low-frequency cut- 
off. The peak-peak amplitude is 167 mV, and the pulse risetime is 2.6 ps as limited by the 
speed of sampling circuits, the NLTL, and the antenna system. 

Because the far-field radiation pattern is frequency-independent, the antenna effective 
aperture size is proportional to A 2 . Consequendy, misalignment selectively attenuates high- 
frequency components and limits the system bandwidth. With poor alignment, the pulse 
risetime degrades due to the reduced bandwidth. 



> 
E 



3 
Q. 

O 

> 
'<D 
O 
<D 

cr 




-250 - 



-300 



2.6 ps risetime, 1 0%-90% 
1 67 mV peak-peak 



-i — i — i — i — | — i — i — i — i — | i i — i — i — | — i — i — i i 

5 10 15 

time, ps 

Figure 2: Received waveform. 



20 



To demonstrate the system accuracy, we measured the insertion loss of a 254 p.m 
alumina substrate (er= 9.9). From 30 to 250 GHz the measurement values correspond well 
to theory. (Fig.3) With three subsequent measurements, the accuracy attained was 0.17 dB 
RMS, and the reproducibility was better than 0.3 dB. 



Third International Symposium on Space Terahertz Technology 



Page 29 



I I I I L_l I L. 



_1 I I ■ i i 




50 100 150 200 250 

Frequency.GHz 

Figure 3: mm-wave measurement of 254 urn-thick alumina test sample. 



S -10 



CO 

03* 
CO 

o 



c 
o 

CD 
CO 

c 



-20 - 



-30 - 



-40 - 



-50 



J. i i_i i L_i i_ 



_L 



a a 



• o • 
x 



o first measurement 
x second measurement 



* o 
x » 

Ox £ 

x 

x * 
o - 



-i 1 1 1 1 1 1 1 1 j—r—r—i 1 1 1 1 1 r- 

50 100 150 200 

Frequency.GHz 

Figure 4: mm-wave measurement of microwave absorber. 



We also measured the insertion loss of a microwave absorbing material (Emerson & 
Cuming FGM-40, 1.0 mm thickness ). (Fig.4) A loss minimum is seen at 60 GHz, with 
the attenuation improving at higher frequencies. 



Page 30 Third International Symposium on Space Terahertz Technology 

Above 150 GHz, this measurement is limited by the ~ 35 dB system dynamic range. 
This dynamic range can be gready improved by using narrowband signal detection (e.g., a 
lock-in amplifier). 

Conclusion 

We have demonstrated a simple and inexpensive system for broadband mm-wave 
electromagnetic measurements. Reproducible, accurate measurements are possible from 30 
to 250 GHz. The combination of the bow-tie antennas and the substrate lenses provides 
acceptable coupling efficiency over a broad bandwidth, despite the high systems loss (« 
-20 dB between antennas) and the additional (= 10 dB) attenuation required to suppress 
standing waves. The bow-tie antenna is readily integrated with monolithic circuits. 

The current system will allow convenient and accurate measurement of materials and 
emerging mm-wave quasi-optical amplifier arrays. With attainable improvements in the 
diode cut-off frequency, system bandwidth can potentially be extended to 1 THz. 

Acknowledgment 

This work was supported by the Air Force Office of Scientific Research under grant 
number (AFOSR-89-0394) 

References 

[1] Tektronix, Inc. 1991 Catalog 

[2] L. R. D'Addario, "An SIS mixer for 90-120 GHz with gain and wide bandwidth", Int. J. of IR and MM 

waves, Vol. 5 , No.ll, pp. 1419-1433, 1984. 

[3] T. H. BQttgenbach, R. E. Miller, M. J. Wengler, D. M. Watson, T. G. Phillips, "A Broadband Low 

Noise Receiver for Submillimeter Astronomy", IEEE, MTT-S. Digest, pp. 469-472, 1988. 

[4] S. Kodaira, J. Inatani, K. Sakai, T. Fukushima, "Phase Locking of SWL Array Junctions in 

Submillimeter Mixing", Jpn. J. Appl. Phys. Vol. 29, No. 3, pp. L463-L465, March, 1990. 

[5] J. Inatani, Y. Konishi, K. Sakai, and S. Kodaira, "Flux-Flow Oscillator connected with a Bow-Tie 

Antenna", ISEC, Tokyo, June, 12-13, 1989. 

[6] D. H. Auston and M. C. Nuss, " Electro-optic generation and detection of femtosecond electrical 

transients", IEEE, Quantum Electron., Vol. 24, pp.184-197, 1988. 

[7] G. Arjavalingam, Y. Pastrol, J. M. Halbut and G. V. Kopcsay, "Broad-band microave measurements 

with transiet radiation from optelectronically pulsed antenna", IEEE, Trans. MTT., Vol. 38, No.5, pp. 615- 

621, May, 1990. 



Third International Symposium on Space Terahertz Technology Page 31 

[8] N. Katzenellenbogen and D. R. Grischkowsky, "Efficient generation of 380 fs pulses of THz radiation 

by ultrafast laser pulse excitation of a biased metal-semiconductor interface", Appl. Phys. Lett., Vol.58, 

No.3, pp. 222-224, January, 1991. 

[9] M. J. W. Rodwell, M. Kamegawa, R. Yu, M. Case, E. Carmen, K. S. Giboney, "GaAs Nonlinear 

Transmission Lines for Picosecond Pulse Generation and Millimeter-Wave Sampling", IEEE, Trans. 

MTT., Vol. 39, No.7, July, 1991. 

[10] R. Yu, M. Case, M. Kamegawa, M. Sandram, M J. W. Rodwell and A. Gossard, "275 GHz 3mask 

Integrated Sampling Circuit", Elect Lett., Vol. 26, No. 13, pp. 949-951, June, 1990. 

[11] R.A. Marsland, C. J. Maden, D. W. Van Der Weide, M. S. Shakouri, and D. M. Bloom, "Monolithic 

Integrated Circuits for MM-Wave Instrumentation", in Technical Digest, GaAs IC Symposium, New 

Orlens.La. October, 7-10, 1990. 

[12] D. B. Rutledge, D. P. Neikirk, and D. P. Kasilingam. "Integrated-Circuit Antenna" in Infrared and 

Millimeter Waves, K. J. Button, Ed., Vol. 10, pp. 1-90, New York: Academic Press, 1984. 

[13] R. C. Compton, R. C. McPhedran, Z. P. Popovic, G. M. Rebeiz, P. P. Tong and D. B. Rutlegde, 

"Bow-Tie Antennas on a Dielectric Half-Space: Theory and Experiment", IEEE, Trans. Antenna Propag., 

AP-35, pp. 622-631, June, 1987. 

[14] M. Kamegawa, Y. Konishi, M. Case, R. Yu, and M. J. W. Rodwell, "Coherent, Broadband 

Millimeter-Wave Specroscopy Using Monolithic GaAs Circuits", LEOS Summer Topical Meetings, 

Newport Beach, July, 24-26, 199 1 . 




Page 32 Third International Symposium on Space Terahertz Technology 

ACTIVE CPW-FED SLOT ANTENNAS FOR POWER 
COMBINING APPLICATIONS 

Brian K. Kormanyos and Gabriel M. Rebeiz 

NASA/Center for Space Terahertz Technology 
Electrical Engineering and Computer Science Department 
University of Michigan . . " 

Ann Arbor, MI 48109-2122 ^93-27729 

ABSTRACT 

We have combined integrated circuit antenna technology with microwave oscillator design 
to build an active slotroscillator. The design is planar, does not require via holes and 
is compatible with monolithic transistor technology. The CPW-fed antenna impedance is 
calcualted using a full-wave analysis technique. Slot-oscillators were built at 7, 13, and 22 
GHz and the predicted oscillation frequencies agree well with experiments. The design is 
easily scaled to millimeter-wave frequencies and can be extended to power combining arrays. 

INTRODUCTION 

Millimeter-wave systems are becoming increasingly important in many military and com- 
mercial applications. Millimeter-wave receivers and transmitters have been traditionally 
waveguide-based systems and these are expensive to build at these frequencies. To solve 
this problem, several groups researched quasi-optical power combining topologies and active 
antennas [1-4]. In this paper, we present a novel active transmitter suitable for low-cost 
millimeter- wave applications. The transmitter consists of a cpw-fed slot antenna (or a dual- 
slot antenna) on a high-dielectric substrate- lens and a three- terminal device (millimeter- wave 
HEMT). The novelty in this approach is that we use the antenna impedance, calculated by 
a full-wave analysis method, as a parameter in the design of the oscillator. This results in 
a much more compact circuit than the conventional approach which consists of an oscillator 
with a 50Q output that is matched to a radiating slot-antenna. In our design, the matching 
network is eliminated (or minimized), the circuit is much smaller than a wavelength and this 



Third International Symposium on Space Terahertz Technology Page 33 

allows the design of a power combining array without trigerring grating lobes. The design 
can be easily scaled to millimeter wavelengths when HEMT transistor technology is available 
at these frequencies. 

OSCILLATOR DESIGN AND MEASUREMENTS 

The oscillator design is based on the S-parameters of the transistor used. An indefinite 
scattering matrix is employed so that short circuited lengths of CPW may be placed at the 
gate and source. Computer optimization is then applied to the lengths of CPW to maximize 
the reflection coefficient at the drain of the device. In this way a reflection magnitude greater 
than one is obtained without the use of an external feedback network and its associated 
complications. A slot antenna is connected to the drain through a length of CPW. In order 
for oscillations to build up, the impedance the slot antenna presents to the drain must have a 
reflection coefficient magnitude at least as large as the reciprocal of the reflection coefficient 
at the drain and the phase must be opposite in sign. The impedance of the CPW fed slot- 
antenna on a substrate must be well known and is calculated by a full wave moment method 
analysis. The terminals of the FET are DC isolated from eachother to allow bias voltages to 
be applied. This is done by integrating metal-insulator-metal capacitors and bypassed slits 
in the ground plane. 

Slot-oscillators were designed and built at 13GHz and 22GHz (Fig 1.) using commercially 
available hetero-junction FETs (NE32100, NE32184). The circuits oscillated near the pre- 
dicted frequency when placed at the focus of a one inch diameter elliptical silicon substrate 
lens (Fig. 1). The radiation patterns of the oscillators on the substrate lens were measured 
(Fig. 2) and are used to estimate the directivity. Total oscillator power is calculated with 
the radar equation. The total radiated power measured was 5.4mW at 13.01GHz and 3mW 
at 22.45GHz. The DC to RF efficiency is 5.4% at 13.01GHz and 3.8% at 22.45GHz. These 
numbers are consistent with the capability of the transistor which is a low noise small signal 
devices operated at maximum bias. In the future medium power transistors will be used. 



Page 34 



Third International Symposium on Space Terahertz Technology 



0.312mm 



0.25mm 



/, mmj^ 




3.38mm 



NEC I „ 
G I NE32184 I D 



S^ 



3.96mm- 






0.073mm 



S tarn > 

1.192 / 
mm • 



0.122mm 



1.758 

ezzzzzzzzzzzzz 

IJ-a 

ZZZZZZZZZZZZZ21 



* E3 



1.32 



2ZZZ2ZZZZZZZZ 



NEC 



X 



/ 



NE32100 ^ S 

/ 









802mm — •■ %£ 

1 



Figure 1: 13GHz and 22GHz slot-oscillator designs. 



1 1 ii i t ii 1 1 1 1 1 1 ii 1 1 ii i u-y i Hi" m ii 1 1 ii 1 1 ii 1 1 ii i ii 1 1 ii i 



CD 

a -io 

a 
o 



4) 






-15 



-20 



-25 



-30 




-Pol (E)- 
-pol (H>- 



■-- X-Pbl (E) " 

x-Poi (H) : 

' I l l II I l I I I I I II I 111 I Ill III I Ml I 



I I I 1 1 I I I I I M l | I II I IL^U !» ^AH M I I | I I I I I I I I I |l I I I I I I I I 



-60 -40 -20 20 40 

Angle (degrees) 



60 




-20 o 20 

Angle (degrees) 



Figure 2: Radiation patterns of 13GHz and 22GHz slot-oscillators on one inch diameter 
silicon substrate lens. 



Third International Symposium on Space Terahertz Technology 



Page 35 



A 7GHz VCO (Fig 3.) was designed using the above method with the incorporation of varac- 
tor diodes (Metelics MSV34-60-E28) at the source terminals of the FET. An oscillator tuning 
range of 850MHz was achieved from 6.68GHz to 7.35GHz. This shows that electronically 
tunable slot-oscillators are possible for phase locked loops or other applications. 



0.25mm 



T 



VA ° 5 P3 

// mm £ 



% 



mm 



0.776 // // 



Z 



g g 



0.312mm 



1 

mm /, 



2.0 



G I 




Metelics 
MSV34-60-E28 



Figure 3: 7 GHz VCO with 850 MHz tuning range 



The oscillators are well suited for use in power combining arrays synchronized by the mutual 
coupling between antennas. One possible array configuration (Fig. 4) is to place a two 
dimensional array on a dielectric block of quarter wavelength thickness. Most of the power 
will radiate out the opposite side of the block. A weak substrate mode will exist in the 
block and may enhance the mutual coupling. If neccessary a reflector may be used on the 
back side of the block to further enhance mutual coupling and improve phase equalization 
between elements. 



Page 36 



Third International Symposium on Space Terahertz Technology 



X/4 



Reflector 



\ 



\ Slot-Oscillators 

TMO 



k * 



Dielectric Block 




T t t Y 
adiated Power 



Figure 4: Possible slot-oscillator array configuration 



ACKNOWLEDGEMENTS 

This work is supported by the AF/ Rome-Air Development Center and by the NASA/Center 
for Space Terahertz Tehnology at the University of Michigan. We thank Prof. Linda Katehi for 
providing us with the full- wave solution of a cpw-fed slot antenna on an infinite dielectric substrate. 



REFERENCES 



[1] J.W. Mink, "Quasi-Opical Power Combining of Solid State Millimeter wave Sources," IEEE 
Trans. Microwave Theory Tech., vol. 34, pp. 273-279, Feb. 1986 

[2] Z.B. Popovic, R.M. Weikle, M. Kim, and D.B. Rutledge, "A 100 MESFET Planar Grid oscilla- 
tor," IEEE Trans. Microwave Theory Tech., vol. 39, pp. 193-199, Feb. 1991 

[3] R.A. York, and R.C. Compton, "Quasi- Optical Power Combining Using Mutually Synchronized 
Oscillator Arrays," submitted to IEEE Trans. Microwave Theory Tech., Oct. 1990 

[4] N. Camilleri and T. Itoh, "A Quasi-Optical Multiplying Slot-Array," IEEE Trans. Microwave 
Theory Tech., vol. 33, pp.1189-1195, Nov.1985 



Third International Symposium on Space Terahertz Technology Page 37 

2x2 QUASI-OPTICAL 
POWER COMBINER ARRAY 

AT 20 GHz N9 3-^53^30 

/6>oSl3 

Shigeo Kawasaki and Tatsuo It oh fL % 

Department of Electrical Engineering 
University of California 
Los Angeles, CA 90024 

ABSTRACT 

Investigation of a power combiner made of two FET oscillators for an active array 
1 are reported. As an approach by a quasi-optical method, a two-dimensional planar array of 
strongly coupled oscillators by direct connection through a microstrip line is used. In- 
phase condition between the oscillators as well as in-phase condition of each radiation wave 
was accomplished by regulating length of feed microstrip lines. The radiation elements of 
lX.-slot are embedded in a circuit ground plane. At an operation frequency of 20 GHz, in 
both H- and E-planes, reasonable L radiation patterns were obtained which have good 
agreement with theoretical patterns. 

INTRODUCTION 

Quasi-optical circuits have been of growing interest for compact and simple 
microwave and millimeter-wave systems. Among many solid-state devices, FETs are 
preferred as an active source for applications based on the MMIC technologyfl]. 
However, due to low efficiency, individual FET has limited capability of the power 
generation at higher operating frequencies. Therefore, a power combining technique is 



Page 38 Third International Symposium on Space Terahertz Technology 

essential for high power and high frequency systems. Further, the need for such systems 
has resulted in a variety of power combining configuration using FETs. 

As one of such power combiners, the planar grid oscillator using MESFETs has 
been developed in a distributed fashion[2]. Also, the weakly coupling oscillator array in a 
periodic fashion supported by a reflector element has been reported[3]. Recently, using 
external injection locking, the power combiner array with feedback FET oscillators was 
made[4]. However, due to a multimoding problem, the power combiner array with strong 
coupling has not made great progress to date. Regarding linear spatial power combiner 
arrays with strong coupling, we have already reported a few results obtained under the 
stratified structure[5],[6]. Through these results, in-phase operation for high combining 
efficiency was facilitated. 

In this paper, we describe design and experimental results of a quasi-optical spatial 
power combiner array made of two negative resistance HEMT oscillators and 4 slots. 
Radiation patterns from the slots aligned as 2x2 were compared with theoretical patterns in 
both H-plane and E-plane. Further, through these patterns, it was confirmed that the 
radiation waves from each slots were in-phase. 

DESIGN 

In order to demonstrate topology for the MMIC technology, the circuit was made in 
a layered structure by adopting slot radiators in a circuit ground plane. Fig. 1 shows the 
configuration of 2x2 spatial power combiner array designed at 20 GHz. The circuit 
structure made of two negative resistance oscillators and feed lines is etched on the top side 
of the substrate, while the slot radiators shown by cross-hatched sections in Fig. 1 are 
embedded in the ground plane of the bottom side of the same substrate. Each oscillator 
was designed with -50 Q for an input impedance. Since both sides of the substrate can be 
effectively used, the slot radiators in the circuit ground plane increase design flexibility. 



Third International Symposium on Space Terahertz Technology 



Page 39 



RF energy generated from one FET oscillator is divided and delivered to two slots 
aligned in the E-plane. In addition, a part of this energy is used to lock the other oscillator 
through a direct coupling between the two oscillators. The locked and amplified signal 
flows into another two slots aligned in the E-plane as well. Each pair of the slots aligned in 
the E-plane are connected in series. Therefore, this 2x2 array consists of two slots aligned 
in the E-plane and two slots aligned in the H-plane with W separation between the centers 

of two slots. The direct connection of two slots in H-plane separated by a thin line 
enhances high packaging density. 



1 



u 



I 



SSS¥ 



m 



11 



jsr 



D 



FET • CU 



a 



n 



3- 



2Zc 



B 

Zo 



J 



FET 



□ 







V/.W 

.vX;.y 






1L 



i+4 



T 



1L 






Slot Radiator 
(Bottom Side) 

*- X/4 

Transition 



Transformer 
/2Zo 



Fig. 1 Circuit Configuration 



Page 40 Third International Symposium on Space Terahertz Technology 

According to the previous results about the direct connection of oscillators[6], when 
the distance between two branch-points from the oscillators (shown as A and B in Fig. 1) 
is an odd number of half wavelength, the radiation waves from the two branching points A 
and B are in-phase. Since the length of a straight line between A and B is IX, a microstrip 
line with X/2 should be added. As a result, the radiation waves from the slots in H-plane 
can have the same phase. 

The distance between two centers of slots is IX. Thus, the divided energy from 

one branching point A or B can reach the centers of both slots at the same time. This 
results in an anti-phase radiation. For the in-phase condition, one of these pass must have 
additional X/2 to invert the phase on one of the slots. By means of this control of pass 
length between an oscillator and a slot as well as between two oscillators, the in-phase 
condition was obtained in both H-plane and E-plane. 

In addition to the consideration of phase condition, input matching condition needs 
to be taken into account. A one-stage X/4 transformer is inserted from a center of a slot 

toward a branched feed line connected to the branch-point A or B. In order to 
electromagnetically couple a feed line with a slot radiator, a X/4 microstrip-to-slotline is 

adopted. 

On the other hand, the slot radiators were also designed at 20 GHz with a IX, slot 
length and a 0.081 X slot width. Since input impedance of the center feed \X slot radiator 
provides the circuit with 50 Q load, the slot and the 50 Q feed line are matched in the 
steady state oscillation condition. 

EXPERIMENTAL RESULTS 

The circuit were fabricated on 31 mil thick substrate with £,-=2.33, using NEC 
NE32484 package HJFETs. The negative resistance FET oscillator was optimized at 21 
GHz with 5 % margin by using small signal S parameters so that the actual circuit can also 



Third International Symposium on Space Terahertz Technology 



Page 41 




be operated at 20 GHz. The maximum operation frequency of 19.5 GHz was observed 
from the fabricated circuit as shown in Fig. 2. 
It is found that, in this two- 
device power combiner case, 
a frequency margin of 7.7 % 
is required under the design 
condition of -50 Q for an 
input impedance, in reference 
[1], we investigated the 
frequency margin of a single 
quasi-optical oscillator with a 

single slot using the same 

v 

NE32484. In this case, the 
frequency margin of 4.4 % 
was obtained. 

Compared with this previous results, the difference between the operation frequency and 
the design frequency is increased. This implies that, in the case of a large number of FET 
combiners, accumulation of a difference of input impedances due to such frequency 
difference becomes a cause of load pulling, and, due to this phenomenon, the frequency 
difference is increasingly enhanced. The tuning range obtained by changing the applied DC 
voltages, V ds (2.6-5.0 V) and V gs (-1.0—1.6 V), was 121 MHz. 

Radiation patterns in both the H-plane and the E-plane are shown in Fig. 3. In both 
cases, it is easy to find out null points around ±30° which result from array factor. For 
comparison, the theoretical analysis was carried out by using the point matching method to 
obtain numerical data from the Pocklington type integral equation. In the calculation, the 
voltage ratio of two source generators to excite each slot was set to 1, while the phase 
difference of these generators was set to 0. Using these data, theoretical radiation patterns 



Fig. 2 Operation Speetrum 



Page 42 



Third International Symposium on Space Terahertz Technology 



10 -i 



m 

•a 



a> 
3 
o 
a. 



a> 

> 







Angle (degree) 



experiment 
theory 



100 



Experiment 
(2x2 H-plane) 

Vds=4.0 V 

lds=45 mA 

Vgs=-1.81 V 

fo=19.14 GHz 
Theory 

Pockilngton I.E. 

Voltage Ratio=1.0 

Phase Difference=0 



Comparision in Radiation Pattern (H Plane) 



m 
■o 



o 
a 

> 

a> 



10-1 



0" 



-10- 



-20 



100 




100 



Angle (degree) 



experiment 
theory 

Experiment 
(2x2 E-plane) 

Vds=4.0 V 

lds=40 mA 

Vgs=-1.82 V 

f0=1 9.15 Ghz 
Theory 

Pocklington I.E. 

Voltage Ratio=1.0 

Phase Difference=0 



Comparision in Radiation Pattern (E plane) 



Fig. 3 Radiation Patterns 



Third International Symposium on Space Terahertz Technology Page 43 

are obtained as shown in Fig.3 for both the H-plane and the E-plane. Agreement around 
the main beams is good. 

Sidelobes in the E-plane becomes lager than those in the H-plane because of a 
single element factor. This fact is shown in the theoretical patterns in Fig. 3. However, the 
sidelobes obtained from the experiment were still large in both H-plane and E-plane. These 
discrepancies may result from the inadequate experimental setup. 

CONCLUSIONS 

Design and experimental results of a 2x2 quasi-optical spatial power combiner was 
reported as a prototype of two dimensional quasi-optical power combiner array. Although 
the impedance matching conditions as well as the in-phase condition have been taken into 
account carefully, the difference of the operation frequency (19.2-19.5 GHz) to the design 
frequency (20 GHz) has increased. In the case of a large array, more attention should be 
paid to avoid increase of this frequency difference resulting in phenomena such as load 
pulling. 

In both the H-plane and the E-plane of a 2x2 quasi-optical power combiner array, £ 

radiation patterns were obtained by controlling the lengths of feed lines. Good agreement 
between the experiment and the theory was obtained about mainlobes in both radiation 
patterns. 

It is believed that, through this prototype circuit, the fundamental data for a 
monolithic quasi-optical power combiner array were obtained. 

ACKNOWLEDGEMENT 

This work was supported by US Army Research Office under contract DAAL 03- 
88-K-0005. 



Page 44 Third International Symposium on Space Terahertz Technology 



REFERENCES 

[1] S. Kawasaki and T. Itoh, "24 GHz FET Oscillator with Slot Antenna for Quasi-Optical 

Transmitter", 16th In't Conf. on IR & MMW, Switzerland, Aug. 1991, pp 286-287 
[2] Z.B. Popovic, R.M. Weikle II, M. Kim and D.B. Rutledge, "A 100-MESFET Planar 
Grid Oscillator", IEEE Trans. Microwave Theory Tech., vol. 39, pp 193-200, Feb. 
1991 
[3] R.A. York and R.C. Compton, "Quasi-Optical Power Combining Using Mutually 
Synchronized Oscillator Arrays", IEEE Trans. Microwave Theory Tech., vol. 39, pp 
1000-1009, June 1991 
[4] J. Birkeland and T. Itoh, "Two-Port FET Oscillators with Applications to Active 

Arrays", IEEE Microwave and Guided Wave Lett., vol.1, ppl 12-113, May 1991 
[5] S. Kawasaki and T. Itoh, "40 GHz Quasi-Optical Second Harmonic Spatial Power 
Combiner Using FETs and Slots", to be published in 1992 IEEE MTT-S Int'l 
Microwave Symposium, Albuquerque, NM. 
[6] S. Kawasaki and T. Itoh, "6-Element Periodic and Nonperiodic Linear Arrays for 
Quasi-Optical Spatial Power Combiner", to be published in URSI Radio Science 
Meeting, Chicago IL, July 1992 



Third International Symposium on Space Terahertz Technology 



Page 45 



MONOLITHIC MILLIMETER- WAVE DIODE ARRAY 
BEAM CONTROLLERS: THEORY AND EXPERIMENT 



N93-27731 



L. B. Sjogren, H-X. L. Liu, F. Wang, T. Liu, W. Wu, X-H. Qin, E. Chung, %~~.33 
C.W. Domier, N. C. Luhmann, Jr. 
Center for High Frequency Electronics /(pO^l^f 

Department of Electrical Engineering 

University of California p /J 

Los Angeles, California 90024-1594 I " 

J. Maserjian 

Jet Propulsion Laboratory 

Pasadena, California 91109 



M. Kim, J. Hacker, D.B. Rutledge 

California Institute of Technology 

Pasadena, California 91125 

L. Florez, J. Harbison 

Bellcore, Inc. 

Red Bank, New Jersey 07701-7030 



~\ 



X 




y H 



I. Introduction 

Power-combining arrays of semiconductor devices offer a promising approach to 
the realization of compact, reliable, economical systems for watt-level operation at 
millimeter- wave and submillimeter- wave frequencies. Such ("grid") arrays have demon- 
strated numerous functions at microwave and millimeter-wave frequencies in recent 
experimental efforts. Monolithic diode arrays have demonstrated phase shifting at 93 
GHz [1], frequency doubling from 33 to 66 GHz [2] and frequency tripling from 33 to 
99 GHz [3]. One dimensional monolithic imaging arrays have been demonstrated at 
94 GHz [4]. Additional quasi-optical functions have been demonstrated at microwave 
frequencies by arrays employing hybrid technology. These include the oscillator grid 
[5-6], mixer grid [7], and amplifier grid [8]. The hybrid grids operate on the same 
basic quasi-optical principle as the monolithic arrays, so the functions demonstrated 
to-date in hybrid form should be feasible also in a monolithically integrated form at 
millimeter- wave frequencies. Additional array design approaches have been suggested 
for further development of millimeter- wave components [9], 

Construction of complete systems based on the millimeter- wave array technology 
requires not only source and detector arrays, but control components for such func- 
tions as amplitude modulation, phase modulation, and beam steering. The first effort 



y 



Page 46 Third International Symposium on Space Terahertz Technology 



at addressing this need by a semiconductor device array was an experimental demon- 
stration of a phase shifter at 93 GHz [1]. In this work, phase control of a reflected 
beam over a range of 70 degrees with 6.5 dB loss was achieved by a monolithic array 
of Schottky diodes. The stacking of more than one array [1] should allow a phase 
range of greater than 360 degrees to be achieved. Operated in a nonuniform phase 
(bias) mode, such an array should be capable of (phased array) beam steering and 
beam focusing. 

In the current work, multi-function beam control arrays have been fabricated, 
and have successfully demonstrated amplitude control of transmitted beams in the 
W and D bands (75-170 GHz). While these arrays are designed to provide beam 
control under DC bias operation, new designs for high-speed electronic and optical 
control are under development. These arrays will fill a need for high-speed watt-level 
beam switches in pulsed reflectometer systems under development for magnetic fusion 
plasma diagnostics. 

A second experimental accomplishment of the current work is the demonstration 
in the 110-170 GHz (D band) frequency range of a new technique for the measurement 
of the transmission phase as well as amplitude [11]. Transmission data can serve as a 
means to extract ("de-embed") the grid parameters; phase information provides more 
complete data to assist in this process. 

Additional functions of the array beam controller yet to be tested include electron- 
ically controlled steering and focusing of a reflected beam. These have application in 
the areas of millimeter- wave electronic scanning radar and refiectometry, respectively. 



II. Theory and Design 

The beam control device consists of a monolithic two-dimensional array of varactor 
diodes embedded in metal strips. Its quasi-optical behavior is represented by a shunt 
impedance across a transmission-line representation of the beam. The impedance is 
that of a series RLC circuit. The inductance, some fixed (undesired bias-independent) 
capacitance, and a small resistance are due to the metalization grid structure; the 
bulk of the capacitance is due to the diode; the bulk of the (undesired) resistance is 
a parasitic effect of the diode and its Ohmic contact. 

The initial objective of the current effort was the design of the proposed stacked 
360 degree reflection phase shifter array [1]. In this approach, an effective load re- 
actance of arbitrary value is obtained by the use of two diode grids, each of which 
possesses a reactance range of at least -ZoaAs to Za a Aa, where Za a Aa is the plane wave 
impedance of Gallium Arsenide (approximately 105 Cl), and the grids are separated 
by an odd multiple of A/4. 

The capability of the array to control beam transmittance is based on its ability 
to be switched, under bias control, between a low and high impedance state. When 
the array is biased at resonance, it appears to the beam as nearly a short circuit, 
and reflects most of the beam back. When the array is biased far from resonance 
(i.e. at high impedance), the beam is affected very little by it. In this case, the beam 
transmittance is large. 



Third International Symposium on Space Terahertz Technology Page 47 



The dimensions and doping profile for the monolithic Gallium Arsenide Schottky 
varactor diode were determined with the assistance of a finite difference solution 
program for Poisson's equation in one dimension. This provided an estimate of the 
diode C-V characteristics. Additional routines were employed to estimate tunneling, 
avalanche, and thermionic emission currents. A hyperabrupt doping profile, similar to 
that employed by W. Lam [1], was chosen to provide linearity of reflection phase versus 
DC bias. Different doping levels were simulated to optimize the capacitance range 
and breakdown voltage. A heterojunction barrier, pioneered at JPL for frequency 
multiplier diodes, has been incorporated to suppress tunneling current under reverse 
bias and thermionic emission under forward bias. The experimental results in this 
paper were obtained with arrays fabricated from MBE wafers with the profile shown 
in Fig. 1. 

Design of the array requires the theoretical prediction of the electromagnetic (grid) 
behavior as well as the C-V characteristics of the embedded varactor diode. For the 
passive (electromagnetic) design, a simple method of moments analysis has been em- 
ployed. This has provided a new model which includes the effect of the discontinuity 
of the current at the site of the diode ("grid capacitance") [12] (Fig. 2). This model 
somewhat underestimates the fixed (bias-independent) capacitance at the diode site, 
since it idealizes the current discontinuity as that for a "gap" at the diode site when 
the diode is (analytically) open circuited. Even so, the capacitive effect was found 
to be considerable (5-10 fF for typical array geometries at 100 GHz), and sufficient 
to reduce the simulated phase range of the stacked phase shifter from well above 
360 degrees (with the "grid capacitance" not included) to a value considerably below 
360 degrees (with it included). A rigorous simulation of the C-V characteristics of 
the grid would require a three dimensional simulation which includes the effect of 
the anode "finger" , absence of dielectric beyond the etched walls of the diode, etc. A 
more precise determination of the grid inductance should be possible with the Hewlett 
Packard High Frequency Structure Simulator electromagentic analysis program. Such 
an analysis may provide a more precise estimate of the grid capacitance. 

The grid impedance model and diode model programs were combined and incor- 
porated into a quasi-optical (transmission line) circuit analysis program based on 
the application of Kirchoff's laws at the dielectric (and grid) interfaces. Simulations 
were performed to determine unit cell and strip dimensions which maximized the grid 
impedance range. The grid capacitance effect was suppressed by use of a "rectangular 
unit cell", in which diodes are effectively placed "in series" by spacing them closer 
along the axis of current flow. Such a configuration has the additional benefits of 
allowing use of a large (easier to reliably fabricate) diode and providing significantly 
higher power-handling capability. The one drawback of this design is a somewhat 
higher loss, since the effective diode resistance is a/b times the actual diode resis- 
tance. The increased size of the diode will partially, but not completely, offset this 
effect. A possible alternative for the reduction of grid capacitance is to employ a 
narrower metal strip in the vicinity of the diode [13]. This was not feasible for the 
current design, since the strip was too narrow to taper. 

The simulations provided the optimized unit cell dimensions a=300 /xm, b=120 
fxm, and w=7 fxm, with diode dimensions of 3 /xm x 13 fim. The diode area assumed 



Page 48 



Third International Symposium on Space Terahertz Technology 



Al Ga As 

O.S 0.5 

(undoped) 




Staircase approximation 
to hyperabrupt profile 



1S0A 


5.0 E17 cm-3 


100 A 


4.0 El 7 cm-3 


150 A 


3.0 E17 cm-3 


250A 


2.0 E17 cm-3 


350A 


1.5 E17 cm-3 


SOOA 


1.0 El7cm-3 


700A 


7.5 E16 cm-3 


1600A 


5.0 E16 cm-3 


3800A 


total hyperabrupt 



hyperabrupt n GaAs 



2000A 

2200A 



6000A 



Semi-insulating 
GaAs substrate 




16000A 



Figure 1: MBE profile of the beam control array. 



ei 




r ^ 



-7 

b *- diode 



\ 



=T<4 



Figure 2: Electromagnetic model of the diode array. 



Third International Symposium on Space Terahertz Technology Page 49 



in the simulations is 26 /im 2 , since the effective diode width after fabrication should 
be about 1 /im less than the drawn dimension. Due to the small cell dimensions, the 
device count per array is very high, about 12,000 for a full sized (2.5 cm x 1.8 cm) 
array. To allow reasonable spacing between wirebonds, the array was layed out so 
that each bias line connects to eight contiguous rows of 60 diodes each. Thus, each 
bias line connects to 480 array diodes. 

As previously stated, the array should show some capability for electronic beam 
steering and focusing. The array model cannot predict the extent of these capabilities, 
however, since it is applicable only to uniform array operation. 

III. Fabrication and Testing 

Array fabrication is based (with some modifications) on the self-aligned Aluminum 
Schottky diode process developed by C. Zah for millimeter- wave imaging diodes [4]. 
Most of the processing was performed at the J PL Microdevices Laboratory, a facility 
with state-of-the-art microfabrication capability. 

Device isolation was performed, as in [4], by proton implantation, with an im- 
plant mask of thick photoresist. A two-step implant of 4xl0 14 cm~ 2 at 200 keV and 
4xl0 14 cm~ 2 at 100 keV provided good isolation. However, photoresist edge bead 
resulted in the corners of the arrays being unisolated. Several of the arrays were 
therefore re-implanted to isolate these areas. However, the capacitance range of the 
diodes was greatly reduced after the second implant. It thus appears that some of 
the implant penetrates the photoresist mask and into the active device. The arrays 
which were not re-implanted had some unuseable areas due to the edge bead problem, 
but have proved sufficient for the experimental proof-of-principle testing. 

Due to the parallel connection of array diodes, short circuited devices must be dis- 
connected from the array. An HP4145 Semiconductor Parameter Analyzer, HP9836 
controller, and Electroglas 1034 wafer prober were combined into a system which 
allowed automated testing and storage of I-V characteristics for every array device. 
Device probing was facilitated by an extra metalization step which creates probe pads 
connected to each device prior to grid (bias) metalization. With the short-circuited 
devices identified, a microprobe was used to sever their connection to the array. Fol- 
lowing final metalization, very few of the bias lines were short-circuited. Therefore, 
further identification of short-circuited devices by "hotspot" detection [1] was not 
neccessary. Photographs of a single array device and a small region of an array are 
shown in Figs. 3 and 4, respectively. The final steps for completion of the array are 
the attachment and wirebonding of the array to a printed circuit "bias" board, and 
the attachment of bias wires to the board. Series resistors of 220 SI are added to the 
bias wires to prevent damage if an array device becomes short-circuited. 

TV. Results 

Millimeter- wave transmission testing was performed with the system shown schemat- 
ically in Fig. 5. A small array (1.8 cm x 1.0 cm, 4800 diodes) successfully demon- 
strated transmitted amplitude control throughout the W and D frequency bands 



Page 50 



Third International Symposium on Space Terahertz Technology 




Figure 3: Photograph of a single device in a beam control array. 




Figure 4: Photograph of a small section of the beam control array. Unit 
cell dimensions are 300 [im x 120 fim. The vertical strips serve as the 
"antenna" elements, while the horizontal strips provide bias voltage 
to the diodes. The marks on the large rectangular test pads are from 
automated device probing. 



Third International Symposium on Space Terahertz Technology p „ 



(75-170 GHz). Results at 99, 132, and 165 GHz are shown in Fig. 6. Substantial 
amplitude control was obtained (except near resonance), despite leakage currents on a 
number of the bias lines preventing most of the bias from showing up their associated 
sections of the array. In addition to amplitude control under DC bias, the array suc- 
cessfully demonstrated low frequency (200 kHz) modulation of a 165 GHz beam (Fig. 
7). The modulation frequency was limited by the bandwidth of the detected beam 
amplifier. Further testing will be performed to determine the maximum modulation 
frequency of the array. 

To allow further verification of the grid behavior, a technique has been developed 
to obtain the phase, as well as amplitude, of the transmitted beam [11]. The method 
involves tilting the incident beam, so that a portion of it is aligned orghogonal to the 
operational ("active") axis of the grid. The orthogonal beam component is employed 
as a reference, with the polarization of the transmitted beam providing the relative 
phase of the beam in the active versus orthogonal ("passive") axis. Since the phase 
of the passive axis is highly predictable, the absolute phase of the transmitted beam 
in the operating axis can be determined by this method. The method was verified 
by application to strip arrays, whose theoretical behavior can be well predicted and 
compared to the experimental results. The method was then applied to obtain the 
transmission coefficient of the beam control array as a function of frequency and bias. 
This was done for an 8600 diode beam control array. Estimates of parameters for the 
series RLC model of the grid were obtained by varying the parameters until a good 
match between the theoretical and experimental transmission curves was obtained. 
This prov ided a fairly precise estimate of grid resistance and resonant frequency f rea = 
(2iry/LC)~ 1 . For the array tested, the grid resistance is approximately 40 ft over the 
entire bias range, and the resonant frequency versus bias is shown in Fig. 8. The 
individual value of L (or C) has some range in which the theoretical and experimental 
curves agree. This range is centered at an inductance value of L= 160 pH, with the 
capacitance ranging from 5.2 fF to 13.9 fF as a function of bias for this value of 
L. The results indicate a cutoff frequency for the beam control array f c = (C~} n — 
C~\ x )(2irR)~ l of about 400 GHz. It appears that both the grid inductance and 
capacitance are considerably lower (about 35 %) than predicted. The deviation in 
the inductance value is probably due to the idealization of the inductive effect as that 
of a uniform (and narrower than actually fabricated) vertical strip. Simulation by the 
High Frequency Structure Simulator will provide a more definitive verification of this. 
The deviation in grid capacitance is with respect to the values expected based on 1 
MHz C-V measurements of sample diodes from the same wafer. This discrepancy 
requires further investigation. 

Initial reflection tests have been performed for "calibration" devices (strip grids), 
with use of a focusing lens [14] and the polarization technique [11]. The phase of the 
reflection coefficient has been successfully obtained by this method. Since the grid 
parameters of the currently fabricated arrays are largely now known, the reflection 
phase shift for a stack of two of these grids can be well-predicted. Predicted results at 
128.3 GHz (at which the thickness of the GaAs can be made the desired odd multiple 
of A/4), along with the originally simulated behavior at the design frequency of 99 GHz 
rescaled to 128.3 GHz, are shown in Fig. 9. The lower than desired C max /C m ,„ ratio 



Page 52 




Third International Symposium on Space Terahertz Technology 

diode grid 

aperture ^ 

diode 
detector 

>-w- 



i 



m 



■ i< 
• i. 

■ i 



1 

frequency ^ 

meter absorber ^ 



plate 



5£ 



Grid bias or 
modulation 



preamplifier ^^ 




Figure 5: Schematic diagram of the test system for transmitted beam 
testing. 



0.8 - 



o 

c 

2 0.6 



E 

W 

C 
CO 



0.4 



0.2 



i i • • • r 



i i i i i i i i 1 1 i i i i I i i i i i i i i i 



i ■ ■ ' ' i 



165 GHz 



99 GHz 




■ 2.5 -2 -1.5 -1 -0.5 0.5 1 

Array Bias Voltage (V) 



Figure 6: Experimental beam transmittance at 99 GHz, 132 GHz, and 
165 GHz as a function of DC bias applied to the array. The basic 
form of the curves can be understood by the fact that the array is 
capacitive at 99 GHz, resonant at 132 GHz, and inductive at 165 
GHz. 



Third International Symposium on Space Terahertz Technology 



Page 53 



AAW 



Detected output 




Bias modulation 



j2us 

Figure 7: Detected output versus array input voltage for a sinusoidal 
modulation of the array at 200 kHz. The lower waveform is the array 
bias modulation voltage, whose range was -3V to +1V. The upper 
waveform is the detected output from the transmitted beam. 



N 

K 
O 

>> 

o 

c 

3 

cr 

u 

c 
a 
c 
o 



i — i — i — i — I — i — i — i — i — 1 — i — i — I — i — i — I — i — i — \ — l — i — I — i — r 




-4 



-2 -1 

Bias (V) 



Figure 8: Resonant frequency versus DC bias for the beam control array. 



Page 54 



Third International Symposium on Space Terahertz Technology 



1 I — i — n — r— | — i — i — i — i — [— i — i — i — r 





3 



Q. 

E 
< 



-i — n — i — I — I — i — i — i — 1 — i — i — i — r 




Q I I I I i_l I I I I I I I I i_l i l_j i I i__i i i I ■ ■ ■ ■ 

-5 -4 -3 -2-10 1 

Bias (V) 



6.28 



CO 




c 


3.925 


CO 




■a 




CO 




k. 


1.57 


CD 




CO 




CO 





■0.785 



•3.14 



~ I I I I i i I r 



I I I — I — I — I — I — I — I I I I | — I — I — I" 




i ■ i i i i 



-5 -4 



3 -2 -1 

Bias (V) 



Figure 9: Predicted reflection coefficient for a two layer stacked phase 
shifter with fused silica "window" at 128.3 GHz. Lines with no mark- 
ers represent the predictions based on originally simulated grid pa- 
rameters. Lines with markers represent the predictions based on es- 
timates of the grid parameters based on transmission measurements. 



Third International Symposium on Space Terahertz Technology Page 55 



results in a phase range less than 360 degrees, contrary to the original simulation. 
This, however, does not preclude the possibility of beam steering to small angles. 
The higher than desired grid resistance results in a reflectance which is lower and is 
much more variable over the bias (phase) range. A preliminary attempt at steering 
of a transmitted beam with the current array was unsuccessful. This may be due 
to an inability of the grid to produce the strong amplitude variation with position 
associated with the desired phase distribution. Since steering to a fixed angle has 
been successfully demonstrated by a (low-loss) passive grid [15], the beam control 
array's ability to steer a reflected beam is likely to be governed largely by the grid 
loss as well as phase shift range. To obtain higher performance for the beam control 
functions, new arrays are being fabricated which should possess a larger C ma x/C m ,„ 
ratio and Ohmic contact resistance. In addition, we are considering the stacking 
of a large number of arrays of the current design and operating them in the high 
impedance region to accomplish transmitted beam steering. 

V. Conclusions 

A "second generation" millimeter-wave beam control array device has been con- 
structed. This array has successfully demonstrated a new function by a millimeter- 
wave quasi-optical array, that of beam transmittance control. Phase of the trans- 
mitted beam has also been measured by a newly-developed technique. Reflection 
measurements, which will test the arrays as phase shifters, beam steerers, and beam 
focusers, will be performed soon. In addition, new arrays with a modified doping 
profile for higher C max arid lower resistance are being fabricated. These "higher per- 
formance" arrays should provide, for example, a greater "contrast ratio" (maximum 
to minimum transmittance) when the array is operated as a beam modulator. 

New array designs have been initiated for high-speed (under 200 psec) electronic 
and optical beam control. For electronic control, the bias lines of the beam control 
array are being designed to function as high-speed guided-wave paths. For opti- 
cal control, monolithic arrays of photoconductive switch devices are being fabricated. 
New concepts for barrier varactor photodiodes are under study for application toward 
optically-controlled modulator and beam steering arrays. High-speed beam switch- 
ing arrays have immediate application in plasma diagnostic reflectometry, and have 
potential additional application for such functions as electronic input beam chopping 
in high-speed imaging systems. Longer term possibilities include exciting possibilities 
such as amplifying beam steeres with two-axis scan control. 

Acknowledgements 

Work supported by Northrop Corporation/ University of California MICRO pro- 
gram, Department of Energy, and the Army Research Office. 

The authors wish to thank the personnel of the JPL Microelectronic Devices 
Laboratory for their assistance toward the fabrication of the beam control arrays. In 
particular, we wish to thank R. Peter Smith, Suzanne Martin, Chuck Manning, Rich 
Muller, Judy Podosek, and Doug Waltman. 



Page 56 Third International Symposium on Space Terahertz Technology 



Essential to the project has been the availability of high-quality MBE wafers gen- 
erously provided by Prof. C. Jou (National Chiao Tung University, Taiwan), Prof. 
M. Spencer (Howard University), John Liu (JPL), as well as Bellcore, Inc. Assis- 
tance in this area has also been provided by Larry Kapitan (formerly with Northeast 
Semiconductor, currently with QED). 

The stepping wafer prober, along with technical assistance, was generously pro- 
vided by Edith Baltram, Jack Hayden, Frank Freeman, Soo Kok Leng, and Gary 
Castleman (Hewlett-Packard, Northwest Integrated Circuits Division). 

We gratefully acknowledge the essential assistance in Ohmic contact alloying by 
Dr. Marko Sokolich of Hughes. 

Indispensible assistance with millimeter- wave measurement systems was provided 
by Matt Espiau and Misti Christianson of the UCLA Millimeter- Wave Laboratory. 

Additional individuals who provided essential assistance for this project include 
Wayne Lam (TRW), Charles Meng, Prof. D.S. Pan, Prof. H.R. Fetterman (UCLA), 
Mike DiLisio (Cal Tech), Prof. R.J. Hwu (University of Utah), Clarence Becwar 
(Becwar Engineering), and Rene Bernescot (Rockwell). 

References 

(1) W.W. Lam, C.F. Jou, N.C. Luhmann,Jr., and D.B. Rutledge, "Millimeter-wave 

diode-grid phase shifters," IEEE Trans. Microwave Theory Tech., 36, No. 5, p. 
902, 1988. 

(2) C.F. Jou, W.W. Lam, H.Z. Chen, K.S. Stolt, N.C. Luhmann,Jr., and D.B. 
Rutledge, "Millimeter Wave Diode-grid Frequency Doubler," IEEE Trans, on 
Microwave Theory and Techniques, 36, No. 11, 1988. 

(3) H-X. King, X-H. Qin, W. Wu, L.B. Sjogren, E. Chung, N.C. Luhmann, Jr., 

W.A. Peebles, "Monolithic Millimeter- Wave Quasi-Optical Frequency Multi- 
plier Arrays", presented at, 1991 International Semiconductor Device Research 
Symposium, pp. 68-72, December, 1991. 

(4) C. Zah, D.P. Kasilingam, J.S. Smith, D.B. Rutledge, T. Wang, and S.E. Schwartz, 

"Millimeter- wave Monolithic Schottky Diode Imaging Arrays", Intl. J. of In- 
frared and Millimeter Waves, 6, pp. 981-997, 1985. 

(5) Z.B. Popovic, R.M. Weikle, M. Kim, K.A. Potter, and D.B. Rutledge, "Bar Grid 

oscillators," IEEE Trans. Microwave Theory Tech. , 38, No. 3, p. 225-230, 
March,1990. 

(6) Z.B. Popovic, R.M. Weikle, M. Kim, and D.B. Rutledge, "A 100-MESFET 
Planar Grid Oscillator," IEEE Trans. Microwave Theory Tech., 39, No. 2 pp. 
193-200, February, 1991. 

(7) J.B. Hacker, R.M. Weikle III, M. Kim, D.B. Rutledge, "A 100 Element Schottky 

Diode Grid Mixer", 1991 IEEE AP-S symposium digest. 



Third International Symposium on Space Terahertz Technology Page 57 



(8) M. Kim, J.J. Rosenberg, R. P. Smith, R. M. Weikle III, J.B. Hacker, M.P. DeLi- 

sio, D.B. Rutledge, "A Grid Amplifier", IEEE Microwave and Guided Wave 
Letters, I, No. 11 pp. 322-324, November, 1991. 

(9) R.J. Hwu, C.F. Jou, N.C. Luhmann,Jr., M. Kim, W.W. Lam, Z.B. Popovic, and 

D.B. Rutledge, "Array concepts for solid state and vacuum microelectronics 
millimeter wave generation," IEEE Trans. Elec. Dev., 36, No. 11, 1989. 

(10) L.B. Sjogren, R.J. Hwu, H-X. King, W. Wu, X-H. Qin, N.C. Luhmann, Jr., M. 
Kim, D.B. Rutledge, "Development of a 94 GHz Monolithic Quasi-Optical 360 
Degree Phase Shifter," Proc. of the 15th Intl. Conf. on Infrared and Millimeter 
Waves, pp. 696-698, 1990. 

(11) L.B. Sjogren, et. al., "A Technique for the Measurement of Complex Transmis- 
sion Coefficient of Millimeter- Wave Grid Arrays", to be submitted, Microwave 
and Optical Technology Letters, 1992. 

(12) L.B. Sjogren and N.C. Luhmann, Jr., "An Impedance Model for the Quasi- 
Optical Diode Array", IEEE Microwave and Guided Wave Letters, 1, No. 10, 
pp. 297-299, October, 1991. 

(13) H-X. King, L.B. Sjogren, N.C. Luhmann, Jr., D.B. Rutledge, "New Concepts 
for High Frequency and High Power Frequency Multipliers and Their Impact 
on Quasi-Optical Monolithic Array Design", Int. J. of Infrared and Millimeter 
Waves, Feb. 1992. 

(14) David R. Gagnon," Highly Sensitive Measurements With a Lens-Focused Re- 
flectometer" , IEEE Transactions on Microwave Theory and Techniques, 39, No. 
12, pp. 2237-2240, December, 1991. 

(15) Moonil Kim, Robert M. Weikle III, Jonathan B. Hacker, David B. Rutledge, 
" Beam Diffraction by a Planar Grid Structure at 93 GHz", 1991 IEEE APS 
Symposium. 



Page 58 Third International Symposium on Space Terahertz Technology 

5Z-33 






N93~t7732 



A Study of Subterahertz HEMT Monolithic 

Oscillators * 

Youngwoo Kwon and Dimitris Pavlidis 

Center for Space Terahertz Technology 

Solid State Electronics Laboratory 

Department of Electrical Engineering and Computer Science 

The University of Michigan, Ann Arbor, MI 48109-2122, USA 



^_ " Abstract 

A detailed study of monolithic InP-based HEMT oscillators for subterahertz 
operation is presented. InAlAs/InGaAs HEMT's have been optimized for high 
frequency operation and showed very high maximum oscillation frequencies (f maa . ) 
of 310 GHz using offset self-aligned T-gate technology. Power characteristics of 
HEMT oscillators are reported. An oscillation power of more than 10 mW was 
evaluated by large-signal analysis at 320 GHz using HEMT's with f max = 450 
GHz, V&r = 10 V and a gate width (W 3 ) of 8 x 22.5 /zm. Oscillator topology 
studies showed that complex feedback schemes such as dual and active feedback 
enhance the negative resistance. Push- push oscillator designs based on harmonic 
signal generation can finally be used to overcome the frequency barrier imposed by 
f 



1 Introduction 

InAlAs/InGaAs HEMT's have shown excellent high frequency characteristics and 
operation capability as discrete devices. A current gain cut-off frequency (fy) of 305 
GHz [1] and a maximum oscillation frequency of 455 GHz have been reported using 
heterostructures of this type [2]. These very encouraging discrete device results indicate 
that InP-based HEMT's can be used to realize monolithic circuits with operation 
frequency well into the millimeter-wave region. A number of such monolithic integrated 



"Work supported by NASA under contract NAGW-1334 



Third International Symposium on Space Terahertz Technology Page 59 

circuits have recently been demonstrated by the authors. These include monolithic 
HEMT mixers at 94 GHz showing conversion gain of 1 dB [3] and HEMT doublers 
at 180 GHz with a conversion loss of 6 dB [4]. Monolithic HEMT oscillators also have 
been realized by the authors up to W-band showing more than 1 mW power with devices 
having 36 jj,m gate periphery [5]. 

Another possibility opened to HEMT technology is its use for space-based remote 
sensing and radiometry, where fundamental sources are required to operate above 100 
GHz. A first detailed study concerning the power characteristics and the upper frequency 
limit of InAlAs/InGaAs HEMT's when used as oscillators, has recently been presented 
by the authors [6]. This paper provides further details on related issues of HEMT use for 
signal generation. It addresses first the ways of further optimization of InP-based HEMT 
technology in view of obtaining enhanced f mai performance (Section 2). Power and 
frequency characteristics of monolithic oscillators evaluated with the help of a large-signal 
analysis are presented in Section 3. Finally, specific designs and topologies of 160 GHz 
fundamental monolithic HEMT oscillators are discussed in Section 4. 

2 Device Optimization for High f max 

A very high i max of several hundred gigahertz is necessary to guarantee the device 
operation as oscillator at millimeter-wave frequencies. Optimization for high i max can 
be achieved by reducing the parasitic resistances and capacitances of the HEMT. The 
parasitic source resistance (R s ) consists of two parts : one coming from the contact region 
(R c ) and the other from the ungated region between the gate and source. In an attempt 
to minimize the ungated region resistance, a self-aligned gate technology has been applied 
to InAlAs/InGaAs HEMT's [7]. The ungated region between source and gate has been 
reduced in this case to less than 0.2 pun and the source access resistance was minimized, 
resulting in a very high extrinsic (j of 250 GHz. Although f max is directly proportional to 



Page 60 



Third International Symposium on Space Terahertz Technology 




Lgs=0 .2\lxa 
(constant) 



Lgd (variable) 



Figure 1: Schematic view of the self-aligned offset T gate HEMT 

fr, it was limited in this case due to the high output conductance (Gds) and gate-to-drain 
capacitance (C g d); this was caused by the proximity of the gate and drain making G,i s 
and Cgd higher than in HEMT's fabricated by conventional technology. 

A better insight to the problem can be obtained by examining the f max expression 
which is given by [8]: 



fn 



G d s 



R s + R„ 



4 Co 



= {^(On/fc + , * "o ) + 77^(1 + 2-5^)(l + G m R s f)-h 



Cgd 

c e 



It l G m v ' l/G m + R s 5 C gs vy gs 

It is obvious from Eq. (1) that a high f mai :/fr ratio can be achieved by increasing both 
C g s /Cgd and Gm/Gds- These two ratios are related to L 3 d/L 5i , where L gs is the distance 
between gate and source, and L g d is the distance between gate and drain. L g d/L gs can be 
increased by offsetting the gate instead of placing it at the center of the source-to-drain 
region. The novel self-aligned offset T-gate developed by the authors [9] and employed 
in the analysis presented here, allows one to satisfy these requirements. An additional 
feature of this approach is that L g d and L gs can be controlled much more accurately in 
this way than in processes where the gate has to be offset aligned between two ohmic 
contacts. Various L g j values ranging from 0.2 f.im to 0.6 /zm were employed while L gs 



Third International Symposium on Space Terahertz Technology 



Page 61 




Figure 2: SEM photograph of self-aligned offset T-gate (L g d = 0.6 /j.m) 



was fixed at 0.2 fim as shown in Fig. 1. 

The devices were fabricated following the self-aligned process described in [9]. The 
SEM photograph of the completed gate after the ohmic metal deposition is shown in 
Fig. 2. The highest f mar values were obtained with L g d = 0.4 /.im and the corresponding 
microwave results are shown in Fig. 3. fr ' s in this case around 150 GHz and f m(ir is 
greater than 300 GHz. By increasing L g d further, the value of f max /fr increases due to 
the higher C gs /C g d and G m /Gd s ratios. However, the magnitude of fy becomes smaller 
with L g d due to the increased gate length and source-to-drain spacing. The increase of 
fmai/fr ratio with L g d is thus compensated by the decrease of fr and the maximum l max 
occurs for L g d = 0.4 /j,m. 

Further f moI optimization is expected by reducing the gate length of the devices which 
had large offsets; due to the increased number of line scans for highly offset gales, the 
gate length becomes larger than in the case of the symmetric/centered realizations. This 



Page 62 



Third International Symposium on Space Terahertz Technology 



_ 350 

X 

S 300 h 



o 
C 

cr 



250 - 



200 - 



? 150 - 



3 
O 



100 



1 


1 1 1 

max >. 


I 








- 


\. f * 


- 


1 


I I I 





0.1 



0.2 0.3 0.4 0.5 
Lgd (|Lim) 



0.6 0.7 



Figure 3: Microwave data of offset self-aligned InAlAs/InGaAs HEMT's (h g d — 0.4 fxm). 
The results show an ij of 150 GHz and an f mar of 310 GHz, corresponding to a high 
imaxlh ratio of 2.0 

drawback can, however, be eliminated by a better optimization of the doses used for the 
footprint and side lobes of the gate. 

3 Evaluation of Oscillator Characteristics Using 
Large- Signal Analysis 

The design of high frequency oscillators is generally based on either small-signal S- 
parameters or measured large-signal S-parameters. The small signal S-parameters predict 
the initial conditions necessary for oscillation build up. However, the steady-state oscilla- 
tion condition can not be accurately predicted from small-signal S-parameters. Designs 
using measured large-signal S-parameters present also certain difficulties arising from 
measurement accuracy and differences between measured and simulated conditions. 



Third International Symposium on Space Terahertz Technology 



Page 63 



G(jw,A) 




V 



cir 



Ideal 
Coupler 



V 



ref 



7777 

Figure 4: The schematic circuit setup for the large-signal oscillator analysis 

An all frequency-domain large signal oscillator analysis method has been developed 
in view of evaluating the HEMT potential as oscillator. It employs small-signal 
S-parameters and a harmonic balance routine with 2-D interpolation functions. The 
method predicts the operation frequency, output power and optimum load termination 
conditions. 

A special circuit set-up is used to perform the large-signal oscillator analysis (Fig. 4). 
It consists of an amplifying unit and frequency-selective feedback loop together with an 
excitation signal. The HEMT is used in common source topology and is considered as an 
amplifying unit with power-dependent gain saturation characteristics. An ideal coupler 
is placed between the amplifying unit and feedback loop to initiate and monitor the 
oscillation (V ctr ). The excitation signal (V re /) is increased from a small signal level until 
the gain of the HEMT saturates and the steady-state oscillation condition is reached. 

The large-signal oscillator analysis method has been applied to study the oscillation 
power dependence on the termination impedance of common source InAlAs/InGaAs 
HEMT oscillators. The simulation results are shown in Fig. 5 for a HEMT with 2 



Page 64 



Third International Symposium on Space Terahertz Technology 







S 
n 

s 

u 

t 

o 

Oh 

o 

03 

i— i 

o 
O 



I I I 1 l" I I I | 

104 GHz 



i nrr"T"r'i i i 



121GHz 



199 GHz 





74 GHz 



-i i i i 1 1 



1 10 

Termination Impedance (Q) 



100 



Figure 5: The oscillation power dependence on the termination impedance at various 
frequencies for a HEMT (L 3 = 0.1 fim) with f max = 200 GHz and gate periphery of 2 x 
45 \im 

x 45 fim gate periphery and i max of 200 GHz. The analysis shows that an optimum 
output power level is obtained when the load impedance is of the order of 1/2 to 1/4 
of the small signal negative resistance at frequencies which are sufficiently away from 
fmox- At very high frequencies, the load impedance determined by the above criteria is 
reduced to very small values (below 5 Q) which are difficult to implement in monolithic 
form. This termination load requirement sets the practical limit of upper frequency at 
which the oscillator circuit can be implemented. These effects were studied and design 
criteria were established on the basis of practical realization constraints imposed by load 
terminations which should exceed 5 Q. 

The available power was evaluated at different frequencies using optimum termination 



Third International Symposium on Space Terahertz Technology 



Page 65 



10" 



I 

% 

£ 

a 
o 

i 



o 

CO 

O 



10' 



10 u 



10 



-1 



10 



-2 



G- 



f =450GHz \ 
tfp8X22.5um ^ 
Vbr=10V 



O- 



f =450GHz 

tfg=2X22.5um 

Vbr=10V 



<9« 




\ 



a 



f m =200GHz 

tfp2X45um 

Vbr=3V 



f mM =450GHz 

T*g^2X22.5um 

Vbr=3V 



■ ■ ■ i 



50 



500 



100 

R~equency(GHz) 

Figure 6: Power delivered by InAlAs/InGaAs HEMT oscillators as a function of 
frequency, gate periphery (W 5 ), maximum frequency of oscillation (f ma x) and breakdown 
voltage (Vbr) 

conditions. Three parameters are used for the simulation: gate periphery (W H ), 
maximum frequency of oscillation (fmax) and breakdown voltage (V(, r ). The results are 
shown in Fig. 6. The oscillation power decreases first slowly and shows a more dramatic 
degradation at high frequencies close to f max . This corresponds to the degradation of 
maximum available gain (G max ) and negative resistance (Rneg) at high frequencies. The 
reduced R ne g imposes a smaller value of termination load with the result of less power 
delivered to the load. The overall characteristics suggest that generation of reasonable 
power levels is feasible up to a frequency of 2/3 of f max . From Fig. 6, it is obvious that 
a higher f max ensures large oscillation power for a given frequency. More than 15 mW 
of output power can be expected up to 300 GHz out of 0.1 /zm HEMT's with eight 22.5 



Page 66 Third International Symposium on Space Terahertz Technology 

fim gate fingers assuming an i max of 450 GHz and V(, r of 10 V. This prediction doesn't 
include any parasitic effects coming from mismatches, losses of transmission lines and 
source grounding. 

Fig. 6 also shows the characteristics of devices with different gate widths in view of 
studying the effect of gate periphery on the output power level. Larger devices provide 
higher oscillation power, but they are harder to implement in oscillator circuits. This 
is due to their lower induced negative resistance, which implies the need for very small 
termination impedance. As a result, the oscillation power degrades fast at the high 
frequency end of operation bandwidth. A compromise has consequently to be made 
between the oscillation power and ease of realization when choosing the device periphery. 

Another important parameter in the oscillator evaluation is the breakdown voltage. 
A higher breakdown voltage allows one to bias the transistor at larger drain bias and 
thus to apply higher DC power to the device. The RF power generated from the 
device is proportional to the DC power and increases consequently with higher V(, r . 
Under conditions of optimum biasing for power, the maximum voltage swing is ~ 
V& r /2. By increasing the breakdown voltage of the HEMT's from 3 V to 10 V, it was 
found that the RF power increased by approximately 6 dB. Breakdown improvements 
in InAlAs/InGaAs HEMT's have recently been reported by Matloubian et al [10] and 
validate this assumption. Further work is, however, necessary to justify this possibility, 
especially at millimeter-wave frequencies. Similar improvements can also be made by 
increasing the current density of the device and may be even easier to achieve in 
InAlAs/InGaAs HEMT's with the help, for example, of multi-heterojunction designs 

[11]. 

It should finally be noted that the simulation results in Fig. 6 were obtained using 
a simple series feedback topology and the evaluated oscillation power values do not 
therefore necessarily reflect the maximum power capability of the devices. 

Overall, the HEMT's can be optimized for generation of adequate power levels at 



Third International Symposium on Space Terahertz Technology 



Page 67 



0.0 



£ -50.0 

JS 
Q, 

<u 
o 

c 

(0 

J_> 

.52-100.0 
w 

OS 

_> 

(0 
60 

4M50.0 



-200.0 



Single Series 

. Feedback ~^^ 


\ ./^Active ? 


c\ Feedback 

a 






Dual 




/ Feedback 
i 

i i i 



154 156 158 160 162 164 166 168 170 

frequency (GHz) 

Figure 7: Comparison of negative resistance for three different feedback schemes: 1) 
single series feedback, 2) dual feedback, 3) active feedback 

high frequencies with the help of: i) high f mox , ii) high V 6r , iii) high current density, iv) 
choice of appropriate gate periphery. 

4 Circuit Topologies for Subterahertz Monolithic 
HEMT Oscillators 

As already discussed earlier on, the negative resistance available at subterahertz 
frequencies is usually rather small. Furthermore, the induced negative resistance is 
present over a narrow frequency range especially when the oscillation frequency is close 
to the ( max of the device. The availability of small negative resistance values make the 
design task very difficult. Therefore, appropriate topologies have to be selected such that 
the negative resistance can be maximized over a wide frequency range. 



Page 68 



Third International Symposium on Space Terahertz Technology 



WW 



H 



Impedance 
Transformer 



O 



7777 

Figure 8: Equivalent circuit schematic of active feedback oscillator for subterahertz 
application 

Dual feedback schemes can be used to improve Rn eg over a narrow frequency range. In 
addition to the series feedback element from source to ground, a parallel feedback element 
can be inserted between the gate and drain. The negative resistance of this topology is 
compared with that of single series feedback topology in Fig. 7. As shown in this figure, 
dual feedback circuits provide the possibility of obtaining larger negative resistance. 
However, the negative resistance is present over a narrower range of frequencies than 
in the case of single series feedback circuits. 

The active feedback approach is another alternative for designing oscillators with 
devices of small R„ ep . It uses a small FET as a phase shifting element between the 
gate and drain (see Fig. 8). Since the feedback loop is provided by active rather than 
passive elements, the feedback phase shift is fairly independent of frequency and therefore 
oscillation is guaranteed over a wide range of frequencies. This can be verified from 
Fig.7. Furthermore, the active feedback approach is less sensitive to the parasitica coming 
from passive elements and interconnects because it does not strongly depend on passive 



Third International Symposium on Space Terahertz Technology 



Page 69 



* 



VgSQ- 



\ 



Impedance 
Transformer 



RFOut 

-o 



6 

Yds 



Figure 9: Equivalent circuit schematic of push-push feedback oscillator for subterahertz 
signal generation 

circuitry for inducing the oscillation. It is therefore very suitable for high frequency 
circuit applications, where accurate modeling of passive circuit elements is not really 
available. 

Another very interesting topology is the push-push configuration. The circuit 
schematic is shown in Fig. 9. It consists of two subcircuits combined in push-push 
arrangement. Each circuit oscillates at half the output frequency. The second harmonics 
are here combined in phase at the output terminal, while the first harmonics cancel 
each other. This configuration has the advantage of essentially doubling the operation 
frequency of the discrete devices. Thus, the circuit may operate beyond the frequency 
limit imposed by f max . This topology also, provides the possibility of lowering the 
phase noise because all the odd harmonics and associated noise are canceled at the 
output. Good balance between the two subcircuits has to be maintained for the successful 
operation of this circuit, but mutual interaction between the two devices is expected to 
make this requirement less stringent. Monolithic technology provides additional means 



Page 70 Third International Symposium on Space Terahertz Technology 

of achieving perfect balance. The monolithic push-push HEMT oscillator is therefore a 
very promising candidate for satisfying the needs for subterahertz signal generation using 
HEMT's. 

5 Conclusions 

The use of submicron InAlAs/InGaAs HEMT technology has been discussed in view 
of the possibility of realizing subterahertz oscillators. InP-based HEMT's have been 
optimized for this purpose and showed very high f max of 310 GHz using offset self-aligned 
T-gate technology. A gate-to-drain separation of 0.4 y.m was used in these devices. 

A large signal modeling method has been developed and applied to the evaluation of 
power characteristics of HEMT oscillators. The optimum termination loads have been 
found to be 1/4 - 1/2 of the small-signal negative resistance of the devices. Upper 
frequency limit criteria have been established and indicated the feasibility of signal 
generation up to ~ 2/3 of the f max of the device. The large signal analysis has also 
been used to evaluate the oscillation power of HEMT oscillators in the subterahertz 
region. The use of HEMT's with f max = 450 GHz, V 6r = 10 V and W p = 8 x 22.5 [im 
should allow an oscillation power of 15 mW at 320 GHz. 

The topology study of subterahertz oscillation has shown that enhanced negative 
resistance can be obtained by using complex feedback schemes such as the dual feedback 
and active feedback scheme. The frequency barrier imposed by f ma x can be overcome by 
harmonic oscillation operation as for example in the case of push-push oscillators. 

Acknowledgment 

The help of T. Brock, G. Munns and G. I. Ng in technology and material growth are 
greatly appreciated. 



Third International Symposium on Space Terahertz Techywlogy Page 71 



References 

[1] L. D. Nguyen, A. S. Brown, M. A. Thompson, L. M. Jelloian, L. E. Larson, 
and M. Matloubian, "650-A Self-Aligned-Gate Pseudomorphic Alo.4sIno.52As / 
Gao.20Ino.80As High Electron Mobility Transistors," IEEE Electron Dev. Lett., vol. 
13, no. 3, pp. 143-145, March 1992. 

[2] P. Ho, M. Y. Kao, P. C. Chao, K. H. G. Duh, J. M. Ballingall, S. T. Allen. 
A. J. Tessmer, and P. M. Smith, "Extremely High Gain 0.15 \im Gate-Length 
InAlAs/InGaAs/InP HEMTs," IEE Electron. Lett, vol. 27, pp. 325-327, 1990. 

[3] Y. Kwon, D. Pavlidis, P. Marsh, G. I. Ng, and T. Brock, "Experimental 
Characteristics and Performance Analysis of Monolithic InP-Based HEMT Mixers 
at W-Band," To appear in IEEE Trans. Microwave Theory Tech., 1992. 

[4] Y. Kwon, D. Pavlidis, P. Marsh, M. Tutt, G. I. Ng, and T. Brock, "180GHz 
InAlAs/InGaAs HEMT Monolithic Integrated Frequency Doubler," in Tech. Digest 
of 1991 IEEE GaAs IC Symposium, pp. 165-168, October 1991. 

[5] Y. Kwon and D. Pavlidis, "Large Signal Analysis and Experimental Characteristics 
of Monolithic InP-Based W-Band HEMT Oscillators," in Proceedings of the 21st 
European Microwave Conference, pp. 161-166, September 1991. 

[6] Y. Kwon, D. Pavlidis, and M. N. Tutt, "An Evaluation of HEMT Potential 

for Millimeter- Wave Signal Sources Using Interpolation and Harmonic Balance 

Techniques," IEEE Microwave and Guided Wave . Letters, vol. 1, pp. 365-367, 
December 1991. 



Page 72 Third International Symposium on Space Terahertz Technology 

[7] U. K. Mishra, A. S. Brown, L. M. Jelloian, M. Thompson, L. D. Nguyen, and S. E. 
Rosenbaum, "Novel High Performance Self-Aligned 0.15 Micron Long T-Gate," in 
Tech. Digest of 1989 International Electron Device Meeting, pp. 101-104, December 
1989. 

[8] M. B. Das, "A High Aspect Ratio Design Approach to Millimeter- Wave HEMT 
Structures," IEEE Trans, on Electron Devices, vol. ED-32, no. 1, pp. 11-17, January 
1985. 

[9] Y. Kwon, T. Brock, G. I. Ng, D. Pavlidis, G. 0. Munns, M. E. Sherwin, and G. I. 
Haddad, "F mai -Enhancement in CBE-Grown InAlAs/InGaAs HEMT's Using Novel 
Self-Aligned Offset-Gate Technology," in ^th Conf. on InP and Rel. Materials, April 
1992. 

[10] M. Matloubian, L. D. Nguyen, A. S. Brown, L. E. Larson, M. A. Melendes, and M. A. 
Thompson, "High Power and High Efficiency AlInAs/GalnAs on InP HEMTs," in 
1991 IEEE Int. Microwave Symp. Dig., pp. 721-724, June 1991. 

[11] G. I. Ng, D. Pavlidis, M. Tutt, J.-E. Oh, and P. K. Bhattacharya, "Improved Strained 
HEMT Characteristics Using Doouble-Heterojunction Ino.65Gao.35 As / Ino.52Alo.4sAs 
Design," IEEE Electron Dev. Lett., vol. 10, no. 3, pp. 114-116, March 19S9. 



Third International Symposium on Space Terahertz Technology Page 73 

VARACTOR DIODES FOR MILLIMETER AND SUBMILLIMETER WAVELENGTHS 

Brian J. Rizzi, Jeffrey L. Hesler, Hasan Dossal and Thomas W. Crowe 

Semiconductor Device Laboratory J> y-*2A 

Department of Electrical Engineering N 9 3 * 2« «S 3 

Thornton Hall /%06'Zr 

University of Virginia 
Charlottesville, VA 22903-2442 £ *D^> 

ABSTRACT 

Whisker-contacted GaAs Schottky barrier varactor diodes are the most common high- 
frequency multiplier element in use today. They are inherently simple devices that have very 
high frequency response and have been used to supply local oscillator power for Schottky 
heterodyne receivers to frequencies approaching 700 GHz. This paper discusses the 
development of improved varactor diode technology for space based applications at 
millimeter and submillimeter wavelengths. 

I. INTRODUCTION 

Whisker contacted GaAs Schottky varactor diodes are presently in use to supply local 
oscillator power at frequencies as high as 700 GHz for ground based and airborne 
applications [1,2,3]. These diodes are also used in the Microwave Limb Sounder on NASA's 
Upper Atmosphere Research Satellite which is now monitoring global ozone depletion [4]. 
Although these devices have proven to be quite useful, there is great interest in developing 
technologies that are more mechanically robust, have higher operating frequency and have 
the potential to generate greater amounts of power. This paper will review recent work at the 
University of Virginia on multiplier elements. This includes both the development of planar 
Schottky varactors and investigation of new devices that have the potential for improved 
performance. 



Page 74 Third International Symposium on Space Terahertz Technology 

Section II will review our first attempt to fabricate a planar varactor diode for use at 
millimeter wavelengths. The preliminary design is presented and the limitations of this 
structure are considered. A next generation device is then proposed. It is hoped that this new 
device will become a standard replacement for a very successful and commonly used 
whisker-contacted varactor diode (U. Va.-6P4). Through development of this device we hope 
to demonstrate the potential of planar varactor technology and investigate the factors that 
will most seriously degrade planar varactor performance at high frequency. 

We are developing varactor diodes for a multiplier chain to 1 THz. This system will 
incorporate two doublers (80 to 160 GHz and 160 to 320 GHz) and a tripler (320-960 GHz). 
The doublers will use multiple diodes integrated on a single chip to enhance power handling 
ability. These chips are designed to be used in a balanced doubler developed by Erickson 
[5]. The prototype doubler design and some preliminary results are presented in section m. 

The tripler to 1 THz will be extremely challenging. Fortunately there is a great deal of 
effort being expended world-wide on new varactor structures which may be useful for this 
work. In section IV we will consider a variety of possible technologies, with special 
emphasis on an integrated 5-doped varactor diode pair and consideration of the new 
heterojunction barrier varactors. Section V is a brief summary of this work. 

n. Development of a Planar Varactor technology 

As a first step in the development of planar varactor diodes we will fabricate planar 
devices to replace two commonly used whiskered diodes. These are the 6P4 diode, which is 
commonly used for doubling in the millimeter wavelength range, and the 2T2, which is used 
to double and triple at submillimeter wavelengths. The parameters of both of these devices 
are listed in Table I. The primary electrical benefits of the whiskered diode technology is the 



Third International Symposium on Space Terahertz Technology 



Page 75 



low shunt capacitance of the whisker and the ability of the diode user to tune the whisker 
inductance to optimize performance. 

A scanning electron micrograph of a prototype planar varactor is shown in Fig. la. The 
surface channel fabrication procedure has been described previously [6,7]. The nominal 
parameters for this diode, designated SC6T1, are also listed in Table I. This diode was 
designed as a replacement for the 6P4 diode, however, there are two problems. First, series 
resistance is substantially higher than the 6P4's and second, the planar diode has a parasitic 
shunt capacitance of 12 fF which is unacceptably high. As might be expected, preliminary 
RF measurements have been disappointing. The excess series resistance is due to the use of 



Table I: Schottky Varactor Diodes 



Batch Type 



Epitaxial Epitaxial Zero-bias Minimum 

Anode Layer Layer Series Junction Junction Breakdown 

Diameter Thickness Doping Resistance Capacitance Capacitance Voltage 
(urn) (urn) (cm" 3 ) (Q) (fF) (fF) (V) 



6P4 Whiskered 6 
2T2 Whiskered 2.5 
SC6T1 Planar 6.2 



1.0 


3xl0 16 


9.5 


20 


5.5 


20 


0.59 


lxlO 17 


12 


5.5 


1.5-2.0 


11 


1.3 


2xl0 16 


20 


20 


4 


30 




S.I. CaAs 



60 urn 



30 



50-150 un 



SX GaAs 



Fig. 1. a) A prototype planar varactor diode. The surface channel technology is used to 
achieve isolation between the contact pads [6,7]. b) A sketch of the second 
generation device which has smaller contact pads and variable finger length. 



Page 76 Third International Symposium on Space Terahertz Technology 

an epitaxial layer that is too thick and too lightly doped. Although this epitaxial layer yields a 
higher breakdown voltage, the penalty in R s outweighs this benefit. Since the fabrication of 
the SC6T1, new material has been obtained and diodes with characteristics closer to those of 
the 6P4 will be fabricated. 

The increased shunt capacitance of the planar diode is a serious problem. This 
capacitance is due primarily to the fringing field between the contact pads through the high 
dielectric constant GaAs substrate. To reduce this capacitance there are three options: 

1) Reduce the pad dimensions, 

2) Increase the pad separation (and therefore the finger length), and/or 

3) Use a substrate with a lower dielectric constant 

The first two improvements will be implemented in our next generation device, as shown in 
Fig. lb. The primary limitations on pad dimensions are the ohmic contact resistance and the 
need to make a reliable solder contact. Although the proposed pad width of 30 |im is about 
the minimum size that most users feel comfortable soldering to, it is clear that if smaller pads 
will lead to better performance, users will develop more elaborate soldering techniques. 
However, the minimum pad size is also limited by our ohmic contacts. We use SnNi/Ni/Au 
plating for our standard ohmic contact and reliably obtain resistivities of 10 -5 Qcm 2 or 
slightly less. Thus, a 30 nm x 30p.m pad should have roughly one ohm of contact resistance. 
Smaller pads will require a significantly improved ohmic contact technology. 

The new mask set will have several finger lengths, from 50 - 150 ^im. This will allow 
evaluation of RF performance as a function of pad-to-pad capacitance and finger inductance. 
It is expected that one specific finger length will give optimum performance in a given 



Third International Symposium on Space Terahertz Technology Page 77 

multiplier mount at a given frequency. Thus, we expect that detailed RF evaluation of these 
devices will yield important guidelines for future chip designs. 

The use of a quartz substrate for planar Schottky diodes has been demonstrated for 
mixer applications [7]. This has led to significant reductions in shunt capacitance which may 
be important for multiplier applications. However, the thermal properties of the GaAs 
Schottky diode on quartz are not well understood, and we have noticed that mixer diodes on 
quartz substrates are more likely to show signs of heating effects than equivalent diodes on 
GaAs. Since the removal of heat from the varactor diode is particularly important, it is not 
clear if quartz substrates will yield an overall performance benefit. Our next batches of 
planar varactors will have GaAs substrates. However, we also hope to investigate quartz and 
perhaps sapphire substrates in the near future. 

III. INTEGRATED SCHOTTKY VARACTORS FOR BALANCED DOUBLING 

The first step in the proposed multiplier chain to 1 THz is a doubler from 80 to 160 
GHz. Since there are sources available that can deliver large amounts of power at 80 GHz, 
our goal is to develop a doubler that is fairly efficient, but, more importantly, can handle 
large input powers. With this goal in mind, a planar chip was designed based on the balanced 
doubler configuration of Erickson, which has generated up to 25mW at 160 GHz using two 
whisker contacted diodes [5]. 

One benefit of the planar diode technology is the ability to integrate several diodes on a 
chip to increase power handling ability. For example, when two diodes are placed in series 
their individual areas can be doubled in order to maintain the same total series resistance and 
junction capacitance as a single device. However, the series pair will have twice as much 
reverse breakdown voltage. The increased area and breakdown voltage will yield improved 



Page 78 



Third International Symposium on Space Terahertz Technology 



power handling ability. Two scanning electron micrographs of our prototype are shown in 
Fig. 2. The chip consists of four varactor diodes, two for each leg of the balanced doubler. 

The design parameters and dc characteristics of the prototype balanced doubler chips 
are shown in Table II. Our goal was to achieve a reverse breakdown voltage of 20V for each 
anode. Also, the anode diameters of 10 and 12 nm were chosen to achieve zero-bias junction 
capacitances of 40 and 60 fF per anode. As is seen in the table, the first batch had excessive 
series resistance and extra breakdown voltage. This is due to the low doping density and 
thickness of the epitaxial layer. For the second batch this problem was corrected at the cost 
of reduced breakdown voltage. However, this trade-off is expected to yield significantly 
improved RF performance. 

The capacitance-voltage (C-V) curves for a single diode and a diode series pair are 
shown in Fig. 3, indicating the increased breakdown voltage of the diode pair. 

Preliminary RF tests for the first prototype balanced doubler were performed by Dr. 
Erickson at the University of Massachusetts and the results are presented in Table II. These 
initial results are quite encouraging, but not yet competitive with the whiskered-diode 





Fig. 2. SEM photographs of the prototype balanced doubler to 160 GHz. 



Third International Symposium on Space Terahertz Technology 



Page 79 





Table II: 


Prototype Balanced Doubler Chips 


DC Characteristics 


Batch 

# 


Epitaxial 
Layer 

Thickness 
(um) 


Epitaxial 

Layer 

Doping 

(cm" 3 ) 


Anode 

Diameter 

(urn) 


Pair 

Series 

Resistance 

(«) 


Pair 

Breakdown 

Voltage 

(V) 


1 

2 


1.3 
1.2 


1.8xl0 16 
2.5xl0 16 


10 
10 


20 
14 


45 
35 


Preliminary RF Data 1, 


Batch 

# 


Input 
Freq. 
(GHz) 


Output 
Freq. 
(GHz) 


Input 

Power 

(mW) 


Output 
Power 
(mW) 


Efficiency 
(%) 


1 
1 


82 
82 


164 
164 


55 
100 


3 
6 


6 
6 


f Preliminary RF data supplied by N. Erickson, University of Massachusetts. 
Measurements have not been performed with batch #2. 



40- 



30- 



Cd(fF) 



20- 



10- 



f. 

r. 








~i r 

10 20 

Reverse Voltage 



I 
30 



Fig. 3. C-V Characteristics of the prototype balanced doubler chip for a single varactor 
diode (dotted), a diode series pair (solid) and for a single diode with anode-to- 
pad connection to eliminate the pad-to-pad shunt capacitance (dashed). 



page 80 Third International Symposium on Space Terahertz Technology 

results. Two changes in the chip design are planned to improve performance. The first is the 
increase in epitaxial layer doping to reduce series resistance, as was achieved with batch #2. 
The second is the reduction of pad-to-pad capacitance. The importance of this is 
demonstrated by the third curve (dashed) in Fig. 3. This curve was measured from the anode 
to ohmic contact pad on a diode that had no ringer, and therefore does not include the pad- 
to-pad capacitance. This curve has much greater modulation and much lower minimum 
capacitance. This clearly demonstrates that the pad-to-pad capacitance is having a major 
effect on performance. The shunt capacitance of future chips will be reduced through a 
redesign of the contact pads and possibly through the use of quartz substrates. 

Once the first stage multiplier has been optimized, the next step is to design a chip for 
the doubler to 320 GHz. Since the second stage will not have to handle as much power as the 
first, we will be able to trade-off some power handling ability in order to increase cut-off 
frequency. It is expected that the optimum diodes for this stage will have smaller anodes and 
higher epitaxial layer doping density. 

There is much work to be done on the integrated balanced doublers. However, the 
prototype devices have yielded encouraging results, and the improvements necessary to 
increase performance are clearly defined. Thus, we expect to achieve significantly improved 
output powers at 160 GHz in the near future. Also, the lessons learned on the first stage 
doubler will be applied to the second stage, so that development of the higher frequency 
chips should progress more rapidly. 



Third International Symposium on Space Terahertz Technology Page 81 

IV. POTENTIAL VAR ACTORS FOR TRIPLING TO 1 THZ 

The development of a tripler to 1 THz is an extremely challenging task. Fortunately 
there are several device technologies that may yield suitable performance. We have chosen 
to investigate five of these, each of which is discussed in the following sections. The 
whiskered Schottky and planar Schottky are considered briefly and the new two- 
dimensional-electron-gas/Schottky (2-DEG/Schottky), which is considered in detail in a 
separate paper, is also only briefly overviewed. The other two technologies, the integrated 
5-doped varactor pair and the heterojunction barrier varactor, are considered in more detail. 

A. Whiskered Schottky Diodes 

The most likely candidate for the first successful tripler to 1 THz is simply a standard 
whisker contacted Schottky varactor. The 2T2 diode has already been successfully used in 
triplers to 500-700 GHz and can probably be extended to the THz range. However, the 
efficiency will certainly be decreased and it is not clear how much output power will be 
achieved. A more optimized diode can probably be developed, perhaps with slightly higher 
doping density and smaller diameter. Although this technology appears to be reaching 
fundamental limitations [8], it should continue to be pursued because the probability of some 
level of success is high. 

B. Planar Schottky Varactors 

There are two advantages of using a planar Schottky device; the elimination of the 
fragile whisker contact and the opportunity to use several integrated diodes to increase power 
handling ability or achieve a more beneficial C-V characteristic. The drawback is the 
increased shunt capacitance that is inherent in the planar diode. There are several areas that 
must be researched. As discussed previously, these include the redesign of the contact pads 



C-X 



Page 82 Third International Symposium on Space Terahertz Technology 

and anode finger, and the use of low dielectric constant substrates. Also, the potential use of 
two Schottky varactors in an anti-series combination to achieve a symmetric C-V 
characteristic may have substantial benefits for tripling applications. It is not yet clear if 
planar Schottky technology will be useful at 1 THz, however we hope to answer many 
important questions through our development of planar diodes for lower frequencies. 

C. The 2-DEG/Schottky Diode 

This device consists of a metal contact to the edge of a two-dimensional-electron-gas 
(2-DEG) formed at a heterointerface. The capacitance is between the Schottky metal and the 
undepleted portion of the 2-DEG. The voltage on the Schottky metal modulates the depletion 
depth in the 2-DEG, thereby varying the capacitance. This device should benefit from 
increased electron mobility and perhaps higher electron saturation velocities compared to 
bulk devices. This may lead to significantly improved high frequency performance. Also, 
this is an inherently planar device. Prototype diodes have demonstrated excellent capacitance 
modulation and high reverse breakdown voltages. This new device is discussed in greater 
detail in a separate paper [9]. 

D. An Integrated S-Doped Diode Pair 

A design for a planar chip with two integrated 8-doped varactor diodes in a back-to- 
back configuration is shown in Fig. 4. The symmetric C-V characteristic of such a diode pair 
will yield significant benefits for tripler applications since an idler circuit at the second 
harmonic is not needed. The 8-doped diodes have been shown to have a sharp C-V 
characteristic [10,1 1]; which is a significant advantage at high frequencies since the available 
input power is quite low. 



Third International Symposium on Space Terahertz Technology 



Page 83 



The planar tripler has been designed to produce a capacitance ratio (C max /Cnun) of 2.5, 
with an estimated cut-off frequency of 6 THz. The material structure is described in Table 
HI. The mask set and epitaxial material for this device are now being purchased. 

E. Evaluation of the Heterostructure Barrier Varactors 

In 1990 Rydberg et al. demonstrated that a thin layer of high band-gap material 
sandwiched between two thicker layers of low band-gap material could yield a symmetric 
C-V characteristic that is ideal for tripler applications [12]. This Quantum (or 
Heterostructure) Barrier Varactor (QBV or HBV) has promise for high frequency multiplier 
applications, and is now being investigated by several groups. The goal of our investigation 





60 un 


50-150 un 








S.I. GaAs 








I 


- 1 






30 
un 


/ 


"^ 








P 






J 






S 








^. 


r 


\ 










j 






a 










S.I. GaAs 






k 



Fig. 4. A sketch of the proposed integrated 6-doped varactor pair. The finger length will 
be variable on the mask set and the anode spacing has not yet been determined 



Table III: Epitaxial Material for the 8-doped Diode Pair 


Layer 


Doping Thickness 


Type 


Density (um) 


NGaAs 


<lxl0 15 cm -3 0.05 


Si atomic layer 


3.4xl0 12 cm -2 


NGaAs 


2xl0 17 cm" 3 0.13 


N+GaAs 


>3xl0 18 cm -3 3.0 


Al x Gai_ x As etch stop 


undoped(x>0.5) 2.0-2.5 


GaAs substrate 


S.I. 



page 84 Third International Symposium on Space Terahertz Technology 

is to determine if HBVs offer significant improvement over standard Schottky technologies, 
and, if so, to demonstrate such improvement. To determine the potential of these devices we 
will discuss the design of HBVs that have characteristics similar to the state-of-the-art 
whiskered varactors whose characteristics were presented in Table I. 

A schematic band diagram of a zero-biased single barrier GaAs/AlGaAs/GaAs HBV is 
shown in Fig. 5a. When a voltage is applied to the device a depletion region is created on 
one side of the barrier which increases in length as the voltage is increased. The capacitance 
of this device is approximated as 

C = * (1) 

Xb/eb + Xo.total/eM 

where A is the device area, e is the permittivity of the barrier (B) and the modulation region 
(M) materials, Xb is the barrier layer thickness and Xo.totai is the total depletion layer 
thickness on both sides of the barrier as a function of voltage. The maximum capacitance 
can be as high as EbA/Xb if there is negligible depletion in the modulation layers at zero- 
bias. The series resistance for a single barrier HBV, including spreading, epilayer and ohmic 
contact resistance, is estimated as 

j 2Xjvi Re 

Rl.s = Rl.spr + Rl.epi + Rl.ec = ^T + ^a" + X (2) 

where d is the anode diameter, o is the conductivity of the substrate (S) and epilayer (E) 
materials, Xm is the length of the n-type modulation regions, and Re is the specific resistivity 
of the ohmic contact. It is important to note that the device area affects not only the junction 
capacitance, but also the resistance of the ohmic contact Therefore, although we can reduce 
the junction capacitance by shrinking the device area, this is not beneficial unless the ohmic 
contact resistivity, R<., is low enough so that the third term in (2) remains negligible. For this 



Third International Symposium on Space Terahertz Technology 



Page 85 



study we will assume a specific contact resistivity of 10 Qcm , which is consistent with 
the best contacts reported in the literature. 

An important parameter for all varactor diodes is the voltage at which the conduction 
current becomes significant. For a standard Schottky varactor, impact ionization in the 
depletion region determines the reverse breakdown voltage and thermionic emission over the 
Schottky barrier determines the forward conduction current For the HBV either thermionic 
emission or avalanche breakdown can play the critical role, depending on the device 
parameters. Figure 5b shows an HBV band diagram with voltage applied, with a depletion 
region on one side of the barrier and an accumulation region on the other. As the HBV is 
biased, the accumulation region grows and therefore the effective barrier height, given by 
AEg-qVacc, decreases. Simultaneously, the electric field strength in the depletion region 
grows. Whether avalanche breakdown or thermionic current occurs first depends primarily 
on the conduction band discontinuity AE C and the band-gap in the modulation region. 

We would like to have a method to compare HBV diodes to standard varactors. A 
simple computer model was developed in order to simulate the operation of the HBV under 
applied bias. The simulation assumes that little current flows through the device, and 



1 








-►; 


A. - - 

AE _'-qv 

C ^ ace 

1 
T 




A 

qv bar 

i 


L 


«-x B -+ 


<- X M~ 


AE c 


.T._ „ ^ 


t 


1 


' 




^ E c 


/■ 








" X 


1 


///// 


AIGaAs n-GaAs 


///// 

n ++ -GaAs 


_ _.T..-.~. 


n^-GaAs n-GaAs 


1 

T... 


D ^-\^ 



Fig. 5. The band-diagram of a simple Heterostructure Barrier Varactor (HBV), a) zero- 
bias and b) bias applied. 



Page 86 Third International Symposium on Space Terahertz Technology 

calculates the quasi-equilibrium band diagram for different bias levels. The approximations 
developed by Delagebeaudeuf et al. [13] are a relatively standard method to analyze a 2- 
DEG at a heterostructure interface. For our case, these approximations were extended to 
include additional energy levels due to the high doping density in the 2-DEG region. The 
most crucial parameter to estimate is the maximum voltage that can be applied before 
conduction current begins to degrade the multiplier efficiency. For this discussion we will 
assume that impact ionization becomes important at the voltage where the electric field 
strength exceeds a critical value. Since there is no experimental data from which to estimate 
the critical field of HBVs, we have assumed that the critical field will be similar to that of 
GaAs pn junctions [14]. For devices dominated by thermionic emission, the maximum 
voltage was assumed to be that voltage at which 

AE C - qV^ = nkT, (3) 

and we have assumed a value of n=5 for this study. 

For GaAs/AlGaAs devices the value of AE C is rather small (AE C =0.35 [15]) and 
thermionic emission becomes important before impact ionization. Simulations show that for 
low doping levels (less than about 10 16 cm" 3 ) these devices can have a maximum voltage 
comparable to a standard Schottky varactor, however, Repj will be extremely large. 
Increasing the doping level decreases Repi, but also decreases V max , which indicates that 
single barrier GaAs/AlGaAs HBVs will have less power handling ability than standard 
Schottky varactors. There are several possible solutions to this problem, two of which will be 
considered in this paper; epitaxial stacking of barriers and the use of different material 
systems. Epitaxial stacking divides the applied voltage among several barriers, thus 
increasing the maximum device voltage. For an HBV with N barrieTS, VN max = NV lmax . 



Third International Symposium on Space Terahertz Technology Page 87 

As N is increased, it is best to increase the device area in order to maintain reasonable values 
of junction capacitance and modulation layer resistance. Assuming that the area is increased 
proportionally to the number of barriers, the series resistance of an N barrier HBV can be 
expressed as, 

R N.s=^^ R l,spr + 2N Rl.epi + ^ R l,oc- (4) 

This equation shows that the increase in area has the important effect of reducing the 
spreading and ohmic contact resistances. In fact, without using multiple barriers it would be 
impossible to fabricate an HBV with reasonable capacitance and series resistance unless the 
ohmic contact resistivity is exceptionally low. 

Other material systems can have significantly higher values of AE C . For example, the 
InGaAs/InAlAs system can yield barriers of near 0.8 eV, while the GaAs/GaN system allows 
0.9 eV barriers. The computer simulation indicates that the maximum voltage in both of 
these material systems is limited by impact ionization, rather than thermionic emission. 

In the following paragraphs, the simulation results for single and multiple barrier HBVs 
are discussed for the previously mentioned material systems. In order to compare the HBVs 
with the 2T2 and 6P4 varactors, barriers are added until V max is greater than that of the 
standard varactor. The area is then chosen so that C^ of the HBV is the same as the 
standard varactor. A common varactor figure-of-merit used in our comparisons and listed in 
Table IV is the dynamic cut-off frequency, given by [16] 

tco " 2tcR s ' (5) 

where Cma* and Cm^ are the maximum and minimum device capacitance. 



Page 88 



Third International Symposium on Space Terahertz Technology 



GaAs/AlGaAs: Figure 6 shows the simulation results for GaAs/AlGaAs HBV's. As the 
modulation doping density is increased, more barriers are needed to achieve the desired 
maximum voltage and the series resistance is reduced due to the increase in both o"e and 
device area. Table IV gives examples of GaAs/AlGaAs HBV's with V max , 0^ and R s 
similar to the 2T2 and 6P4. However, the HBVs will have the added benefit of a symmetric 
C-V curve. 

InGaAs/InAlAs: With the InGaAs matched to InP (i.e. 53% In), Ino.32Alo.68 As will give a 
AE C of about 0.8 eV with a 1% lattice mismatch. However, InGaAs has a narrower band gap 
than GaAs, and will thus have a smaller critical field for impact ionization. In these 
simulations, we used the critical field data versus doping for a Ge abrupt p-n junction [14] 
since Ge and Ino.53Gao.47 As have similar bandgaps. Because of the lower critical field, 
single barrier InGaAs/InAlAs HBV's will not have sufficient V max , and thus multiple 



Table IV: Heterostructure Barrier Varactors 


Material System 


N 


X B 


N mo d 


Diam. 


"max 


R, 


*-niin 


r 

^-max 


tco 






(urn) 


(cm" 3 ) 


(um) 


(V) 


(") 


(fF) 


(fF) 


(THz) 


GaAs/AlGaAs 


1 


0.02 


5xl0 16 


2.3 


4.0 


37 


1.5 


22 


2.6 




1 


0.02 


lxlO 17 


1.7 


2.6 


21 


1.5 


12 


4.4 




1 


0.02 


5xI0 17 


1.0 


1.2 


19 


1.5 


4 


3.4 


(6P4-like) 


5 


0.02 


5xl0 16 


9.0 


20 


7 


4.5 


66 


4.6 


(2T2-like) 


5 


0.02 


lxlO 17 


3.8 


13.2 


11 


1.5 


12 


8.6 


InGaAs/InAlAs 


1 


0.02 


5xl0 16 


2.6 


6.7 


25 


1.5 


32.3 


4.0 




1 


0.02 


lxlO 17 


1.9 


4.3 


16 


1.5 


17.2 


6.0 




1 


0.02 


5xl0 17 


1.2 


2.8 


16 


1.5 


6.4 


5.1 


(6P4-like) 


3 


0.02 


5xl0 16 


8.0 


20.0 


5.5 


4.5 


97 


6.0 


(2T2-like) 


3 


0.02 


lxlO 17 


3.4 


12.9 


9.5 


1.5 


17.2 


10.4 


GaAs/GaN 


1 


0.008 


5xl0 16 


6.0 


19.2 


13 


4.5 


367 


2.7 




1 


0.008 


lxlO 17 


2.5 


11.4 


20.5 


1.5 


63 


5.1 


(6P4-like) 


2 


0.008 


lxlO 17 


6.1 


23 


5 


4.5 


190 


7 


(2T2-like) 


2 


0.008 


2.3xl0 17 


2.8 


18.2 


9.5 


1.5 


38.6 


10.7 



Third International Symposium on Space Terahertz Technology Page 89 

barriers must be used. Figure 7 and Table IV show that devices similar to the 2T2 and 6P4 
varactors can be achieved with three barriers. 

GaAs/GaN: The GaAs/GaN material system has a high AE C and the critical field of GaAs. 
The major disadvantage is that it is a relatively new material system on which little 
experimentation has been performed [17]. Our simulations showed that impact ionization 
will be the limiting factor for these devices. Single barrier GaAs/GaN HBV's have sufficient 
V max , but tend to have higher R s than comparable Schottky varactors due to modulation 
region resistance. Characteristics of single barrier GaAs/GaN HBV's are given in Table IV 
for several dopings. By using higher N mo< j and multiple barriers, HBV's with low R s and 
very little conduction current should be possible. Figure 8 and Table IV show that only two 
barriers are required to achieve device characteristics similar to the 2T2 and 6P4. 

V. Summary 

Whisker-contacted GaAs Schottky barrier varactor diodes are the best multiplier 
elements available for millimeter and submillimeter wavelength applications. However, the 
development of planar diode technology and new devices promise to improve both system 
reliability and performance. Our prototype planar Schottky varactors are not yet competitive 
at millimeter wavelengths, but several straight-forward improvements in the chip design 
should alleviate the problems of high series resistance and shunt capacitance. The ability to 
integrate several varactor diodes onto a chip is being exploited to increase power handling 
ability, and an integrated balanced doubler for millimeter wavelengths has been described. 
The prototype devices have shown promising performance at 160 GHz and the second 
generation chips have greatly improved dc characteristics. Both the single-diode planar 
Schottky varactor and the balanced doubler will benefit from improved contact-pad/finger 



Page 90 



Third International Symposium on Space Terahertz Technology 



geometries and lower ohmic contact resistances. The use of low dielectric substrates is also 
being investigated. 

Several device technologies may be useful for a proposed tripler to 1 THz. While the 
whisker-contacted Schottky diode is likely to be the first device to yield reasonable output 
power at this frequency, planar diodes and other device structures promise improved 
performance. The 2-DEG Schottky, 8-doped Schottky and the quantum (heterostructure) 
barrier varactors (QBV or HBV) are being investigated at U.Va. The 2-DEG Schottky 
research is described elsewhere [9]. An integrated 5-doped varactor pair with symmetric C-V 
was described and will be fabricated in the near future. The HBV devices are particularly 
promising. Our simple analysis has shown that HBVs with parameters similar to the state- 
of-the-art Schottky varactors can be designed if multiple barriers are used and ohmic contact 
resistances are in the 10 _7 £2cm 2 range. These devices will also have the benefit of a 
symmetric C-V curve. The use of InGaAs/InAlAs or GaAs/GaN for the HBVs promises the 
best performance if epitaxial layers of the required quality can be obtained. 




5el6 



le17 




5el6 



1 el 7 



2.3el7 5el7 

N mod( OT - 3 ) 



2.3e17 5e17 

N modK4) 

Fig. 6 The number of barriers necessary to achieve a GaAslAlGaAs HBV with the same 
maximum voltage as the 2T2 (left) and 6P4 (right) varactors as a function ofN mod . 
Also, shown is the series resistance when the area is chosen to yield the same 
minimum device capacitance as the standard varactors. Additional parameters are 
listed in Table TV. 



Third International Symposium on Space Terahertz Technology 



Page 91 




Se16 



Iel7 



2.3el7 



N mod (a" -3 ) 



5el7 




2.3elS 



5el6 



Iel7 ■ 2.3e!7 

N mod ( cm_3 ) 



Fig. 7. The same as Fig. 6, but for InGaAslInAlAs. 




5e17 1e18 

N mod ( cm_3 ) 




mod 



(cm-3) 



Fig. 8. The same as Fig. 6, but for GaAslGaN. 



Page 92 Third International Symposium on Space Terahertz Technology 



ACKNOWLEDGEMENT 

The authors would like to thank Dr. Neal Erickson (U. Mass, Amherst) for supplying 
the initial RF measurements on the prototype balanced doubler and Dr. Peter Siegel (Jet 
Propulsion Laboratory) for initial evaluation of the SC6T1 planar varactor. This work has 
been supported by the National Science Foundation under Grant ECS-8720850, NASA and 
the Jet Propulsion Laboratory. 

REFERENCES 

[I] R. Zimmermann, R. Zimmermann, and P. Zimmermann, "All Solid-State Radiometers for Environmental 
Studies to 700 GHz," Third Int'l Symp. Space THz Tech., Ann Arbor, MI, March 1992. 

[2] H. Nett, S. Crewell, K. Kunzi, "A 625-650 GHz Heterodyne Receiver for Airborne Operation," 16th Int'l 
Conf. IR and MM Waves, Lausanne, Switzerland, August 1991. 

[3] S. Crewell and H. Nett, "Measurements of the Single Sideband Suppression for a 650 GHz Heterodyne 
Receiver," Third Int'l Symp. Space THz Tech., Ann Arbor, MI, March 1992. 

[4] A.L. Riley, UARS Microwave Limb Sounder Instrument Description, Jet Propulsion Laboratory 
Document D-1050, 1984. 

[5] Nil. Erickson, "High Efficiency Submillimeter Frequency Multipliers," 1990 IEEE MTT-S Int'l. 
Microwave Symp., Dallas, TX, May 1990. 

[6] Wl. Bishop, K. McKinney, RJ. Mattauch, T.W. Crowe, and G. Green, "A Novel Whiskerless Schottky 
Diode for Millimeter and Submillimeter Wave Applications," Proc. 1987 IEEE MTT-S Int'l. Symp., Las 
Vegas, NV, pp. 607-610, June 1987. 

[7] W.L. Bishop, E.R. Meiberg, RJ. Mattauch and T.W. Crowe, "A Micron Thickness, Planar Schottky 
Barrier Diode Chip for Terahertz Applications with Theoretical Minimum Parasitic Capacitance," 1990 
IEEE MTT-S InL Microwave Symp., Dallas, TX, May 1990. 

[8] T.W. Crowe, W.C.B. Peatman and E.M. Winkler, "GaAs Schottky Barrier Varactor Diodes for 
Submillimeter Wavelength Power Generation," Microwave and Optical Technology Letters, Special Issue 
on Space THz Technology, Vol. 3, No. 1, pp. 49-53, Jan. 1991. 

[9] W.C.B. Peatman, T.W. Crowe, and M. Shur, "A 2-DEG Varactor Diode for Millimeter and Submillimeter 
Wave Multiplier Applications," Third Int'l Symp. Space THz Tech., Ann Arbor, MI, March 1992. 

[10] T.W. Crowe, W.C.B. Peatman and W.L. Bishop, "GaAs Schottky Barrier Diodes for Space Based 
Applications at Submillimeter Wavelengths," The First International Symposium Space Terahertz 
Technology Proceedings, pp. 256-272, Ann Arbor, Michigan, March 1990. 

[II] B.J. Rizzi, T.W. Crowe, and W.C.B. Peatman, "A 5-Doped Varactor Diode for Submillimeter 
Wavelengths," The Digest of the 15th International Conference on Infrared and Millimeter Waves, pp. 
478-480, Orlando, Dec. 1990. 

[12] A. Rydberg, H. Gronqvist and E. Kollberg, "Millimeter- and Submillimeter- Wave Multipliers Using 
Quantum-Barrier- Varactor (QBV) Diodes," IEEE Electron Device Letters, Vol. 11, No. 9, pp. 373-375, 
1990. 

[13] D. Delagebeaudeuf and N.T. Linn, "Metal-(n)AlGaAs-GaAs Two-Dimensional Electron Gas FET," IEEE 
Trans. Electron Devices, Vol. ED-29, No. 6, pp. 955-960, 1982. 

[14] S.M. Sze, Physics of Semiconductor Devices, Wiley, New York, p. 103, 1981. 

[15] J. Batey and S.L. Wright, "Energy band alignment in GaAs:(Al,Ga)As heterostructures: The dependence 
on alloy composition," J. Appl Phys., Vol. 59, No. 1, pp. 200-209, 1986. 

[16] P. Penfield and RJ? Rafuse, Varactor Applications, MTT Press, p.86, 1962. 

[17] G. Martin, S. Strite, J. Thornton, and H. Morkoc, "Electrical properties of GaAs/GaN/GaAs 
semiconductor-insulator-semiconductor structures," Appl. Phys. Lett., Vol. 58, pp. 2375-2377, 1991. 



2- 

Third International Symposium on Space Terahertz Technology Page 93 



A Schottky/2-DEG Varactor Diode for N93*27734 
Millimeter and Submillimeter Wave Multiplier Applications 

W. C. B. Peatman, T. W. Crowe, M. Shur, and B. Gelmont S& _^ *Z 

Semiconductor Device Laboratory /, _ ; 

Department of Electrical Engineering / (?0 b £- ^—^ 



University of Virginia 
Charlottesville, VA 22903 

ABSTRACT 



A il 



A new Schottky diode is investigated for use as a multiplier element in the millimeter and 
submillimeter wavelength regions. The new diode is based on the Schottky contact at the edge 
of a 2-dimensional electron gas (2-DEG). As a negative voltage is applied to the Schottky 
contact, the depletion layer between the Schottky contact and the 2-DEG expands and the 
junction capacitance decreases, resulting in a non-linear capacitance-voltage characteristic. In 
this paper, we outline the theory, design, fabrication and evaluation of the new device. Recent 
results include devices having cutoff frequencies of 1 THz and above. Preliminary multiplier 
results are also presented. 

I. BACKGROUND 

Schottky barrier varactor diodes are used as frequency multiplier elements for local 
oscillator (LO) sources for the millimeter and submillimeter wavelength region. These sources 
are used in heterodyne receivers for a variety of applications including radio astronomy, 
atmospheric studies and plasma diagnostics. For space-based receiver systems, the LO source 
must be compact, lightweight and reliable; and power and cooling requirements must be 
minimized. While molecular gas lasers have been used as LO sources for airborne 
radioastronomy measurements at frequencies as high as 2.5 THz [1,2], the stringent 
requirements for space applications will require the use of a solid-state LO source. Although 
standard varactor diodes have been used to generate 0.7 mW at 474 GHz [3] and 0.2 mW at 
640 GHz [4], these devices will not provide usable amounts of LO power above about 1 THz 
[5]. Schottky barrier varistor diodes and quantum well oscillators have been proposed as 
sources of LO power but these technologies will not provide sufficient power to drive the GaAs 
Schottky barrier mixer diodes used in these receivers [6,7,8]. We report here on a new planar 
varactor diode in which the Schottky contact is formed at the edge of the 2-dimensional 
electron gas (2-DEG). This new device, which is essentially the 2-d analog of the standard (3- 
d) Schottky diode, has unique properties and is a promising candidate for use in millimeter and 
submillimeter wave multiplier applications [9,10]. In addition, it may be possible to use this 
device to investigate conduction in a 2-d electron gas at frequencies significantly above 100 
GHz. 



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Third International Symposium on Space Terahertz Technology 



Schottky Contact 
Pad 




Fig.l. 



Schematic of the planar Schottky 12 -DEG varactor diode. 



In Section II, the theory of the device and the design for multiplier applications is 
reviewed. In Section IE, the fabrication of the devices is briefly described. The low frequency 
evaluation is presented in Sections IV and V for the interdigital-type and the "refined 
prototype" devices, respectively. The preliminary multiplier performance of the "refined 
prototype" devices are presented in Section VI. Finally, a summary of the work and outlook for 
future research is presented in Section VII. 

II. THE SCHOTTKY/2-DEG DIODE 



A. Overview 

A sketch of the Schottky/2-DEG diode is shown in Fig. 1. Also shown is an expanded 
view of the Schottky contact region. The chip dimensions are typically 100 um by 200 \xm by 
about 50 fJ.m thick. Two device configurations are discussed in this paper. These are the 
interdigitated contact device (also described in [9,10]) and the "refined prototype" device 
which is similar to that shown in Fig. 1. The interdigitated devices have been realized in both 
single and dual anode configurations, the latter being intended for symmetric C(V) 



Third International Symposium on Space Terahertz Technology 



Page 95 



V a =0 



V =V 

v a v r 




Fig. 2. Conduction band diagram of the Schottky/2-DEG diode. The 2-DEG is bounded on 
the left by a Schottky contact and on the right by an ohmic contact. The depletion 
depths for two applied voltages are shown (left). A qualitative sketch of the potential 
well in the undepleted channel is shown (right) with the Fermi and lowest sub-band 
energies indicated. 

applications. These devices had Schottky contact widths in the range 150-350 fim. The 
"refined prototype" devices are single anode devices (whose cathode is an ohmic contact) with 
widths of about 100 |im. The isolation between pads is achieved by etching through to the 
semi-insulating GaAs substrate everywhere except in the channel region and beneath the pads. 
Further details of the fabrication process are outlined in Section HI. 



B. Physics and Equivalent Circuit 

The conduction band diagram of the Schottky/2-DEG diode is shown in Fig. 2. The 
theory of the metal/2-DEG junction was first considered in [1 1] and more recently extended in 
[12]. In [12], the junction capacitance was derived using a conformal mapping technique and 
by making suitable assumptions about the boundary conditions. The capacitance-voltage 
characteristic was given by the expression 

(R2 + d| ep ) 1/2 +R 1 



Cj(V) = ^L ln[- 



% 



(1) 



where W is the width of the contact (see Fig. 1), e is the permittivity of GaAs and R is the 



Page 96 



Third International Symposium on Space Terahertz Technology 



O- 



C.(V) 



H \-^W- 



■O 



'sh 



Fig. 3. A simple equivalent circuit model of the Schottky/2-DEG diode. 

half-height of the Schottky metalization. The depletion depth ddep of the 2-DEG is given as 

2e(V bi -V a ) 



ddep = 



qn s 



(2) 



where V bi is the built-in voltage (0.7-1.0 V), V a is the applied voltage (which is negative for 
reverse-bias), and n s is the 2-DEG sheet charge density. For a 2-DEG sheet charge density of 
10 12 cm -2 (10 13 cm -2 ) and assuming e of GaAs, the 2-DEG depletion depth is 3 |im (0.3 Jim) 
at 20V reverse bias. The total capacitance is equal to the junction capacitance in parallel with 
the shunt capacitance associated with the pad-to-pad fields and anode-to-ohmic fields. 

The equivalent circuit is shown in Fig. 3. For simplicity, the junction conductance is 
neglected. [The skin effect is probably not important for this device since the length and depth 
of the 2-DEG are small]. Also, the inductive effect of charge carrier inertia is neglected 
although this effect may be important at cryogenic temperatures due to the long momentum 
relaxation time of the 2-DEG. The junction capacitance Cj can be estimated using Eq. 1. The 
series resistance, R s , is composed of the resistance of the undepleted channel and the ohmic 
contact resistance. The former is given by 



r _ (L-ddep) 
s q^n s W 



(3) 



where L is the channel length and m is the electron mobility in the undepleted channel. The 
ohmic contact resistance is simply Roc = r sc/W where r^ is the specific contact resistivity 
(normally specified in units Qmm) of a HEMT-like ohmic contact, and W is the device width 
(in mm). The shunt capacitance term C S h includes pad-to-pad capacitance and Schottky-to- 
ohmic capacitance. For high frequency design, C S h and the Rs must be minimized. 

The theory of the junction breakdown has not been fully developed. However, Vb r in long 
channel devices is assumed to be caused either by impact ionization or by tunneling. 
Experimental observations of very large breakdown voltages in prototype devices [9] lead to 
the conclusion that for short channel length devices, the breakdown voltage was limited by 
punch-through. In this paper, those devices with channel lengths of 1 |im are probably punch- 
through limited whereas the 2-3 ^m length devices may be limited by impact ionization or 



Third International Symposium on Space Terahertz Technology Page 97 



tunneling, or both. The breakdown voltage may also be limited by the geometry of the anode 
metalization (for example, the half-height R) or by processing and material defects. A more 
detailed discussion of the breakdown in these devices will be presented in a later paper. 

C. Frequency and Power Limitations 

Several factors limit the frequency response and power performance of the multiplier. The 
frequency response may be limited by the dynamic cutoff frequency which is usually defined 
[13] as 

_ S m ax~Smin ,,v 

Vco 27tR s () 

where S max (Smin) is 1/Qnin (1/QnaxX and R s is the series resistance of the varactor diode. It is 
desirable for the device to have a v co value much higher than the operating frequency to ensure 
that the multiplier efficiency is not degraded. To achieve high v co , the series resistance should 
be as small as possible, the minimum capacitance (near breakdown) should be small and the 
capacitance modulation ratio Qnax/Qnin should be large. 

Another important quantity which may limit both the frequency response and maximum 
output power is the finite velocity of the electrons traversing the modulation region (the 
epilayer in GaAs diodes or the 2-DEG channel in the Schottky/2-DEG diode). Recently, 
Kollberg et al [14] showed how the finite electron velocity limited the current in the 6P4 diode 
used by Erickson [3]. Kollberg argued that the ac displacement current could not exceed the 
saturation current which in turn is limited by the electron drift velocity. Using Monte Carlo 
analysis, the effective velocity and the saturation current in the 6P4 diode were estimated (in 
[14]) to be 2.4 x 10 7 cm/s and 44 mA, respectively. At input powers beyond that which causes 
the current to saturate, the diode's rf impedance increases (since the current cannot). Kollberg 
used this analysis to simulate the roll-off in efficiency with input power, which was observed 
by Erickson. 

The velocity saturation current is written here for the 2-d case as 

I vs = qn s v e ffW (5) 

where n s is the 2-dimensional sheet charge density, W is the contact width and v e ff is the 
effective velocity of the electrons in the channel. As will be shown, a Schottky/2-DEG diode 
with W = 100 |im and n s = 1.85 x 10 12 cm -2 has roughly the same characteristics as the 6P4 
diode. Assuming the same effective velocity as was used for the 6P4, namely v e ff = 2.4 x 10 7 
cm/s, I vs is about 70 mA. 

The finite electron velocity may also limit the frequency response if the transit-time for 
electrons traversing the modulation region is comparable to the period of the LO frequency. 
The transit-time corner frequency has been proposed [10] as a useful design parameter for the 
diode's frequency response, and is defined as 

where L is the channel length (or the epilayer thickness in the standard diode). Note that the 



Page 98 Third International Symposium on Space Terahertz Technology 



frequency given by Eq. 6 is a corner frequency since, as the frequency increases beyond this 
value, the maximum if modulation length shortens, resulting in a smaller capacitance ratio and 
thus to a roll-off in the multiplier efficiency. For example, assuming an effective electron 
velocity of 2.4 x 10 7 cm/s (as was used in [14]), a varactor diode with input frequency of 80 
GHz will have a maximum modulation length of about 0.48 urn. In comparison, the 6P4 diode, 
which is often used at this frequency, has an epilayer thickness (and maximum dc modulation 
length) of about 1.0 um. Thus, both standard and 2-DEG diodes should be designed to achieve 
a large capacitance modulation ratio within the length given by Eq. 6. 

Having outlined the equations for the capacitance, resistance, punch-through voltage (Eq. 
2 with ddep = L), the dynamic cutoff and the transit-time frequencies and the saturation current, 
the Schottky/2-DEG diode may be designed for particular applications. This procedure is 
simplistic but is similar to the design of the state-of-the-art varactors currently in use. This 
design procedure was used for the "refined prototype" devices whose results are given in 
Section V. Before presenting the device results, the fabrication will be briefly reviewed. 

HI Fabrication 

The prototype Schottky/2-DEG devices discussed here were fabricated on a 
pseudomorphic Al .25Gaj5 As/In. isGa 85 As/ GaAs structure shown in Fig. 4. This structure was 
grown by MBE and analyzed using the Van der Pauw method to determine the mobility and 
sheet charge density. The electron sheet charge density at both 77 K and 300 K was 
1.85xl0 12 cm -2 and the electron mobilities were 31,400 cm 2 /V-s and 6640 cm 2 /V-s at 77 K 



1 


GaAs 


5 x 10 18 cm" 3 


40 A 


2 


Alo.25Gao.75As 


5 x 10 17 cm" 3 


300 X 


3 


Si Atomic Plane 


5 x 10 12 cm" 2 


~ 


4 


Alo.25Gao.75As 


~ 


50A 


5 


In 15 Ga 85 As 


— 


150 A 


6 


GaAs 


— 


5000 A 


7 




SI GaAs Substrate 





Fig. 4. AlGaAslInGaAslGaAs heterostructure used for the devices discussed here. 



Third International Symposium on Space Terahertz Technology 



Page 99 





' "" ' j y' , '^ : .^-" :i ^"*'-"l> '• --''* L^Sr*-" 4. 


W'r^' ■ ■> " -^ 


• 








8.58kx 18ku 123 



Fig. 5. Scanning electron micrographs of Schottky/2-DEG devices. The interdigitated 
device (top) has anode width of 250 \xm and channel length of 2 Jim. The "refined 
prototype" device (bottom) shown here is similar to the devices discussed in Section 
V. Here, the anode width is 100 |im and the channel length is 5 \\m. 



Page 100 Third International Symposium on Space Terahertz Technology 



and 300 K, respectively. The supply and cap layers are substantially depleted to the 2-DEG 
and/or the surface, to minimize parallel conduction. The ohmic contact consists of an 
electroplated SnNi/Ni/Au trilayer which is alloyed at about 380°C. To form the Schottky 
contact, a trench is etched through the 2-DEG layer and a Pt/Au contact is electroplated into 
the trench. Next, the contact pads are plated and finally a 2-3 micron deep NaOH:H202 etch to 
the SI-GaAs substrate is performed to isolate the two pads. All lithography levels are 
performed using a Karl Suss MJB-3 (405 nm). An SEM photo of the interdigitated device is 
shown in Fig. 5 (top). A sketch of a "refined prototype" is also shown (bottom). The rough 
surface of this device was due to the isolation etch, performed using chlorine reactive ion 
etching. A subsequent wet chemical etch reduced the surface roughness considerably. 

IV. LOW FREQUENCY EVALUATION OF INTERDIGITATED DEVICES 

The dc evaluation of the Schottky/2-DEG diodes include I(V), C(V) and reverse 
breakdown voltage measurements. First, the interdigitated device results are presented (these 
results were also presented in [10]). The forward and reverse I(V) of a single Schottky/2-DEG 
device is shown in Fig. 6 (top), measured at room temperature. The exponential diode 
characteristic is seen as the linear portion of this semi-log I(V) plot, in the range 0.35 - 0.7 V. 
The "knee" voltage (at 1 |J.A) was 0.512V. The AV values for the current intervals 0.1 - l.OjiA, 
1.0 - IO.O^iA and 10.0 - 100(iA are 74mV, 74mV and 81mV, corresponding to inverse slope 
parameters, V , of 32.1mV, 32.1mV and 35.2mV, respectively. This corresponds to a diode 
ideality factor of 1.26. The series resistance of this device was determined to be 56Q. The 
expected 2-DEG channel resistance at room temperature is 6 Q.. Allowing for a pessimistic 
value of the ohmic contact resistivity, r^ of 2.5 Qmm, the total series resistance expected for 
this device was about 16 Q. The remaining 40 Q series resistance is most likely due to 
insufficient plating of the ohmic contacts, as was substantiated by inspection using scanning 
electron microscopy. The dual anode devices have no ohmic contact resistance. In these 
devices, the I(V) is dominated by the characteristic of the reverse-biased junction and a series 
resistance measurement cannot be made. However, using Eq. 3, the L = 2|im, W = 250um 
device resistance is about 4 Q at 295K and about 1.0 Q at 100 K due to the increase in mobility 
uponcooling. 

The C(V) curves of the single and dual anode interdigitated devices are shown in Fig. 6 
(bottom). The channel length (gap between fingers) is 2 |im (3 p.m) for the dual (single) 
Schottky device. The anode widths were 250 jxm for both devices. As expected, the dual 
Schottky device has a nearly symmetric C(V) characteristic and it's zero-bias capacitance is 
about half that of the single anode device. Subtracting the pad-to-pad capacitance which was 
measured to be 4 fF, this symmetric C(V) device had a dynamic cutoff frequency of about 1 
THz at 295K and about 4 THz at 100 K. The velocity saturation current (Eq. 5 using 
v e ff = 2.4 x 10 7 cm/s) is 178 mA. Also, the transit-time corner frequency of the 2 |im channel 
length device is about 19 GHz, calculated using Eq. 6. The capacitance levels of the dual 
anode device is probably too high for most multiplier applications. Nevertheless, multiplier 
testing of these devices will be performed in the near future. 



Third International Symposium on Space Terahertz Technology 



Page 101 




3- ie-06 



4-1 
C 
d) 
M 

P 
O 



' ' ' ' 



10 15 20 23 30~~ 

Revarsa Voltage (V) 

■ 



1.4 



Forward Voltage (V) 




Voltage (V) 



Fig. 6. Forward and reverse I(V) of single anode interdigitated device (top) with 
W = 250 |im and L = 3.0 |im. C(V) characteristics (bottom) ofL = 3.0 (im single and 
L = 2.0 |im dual anode interdigitated devices of width 250 p.m. 



Page 102 



Third International Symposium on Space Terahertz Technology 



40 





35 




30 


b4 




4-1 




^^ 


25 


<U 




O 




C 
(0 


20 


■P 




-H 




u 


15 


a 




nJ 




u 


10 




5 



I I 



Data -» 
Fit - 



4 6 8 10 

Reverse Voltage (V) 



12 



14 



Fig. 7. C(V) of refined prototype device A. Also shown is a fit (Eq. 1) using R = 0.75 fim, 
C sh = 8.0 fF, V bi = 0.7 V andW = 90 ^im. 

V. LOW FREQUENCY EVALUATION OF REFINED PROTOTYPE DEVICES 

The refined prototype devices had two anode width/channel length combinations. The 
"A" devices had anode widths of 90 |im (on average) and channel lengths of 2.5 |im while the 
"B" devices had anode widths of 80 [im and channel lengths of 3.0 (im. The C(V) characteristic 
of device A is shown in Fig. 7. The theoretical capacitance, shown fitted to the data, agrees 
well with the data except near zero-bias where the fit is lower than the data. The fit assumed a 
reasonable value of the anode metal half-height (R = 0.75 |im) and a shunt capacitance of 8.0 
fF. This value of C S h is higher than expected since the pad-to-pad capacitance was measured to 
be about 2 fF. The additional shunt capacitance may be due in part to fields between the anode 
and ohmic metals. This contribution to the capacitance is not easily determined and is also not 
substracted for the v c0 calculations. The difference between the theory and the data near zero- 
bias is either due to inaccuracy of Eq. 1 for the geometry of this device or to effects related to 
the leakage current at low bias. We are currently investigating a more general theory of the 
junction capacitance for devices of various geometries. 

The forward I(V) as a function of temperature of device A is shown in Fig. 8. As the 
temperature decreased, several changes occured. First, the entire I(V) curve shifted to higher 
voltages, as expected due to the temperature dependence of the saturation current (the theory of 
the thermionic saturation current of the Schottky/2-DEG junction is being investigated [15]). 
At lower currents, the "leakage" current which has been routinely observed at room 
temperature is seen to decrease substantially so that, at 220 K, it is much less than one 
nanoamp. Finally, the strong temperature dependence of the series resistance is evident at high 
currents. 



Third International Symposium on Space Terahertz Technology 



Page 103 



le-01 



le-02 - 



le-03 



le-04 





le-05 


4-> 




c 




u 


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le-10 



CO 

6 
O 



O 
C 
(0 

4-> 
CO 

•H 
CO 
0) 

PS 



0.2 0.4 0.6 0.8 
Forward Voltage (V) 



1 1 1 1 1 1 r— i i 


1 


— r- 


- 


























syyyjfow 








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.'//.•//'•'c'ty 








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////////''•/& 


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<■' '/■■''/.'''■'■'// 
< ' / ; / /.'a.' /if 


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~ 


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K 





' / - f / ' iliiff 


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K 






160 


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- 


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' / ,-'•"' ..-•'"" /////if 


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K 


— 




280 


K 


. 


1 1 1 1 1 1 1 1 1 


299 


K 





1.0 



1.2 




100 150 200 

Temperature (K) 
Fig. 8. Forward I(V) (top) and series resistance (bottom) versus temperature of device A. 



Page 104 



Third International Symposium on Space Terahertz Technology 



-p 
c 

CD 

u 
u 

o 

<v 

co 

u 

CD 

> 

OS 



le-04 


1 1 1 1 1 1 1 — 


— 1 1 1 1 T 1 1 1 


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: / / S* 






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- 


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K " 




ki! '■ '• : : 


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K 




Ki! ! ■ ij. 

p!>-~- 1 -'.. 


*//.-■■' 80 


K - 






//-•'' 100 


K 


le-07 


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'' 140 


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K — - " 




168 


K 




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K - 




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le-08 


1 / / /j0- 


220 
240 


K 

K - — " 






260 


K 




.' t / .'IB?*' 


295 


K - 


le-09 


/ / § 

1 1 1 1 1 1 1 


-1 1 1 .1,. J 1 L 1 


1 



Fig. 9. 



2 4 6 8 10 12 

Reverse Voltage (V) 
Reverse I(V) versus temperature of device A. 



14 



16 



18 



The measured series resistance of device A is plotted as a function of temperature in Fig. 
8 (bottom). The resistance decreased nearly linearly from 19.9 Q at 300 K to 10.5 Q. at 130 K, 
and the resistance at 77 K was 8.9 Q. The 2-DEG resistances calculated using Eq. 3 with the 
low field mobility values at 300 K and 77 K are 14.1 Q. and 3.0 Q, respectively. Thus, the 
ohmic contact resistivity (for the width 90 Jim) is estimated to be about 0.6 Qmm. This value 
of Tsc is much lower than was achieved on the earliest Schottky/2-DEG devices. The 
improvement is probably a consequence of the higher doping at the heterostructure surface and 
an improved ohmic plating and alloying procedure. Further reduction of r sc (perhaps to 0. 1 
Qmm) should be possible using evaporated Ni/Ge/Au ohmic contacts. 

The reverse I(V) of device A was measured as a function of temperature and is shown in 
Fig. 9. As in the forward I(V), the leakage current is seen to decrease upon cooling. At the 
highest reverse current (100 (j.A),"the reverse voltages decreased with temperature from 17 V at 
300 K to about 9 V at 40 K. This temperature dependence of the breakdown voltage is 
qualitatively consistent with impact ionization theory. Since the mean free path for electron 
phonon interactions increases with decreasing temperature, the electrons can achieve higher 
kinetic energies before phonon scattering occurs. Consequendy, as the temperature decreases, 
electrons reach the impact ionization energy at lower field strengths (lower reverse voltages) 
and the breakdown voltage decreases. 



Third International Symposium on Space Terahertz Technology 



Page 105 



le-02 
le-03 
le-04 



le-05 

c 

u le-06 
u 

5 le-07 



le-08 

le-09 - 




1 1 1 1 1 1 1 1 1 1 1 


-i r 


i 








/ ■ ' / ' / / / 






/ / / / / / if 






/ / •' ' / / 1 






■■ ■■ ■' 1 i i 






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- 


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/flllli 




- 


! ••" / / / /// 


299 


K 


/////:.'/ 


260 


K 




220 


K — - 


••' / / / / : '• 


180 


K — . 


/////// 


140 


K — 


100 


K 


/ / / / / / ! 


60 


K 


////// 


24 


K — 


////// 






i i i i i i i i i i i 


< i 


i. 



le-04 



le-05 



~ le-06 

■p 
c 

m le-07 
o 



le-08 



le-09 



0.2 0.4 0.6 0.8 1.0 1.2 1.4 
Forward Voltage (V) 

ii i i i ■ — 



"22K" 

"5 OK" 
"100K" 
"142K" 
"200K" 
"263K" 
"299K" 




10 15 20 

Reverse Voltage (V) 



25 



30 



Fig. 10. Forward I(V) (top) and reverse I(V) (bottom) vs. temperature of planar GaAs 
Schottky diode SC6T1 . 



Page 106 Third International Symposium on Space Terahertz Technology 



The temperature dependence of the breakdown voltage of the Schottky/2-DEG diode 
raised the question of whether the breakdown voltage in standard GaAs varactors had a similar 
temperature dependence. To investigate further, the I(V) of several planar GaAs varactor 
diodes (UVA-SC6T1) were measured as a function of temperature. The doping concentration 
and anode diameter for this diode were 2 x 10 16 cm -3 and 6 urn, respectively, compared to the 
respective values 3 x 10 16 cm -3 and 6 Jim for the whisker-contacted UVA-6P4 diode. The 
forward and reverse I(V) of one device is shown in Fig. 10. The forward I(V) is typical for a 
low doped diode. Here, the knee voltage increased from 0.678 V at 299K to 1.015 V at 24K. 
The inverse slope parameter decreased from 28.7 mV at 299 K to 12.7 mV at 24 K. The series 
resistance was constant and equal to 30 Q (± 1 Q) over the range 24-299 K. The reverse I(V) 
was measured from 10 nA to 10 ^lA. The breakdown voltage characteristics are all very sharp. 
Of the three SC6T1 diodes tested, all had breakdown voltages of 18-19 V at 24K, although the 
room temperature breakdown voltages were 26 V, 29 V and about 30 V. Thus, the average 
reduction in Vbr was about 35 percent upon cooling, compared to a reduction of about 50 
percent for the Schottky/2-DEG diode. A decreasing breakdown voltage in these devices upon 
cooling may impact the low temperature multiplier performance. On the other hand, the large 
decrease in the series resistance of the 2-DEG upon cooling should result in higher multiplier 
efficiency. 

Finally, the saturation drift current of device A was calculated, using Eq. 5 with 
v e ff = 2.4 x 10 7 cm/s, to be 64 mA. This may be a conservative estimate since higher velocities 
may be possible in the 2-DEG than in bulk GaAs. In any case, the Schottky/2-DEG diode of 
roughly equivalent properties as those of the 6P4 diode has a significantly higher saturation 
current (at least 64 mA) than does the 6P4 diode (44 mA according to [14]). 

VI. PRELIMINARY MULTIPLIER RESULTS 

The first rf measurements of a Schottky/2-DEG diode were performed at the National 
Radio Astronomy Observatory using a modified NRAO tripler which was designed [16] for 
whisker contacted diodes (such as the UVA-6P4). The Schottky/2-DEG diode chip was 
soldered across the output waveguide. A klystron was used as a source and two power meters 
were used to measure the power at the input and output ports of the multiplier. Further details 
of the measurement setup may be found in [17]. 

The preliminary measurements were of device A having chip width and length 
dimensions of 100 (im and 180 |im. The chip thickness was about 60 fim. After soldering, the 
clearance between the back of the chip and the waveguide wall was estimated to be only 2-3 
(jm, due in part to the thick solder "bump". The I(V) of the device was checked after soldering 
and found to be the same as before soldering. The room temperature R s and Wy^ of this device 
were 20 Q and 12 V, respectively. 

The first multiplier test was to determine the optimum frequency. Using 50 mW input 
power, the multiplier tuners and dc bias were optimized for maximum output power at several 
frequencies over the range 70-79 GHz. The result is shown in Fig. 1 1 where the input power 
was 50 mW. The best performance was obtained at 75 GHz where P out was 160 p.W at 225 
GHz. The input return loss was measured at 75 GHz to be about -20 dB so the input tuning was 
relatively well optimized. Next, the output power at 225 GHz was measured as a function of 



Third International Symposium on Space Terahertz Technology 



Page 107 





200 




180 




160 


3 


140 


u 


120 


0) 




15 

o 


100 


4-> 

3 

a 

4-> 

3 

o 


80 
60 




40 




20 









550 




500 




450 


s 

3 


400 
350 


M 


300 


o 


250 


-p 




3 

-p 


200 


3 
O 


150 




100 




50 




72 74 76 

Input Frequency (GHz) 



80 




40 60 80 

Input Power (mW) 



100 



120 



Fig. 11. Tripler output power versus input frequency (top) with Pin = 50 mW and output 
power versus input power (bottom) for tripling to 225 GHz. 



Page 108 Third International Symposium on Space Terahertz Technology 



the input power, as shown in Fig. 11 (bottom). P^ was varied over the range 20 -100 mW. As 
the input power increased, the output power increased, reaching a maximum value of 500 |iW 
at 100 mW at the input. This corresponds to an efficiency of 0.5 percent. In comparison, 
Bradley [17] used this multiplier with a planar GaAs varactor of doping 1.1 x 10 17 cm -3 and 
diameter 7 |im and obtained 3.7 mW output power and 4.1 percent efficiency at 219 GHz. This 
first multiplier measurement of the Schottky/2-DEG diode is encouraging but much higher 
performance should be possible. Even the current devices should yield higher multiplier 
efficiencies if they are thinned to 25 [im or less and if a thinner solder layer is used to reduce 
the shunt capacitance between the chip and the waveguide wall. 

vn. Summary and future research 

In summary, we have reported on the recent progress in the research of a novel 
Schottky/2-DEG varactor diode. Observations of reduced breakdown voltages upon cooling in 
both the standard GaAs and the novel Schottky/2-DEG diodes were in agreement with the 
theory of impact ionization. The problem of current saturation was discussed and the 
Schottky/2-DEG diode of width 90 Jim was found to have a significantly higher saturation 
current than the comparable GaAs 6P4 varactor. Recent improvements to the design and 
fabrication procedures have resulted in devices having lower series resistance and lower 
capacitance. Both single anode and dual anode (with symmetric C(V)) devices have been 
investigated. The cutoff frequency of the dual anode device was estimated to be 1 THz (4 THz) 
at 300 K (100 K), whereas the single anode device had cutoff frequency of about 0.6-1.0 THz, 
depending on the temperature. Preliminary multiplier measurements of a single anode device 
were encouraging, resulting in 500 fiW output power at 225 GHz with 0.5 percent efficiency. 
Ongoing research will include more extensive multiplier testing of both the single anode and 
the symmetric C(V) devices. Also, shorter channel length devices with Ni/Ge/Au ohmic 
contacts will be fabricated to achieve much higher cutoff and transit-time frequencies. In 
addition, AlInAs/InGaAs/InP heterostructures will be investigated. Finally, the theories 
relating to the junction capacitance and breakdown are being developed and the current 
transport in these devices will be investigated using Monte Carlo simulations. 

ACKNOWLEDGEMENTS 

This work has been supported by the National Science Foundation under contracts ECS- 
8720850 and ECS-91 13123 (contract monitors Dr. T. Hsiang and Dr. B. Clifton) and by. the 
Office of Naval Research under contract #N00014-90-J-4006 (contract monitor Dr. Y.S. Park). 
The authors thank R. Bradley and N. Horner of the National Radio Astronomy Observatory for 
assistance with the multiplier measurements, and T. Hierl of Quantum Epitaxial Designs, Inc. 
for providing the MBE material and Van der Pauw data. 



Third International Symposium on Space Terahertz Technology Pagel09 



REFERENCES 

[I] H. P. Roser, "Heterodyne Spectroscopy for Submillimeter and Far-Infrared Wavelengths From 100 
urn to 500 urn," Infrared Physics, Vol. 32, pp. 385-407, 1991. 

[2] J. Zmuidzinas, A.L. Betz, and R.T. Boreiko, "A Corner-Reflector Mixer Mount for Far Infrared 
Wavelengths," Infrared Phys., Vol. 29, No. 1, 119-131, 1989. 

[3] N. Erickson, "High Efficiency Submillimeter Frequency Multipliers," The 1990 IEEE MTT-S Int. 
Microwave Symposium, Dallas, TX, May 1990. 

[4] P. Zimmermann, private communication, March and May 1990. 

[5] T. W. Crowe, W.C.B. Peatman, and E. Winkler, "GaAs Schottky Barrier Varactor Diodes for 
Submillimeter Wavelength Power Generation," Microwave and Optical Tech. Lett., Special Issue: 
Space Terahertz Tech., Vol. 3, No. 1, January 1991. 

[6] K. Benson and M.A. Frerking, "Theoretical Efficiency for Triplers Using Nonideal Varistor Diodes 
at Submillimeter Wavelengths," IEEE Trans. Microwave Theory Tech., Vol. MTT-33, No. 12, 
1367-1374, Dec. 1985. 

[7] C. Kidner, I. Medhi, J. East, G. Haddad, "Performance Limitations of Resonant Tunneling Diodes," 
The First Int'l. Symposium on Space Terahertz Technology, Ann Arbor, MI, March 1990. 

[8] H. Rothermel, T.G. Phillips, J. Keene, "A Solid-State Frequency Source for Radio Astronomy in 
the 100 to 1000 GHz Range," Int. J. Infrared and Millimeter Waves, Vol. 10, No. 1, 83-100, 1989. 

[9] W. C. B. Peatman, T. W. Crowe and M. Shur, "Design and Fabrication of Heterostructure Varactor 
Diodes for Millimeter and Submillimeter Wave Multiplier Applications," Proc. IEEE/Cornell 
Conf. on Advanced Concepts in High Speed Semic. Dev. and Circuits, Ithaca, NY, 1991. 

[10] W.C.B. Peatman, T.W. Crowe and M. Shur, "A Novel Schottky/2-DEG Diode for Millimeter and 
Submillimeter Wave Multiplier Applications," IEEE Electron Device Lett., Vol. 13, No. 1, pp. 11- 
13, January 1992. 

[II] S.G. Petrosyan and A. Ya Shik, "Contact Phenomena in a two-dimensional electron gas," Soviet 
Physics Semicond., 23 (6), pp. 696-697, June 1989. 

[12] B. Gelmont, M. Shur and C. Moglestue, "Theory of Junction Between Two-Dimensional Electron 
Gas and P-Type Semiconductor," to be published, IEEE Trans. Electron Devices, Vol. 39, No. 5, 
May 1992. 

[13] P. Penfield and R. Rafuse, Varactor Applications, MIT Press, 1962, p. 86. 

[14] E. Kollberg, T. Tolmunen, M. Frerking and J. East, "Current Saturation in Submillimeter Wave 
Varactors," Proc. Second Int'l. Symp. Space Terahertz Technology, Pasadena, CA, pp. 307-322, 
1991. 

[15] B. Gelmont, M. Shur (unpublished). 

[16] J. Archer, "An Efficient 200-290 GHz Frequency Tripler Incorporating A Novel Stripline 
Structure," IEEE Trans. Micrwave Theory and Tech., Vol. 32, No.' 4, pp. 416-420, 1984. 

[17] R. Bradley, Ph.D. Thesis, University of Virginia, pp. 42-65 May, 1992. 



Page UO Third International Symposium on Space Terahertz Technology 

3?~33 N9 3 „ a? 7 35 



I 



THERMIONIC EMISSION CURRENT IN A SINGLE BARRIER V ARACTOR 

Hans Hjelmgren 3 ), Jack East b >, and Erik Kollberg 2 ) 

a) Applied Electron Physics, Chalmers University of Technology, S-412 96 Goteborg, 

Sweden 

b) Solid-State Electronics Laboratory, University of Michigan, Ann Arbor, MI 48109- 

2122 

/ Abstract — From I-V measurements on Single Barrier Varactors (SBV) at different 
, temperatures we concluded that thermionic emission across the barrier of the actual device 
: is mainly due to transport through the X band. The same structure was also modelled with 
a one-dimensional drift-diffusion model, including a "boundary condition" for thermionic 
emission across the heteroj unction interface. By including thermionic field emission 
through the top of the triangular barrier of a biased diode and the effect of a non-abrupt 
interface at the heterojunction, we obtained good agreement between the modelled and 
measured I-V characteristics. 

1. Introduction 

SBV diodes have been proposed as an alternative to Schottky barrier diodes for harmonic 
multiplier applications [1]. However, the device discussed in [1] showed a higher than 
expected current. We will present experimental data and the results of a numerical model to 
explain the current vs. voltage characteristics of SBV diodes. 

2. Experimental results 

The device consists of a 200 A wide undoped Alo^GagjAs barrier, surrounded by 5300 A 

wide GaAs layers, doped to lxlO 17 cm -3 . On both sides of the AlGaAs layer there is a 50 
A wide undoped GaAs spacer layer. If the effect of field emission is neglected the current is 
mainly limited by thermionic emission across the barrier and can be described by the 
Richardson law, 

J = A*T 2 e-W v )/ kT (1) 

where A* is the modified Richardson constant and <{>b is the bias dependent barrier height 
Both can be determined from the experimental data by plotting InCJ/T 2 ) against 1000/T for 
different voltages [2]. The intersection with the y-axis gives us the Richardson constant, 
while the slope is proportional to the barrier height for that specific bias voltage. Since our 
barrier is comparatively thin, the influence of tunneling is observable even at quite high 
temperatures, causing a deviation from a straight line in Fig. 1. This makes the 
determination of the Richardson constant quite cumbersome. It must be chosen in such a 
way that the barrier heights at low voltages are close to the expected barrier height at zero 
bias voltage. A Richardson constant of 0.30 Acm _2 K -2 , much lower than the value 
expected for thermionic emission in GaAs but in fairly good agreement with that measured 
by Solomon et al. [2], corresponds to a barrier height at low voltages of about 0.29 eV. 
This low Richardson constant together with the high current indicates that the electrons are 

transferring to the X valley within the barrier. The assumed offset in T conduction band 
corresponds to 59 % of the difference in bandgap [3]. 



Third International Symposium on Space Terahertz Technology Page 111 



3. Numerical results 

A drift-diffusion model for one-dimensional heterojunction structures was also used to 
study the device characteristics. It accounts for thermionic emission across the barrier by 
calculating the current at the heterojunction interfaces from, 

J = qTl 2 n 2 v r2 - A^erMYJ/kT (2) 

where the first term describes the current from the AlGaAs side [4] while the second term is 
the current from the GaAs side emitted above the barrier. Without this "bottleneck" for the 
current, the drift-diffusion model results in a much too high current. Since a theoretical 
determination of the actual emission constant across the interface is quite complicated, we 
have used the experimentally determined Richardson constant in Fig. 1. The probability of 
emission above the barrier depends on several parameters, such as thickness of the barrier, 
crystal orientation, and roughness of the interfaces. The thermally emitted electrons were 
assumed to be transferred to the X-valley when they reached the AlGaAs barrier, causing 
an effective barrier height at zero bias of 0.29 eV. During a simulation the barrier height is 
determined self consistently from the modelled conduction band, Fig. 2. Even at 300 K the 
effect of thermally assisted tunneling is considerable, and it has to be included in the model. 
Since the top of the biased barrier is triangular, the probability of tunneling as a function of 
electric field and carrier energy can be estimated from the WKB approximation [5], 

where E m ax is the electric field in the barrier and Ae is the energy distance from the top of 
the barrier. The effect is included in the model by reducing the barrier height in eqn. (2) 

with an amount Ae corresponding to a tunnelling probability of e -1 . We assumed the 
tunneling process to be indirect tunneling in the AlGaAs X band [6], and used a transverse 

electron mass in the X band of 0.20mo. Tunneling through the T band is less probable due 
to the much higher barrier, and tunneling from X valleys with the longitudinal mass 
perpendicular to the interface is less probable due to the higher mass. Apart from band 
bending due to accumulation of electrons at the interface, and thermionic field emission, the 
fact that the transition between the two materials is not completely abrupt will also result in 
a field dependent barrier height [7]. In Fig. 2 we roughly modelled this effect by using a 
grid distance of 25 A at the interface. The obtained I-V characteristic in Fig. 3 is very 
sensitive to the grid distance at the right interface of the barrier, while its value elsewhere is 
of less importance. 

The model has also been used to predict the capacitance vs. voltage. The capacitance is 
determined from the change of charge concentration in the depletion layer for an 
incremental change in voltage. As can been seen in Fig. 4, the agreement between modelled 
and measured C-V characteristics is fairly good. In order to get reasonably good accuracy 
the grid-distance in the depletion layer and the voltage step should not be too large. 

4. Conclusions 

The presented expression for the current between the two grid-points adjacent to the 
heterojunction interface (eqn. 2) includes effects caused by the existence of two interfaces 
and other effects, which for a theoretical determination require quantum mechanical 
calculations [8]. It also gives us a possibility to model the effect of a non-abrupt transition 
between the two materials. Similar expressions, have been used before in drift-diffusion 
models, but they are restricted to a single interface [3,4]. The drawback with the model 
used here is that it relies on measured results in order to find a value for the Richardson 



Page 112 Third International Symposium on Space Terahertz Technology 



emission constant. However, since the actual emission constant may be quite difficult to 
calculate, we considered it as a device parameter specific for the actual device dimensions 
and growing conditions, which could be measured instead of calculated. The good 
agreement with measured results indicates that the current is mainly due to thermionic 
emission and thermionic field emission across the X-valley of the barrier. 

The inaccuracy in the experimental determination of the Richardson constant can be reduced 
by using slightly thicker barriers and by performing measurements at elevated temperatures 
and lower bias voltages. 

References 

1 . A. Rydberg, H. Gronqvist, and E. Kollberg, "Millimeter- and submillimeter-wave 
multipliers using quantum-barrier-varactor (QBV) diodes," IEEE El. Device Lett., vol. 
11, pp. 373-375, 1990. 

2. P. Solomon, S. Wright, and C. Lanza, "Perpendicular transport across (Al,Ga)As and 

the r to X transition" Superlattices and Microstructwes, vol. 2, pp. 521-525, 1986. 

3 . G. B. Tait and C. R. Westgate, "Electron transport in rectifying semiconductor alloy 
ramp heterostructures," IEEE Trans. Electron Devices, vol. 38, pp. 1262-1270, 1991. 

4. K. Horio and H. Yanai, "Numerical modeling of heteroj unctions including the 
thermionic emission mechanism at the heterojunction interface," IEEE Trans. Electron 
Devices, vol. 37, pp. 1093-1098, 1990. 

5. E. H. Rhoderick and R. H. Williams, "Metal-Semiconductor Contacts, 2nd edition," 
Oxford, England: Claredon, 1988. 

6. E. E. Mendez, E. Calleja, and W. L. Wang, 'Tunneling through indirect-gap 
semiconductor barriers," Physical Rev. B, vol. 34, pp. 6026-6029, 1986. 

7. S. C. Lee and G. L. Pearson, "Rectification in A^Ga^As-GaAs N-n heterojunction 
devices," Solid-State Electronics, vol. 24, pp. 561-568, 1981. 

8. M. Rossmanith, J. Leo, and K. von Klitzing, "Model of T to X transition in thermally 
activated runnel currents across Al x Gai_ x As single barriers," J. Appl. Phys., vol. 69, 
pp. 3641-3645, 1991. 



Third International Symposium on Space Terahertz Technology 



Page 113 




1 2 3 4 5 6 7 8 9 10 11 12 13 
1000/T, K-l 

Fig. 1. Plotting of experimental data in a InCJ/T 2 ) vs. 1000/T diagram. The applied 
voltages are 0.1 V, 0.5 V, 1.0 V, 1.5 V, 2.0 V, and 2.5 V (filled squares). 




— o 

■- i 

- 2 



100 200 300 400 500 600 700 800 900 
Distance, nm 

Fig. 2. Simulated conduction bands for different applied voltages (0.0 V, 1.0 V, and 
2.0 V). 



Page 114 



Third International Symposium on Space Terahertz Technology 



1.0000 




Modelled 

O Measured 



0.0001 



0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 
Voltage, V 

Fig. 3. Comparison between modelled and measured I-V characteristics at 300 K. The 
effects of thermally assisted tunneling and a nonabrupt heterojunction interface are 
included in the simulation. 



0.0022 

0.00 
0.0018 



<n 0.0016 

e 

^ 0.0014 

o 0.0012 

£ 0.001 

o 

* 0.0008 

<o 

u 0.0006 

0.0004f 

0.0002 






— .— Modelled 
O Measured 



0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 5.5 6 
Voltage, V 

Fig. 4. Comparison between modelled and measured C-V characteristics at 300 K. 



Third International Symposium on Space Terahertz Technology Pagell5 

N93-27736 

PROGRESS ON SINGLE BARRIER VARACTORS FOR 

SUB MILLIMETER WAVE POWER GENERATION S/o "^33 

Svein M. Nilsen, Hans Gronqvist, Hans Hjelmgren, Anders Rydberg, cf 

and Erik L. Kollberg I - ' 

The Millimeter Wave Group 
Department of Applied Electron Physics, Chalmers University of Technology, 

S -4 12 96 Goteborg, Sweden 

ABSTRACT 

Theoretical work on Single Barrier Varactor (SBV) diodes, indicate that the 
efficiency for a multiplier has a maximum for a considerably smaller capacitance 
variation than previously thought. The theoretical calculations are performed, both with 
a simple theoretical model and a complete computer simulation using the method of 
harmonic balance. Modelling of the SBV is carried out in two steps. First, the 
semiconductor transport equations are solved simultaneously using a finite difference 
scheme in one dimension. Secondly, the calculated IV-, and CV characteristics are input 
to a multiplier simulator which calculates the optimum impedances, and output powers at 
the frequencies of interest. Multiple barrier varactors can also be modelled in this way. 
Several examples on how to design the semiconductor layers to obtain certain 
characteristics are given. The calculated conversion efficiencies of the modelled 
structures, in a multiplier circuit, are also presented. Computer simulations for a case 
study of a 750 GHz multiplier show that InAs diodes perform favourably compared to 
GaAs diodes. InAs and InGaAs SBV diodes have been fabricated and their current vs. 
voltage characteristics are presented. In the InAs diode, was the large bandgap 
semiconductor AlSb used as barrier. The InGaAs diode was grown lattice matched to an 
InP substrate with InAlAs as a barrier material. The current density is greatly reduced 
for these two material combinations, compared to that of GaAs/AlGaAs SBV diodes. 
GaAs based diodes can be biased to higher voltages than InAs diodes. 



1. INTRODUCTION 

A SBV device, in its simplest form, consists of a thin layer of a large bandgap 
material which acts as a barrier, and a thicker layer of a small bandgap material on each 
side of the barrier. A cross section of a typical SBV diode is shown in Fig. 1. The low 
bandgap material at both ends of the device is normally highly doped in order to make 
possible, the formation of low resistance contacts. Provided the layer thicknesses and 
doping concentrations are symmetrical around the barrier, the current I vs. voltage V and 
the capacitance C vs. voltf.ge V will be be symmetrical around zero volt. An applied rf- 
voltage will then generate only odd harmonics. This makes it possible to design higher 
order multipliers with fewer idler circuits and thus less losses. Also the design 
procedure and mechanical construction of a higher order multiplier, making use of a 
SBV device as the non linear element will become much easier compared to that of one 
using a Schottky diode. It is the purpose of this paper to give an overview of our work 
on SBV diodes [1][2][15]. 



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Ohmic contact 




Substrate 



Ohmic contact 



Fig. 1 . Schematic cross section of a SBV mesa diode. LI is the highly doped contact 
layer, L2 n and L4 n are the depletion regions on either side of the barrier, index 
n denotes sublayers, L3 n is the barrier and L5 is part of the buffer/substrate 
layer. In the simulations LI and L5 are set equal. Mesa diameter = a. 



2. DIODES 

2.1 Quality factor of SB V's 

The cutoff frequency for a varactor, as defined in Eq. 1, is often used as a quality 
factor, suggesting that C ma x should be as large as possible for a fixed C m i n . 



fr = 



1 



1 



1 



2 k R s i \ Cmin Q 



max 



(1) 



where Rslo is the dc series resistance of the device, C max and C m in are the calculated 
device capacitances for accumulated and depleted low doped epilayers, L2 and L4 in 
Fig.l, respectively. 

However, a simple analysis show that an optimum C max /C m in ratio exist. Choosing a 
CV characteristic as shown in Fig. 2 and assuming that all harmonics, except the first 
and third harmonics are shortcircuited over the variable capacitance, only these two 
frequency components have to be considered. The bias voltage is of course always zero 
volts. This makes the analysis quite simple. In Fig. 3 we have plotted the efficiency vs. 

C m ax/Cmin for different values of R s , assuming l/a)C m i n = 180 Q. The important 
conclusion from this graph is that there is an optimum value for C ma x/C m in , and that 
too large C max values will deteriorate the multiplier's efficiency. Table 1 shows input 
and output impedances. Although this model is much simplified the results show that the 
input is highly reactive with a fairly low resistance while the output seems to be easier to 
match. This result will be confirmed below in the harmonic balance simulations 
employing a modified version of the computer program by P.H. Siegel et.al. [8]. 



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-max 




C = (C max + Cmin)^ + [(Cmax - Cmin)/2] cos(V) 



C = Co = (C max + Cmin)/2 



Fig. 2 Approximate CV characteristic of SB V diode used in the simplified analysis. 
Vmax is the maximum voltage across the diode. 




(C /C . ) 
v max mm 7 



Fig. 3 Efficiency vs. the C m ax/Cmin ratio for different values of R$. 

Table 1 Calculated efficiencies, optimum input and output imedances for a SBV diode 
tripler with the approximate cosine shaped CV characteristic for different series 
resistances. 







SOURCE IMPEDANCE 


LOAD IMPEDANCE 


RsQ 


Tlmax % 


Rl n 


XI fl 


R3ii 


X3Q 


2.5 


65.5 


8.6 


91.0 


29.5 


34.0 


5 


46.7 


10.9 


90.0 


30.4 


33.0 


10 


27.2 


15.7 


95.0 


30.2 


36.0 


20 


11.8 


25.0 


90.0 


23.0 


33.0 


40 


2.9 


44.1 


106.0 


14.0 


38.0 



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Third International Symposium on Space Terahertz Technology 



2.2 



MEASUREMENTS 



2.2.1 DC measurements 

We have fabricated several diodes based on the materials: GaAs, InGaAs and InAs, 
see Table 2 for details. Our efforts at modelling these devices are however, more recent. 
For the GaAs based diode we use a barrier of Al x Gai- x As, in the InGaAs case we use 
AUnAs as a barrier. The latter diode is grown lattice matched on an InP substrate. For 
the InAs diodes we use an AlSb barrier which gives a very high barrier, 1.3 eV for 
indirect transitions. It should therefore be very efficient in blocking the current through 
the varactor. The InAs device is grown on a GaAs substrate with thick GaAs and InAs 
buffer layers. 

Table 2 Material from which diodes of various sizes have been 
fabricated. 



Wafer #ID& 

Crystal 

Grower 



Material system 



Layer thickness (top/down) and 
doping concentration 



= 400 nm, Nd = 3.4* 10 18 [cm' 3 ] 
= 533 nm, Nd = 1.0*10 17 [cm* 3 ] 
= 5.3 nm, undoped 
: 21.3 nm, undoped 
= L22, L4i = L2i 
1998 n m, Nd = 3.4* 10 18 [cm' 3 ] 



#614 
IMEC 



GaAs/Al x Gai. x As/GaAs 
x = 0.70 in L3 

Buffer n + GaAs 
Substrate: n + GaAs 



LI 
L2i 
L22 
L3 = 
L42 
L5 = 



#1566 
Chalmers 



InAs/AlSb/InAs 
Buffer n + InAs, and n + GaAs 
Substrate: n + GaAs 
Barrier thickness = L3 



LI = 200 nm, Nd = 5.0* 10 18 [cm" 3 ] 

L2i = 400 nm, Nd = 6.0*10 16 [cm" 3 ] 

L22 = 5 nm, undoped 

L3 = 20 nm, undoped 

L42 = L22, L4i = L2i 

L5 = 2000 nm, Nd = 5.0*10 18 [cm' 3 ] 



200 nm, Nd = 5.0* 10 18 [cm' 3 ] 
= 320 nm, Nd = 1.0*10 17 [cm" 3 ] 
= 5 nm, undoped 
= 25 nm, undoped 
= L22,L4i = L2i 

2000 nm, Nd = 5.0* 10 18 [cm' 3 ] 



#1567 
Chalmers 



InAs/AlSb/InAs 
Buffer n + InAs, and n + GaAs 
Substrate: n + GaAs 
Barrier thickness = L3 



LI i 
L2i 
L2 2 
L3 ■■ 
L42 
L5 = 



400 nm, Nd = 4.0* 10 18 [cm' 3 ] 

400 nm, Nd = 6.0* 10 16 [cm" 3 ] 

25nm,Nd = 6.0*10 16 [cm- 3 ] 



#ST1 
Tampere 



Lattice matched 
InGaAs/AlInAs/InGaAs 
Buffer. n + InGaAs 
Substrate: n + InP 
Barrier thickness = L3 



LI 
L2 
L3 
L4 
L5 



= L2 



= 1000 nm, Nd = 4.0* 10 18 [cm' 3 l 



" •" " iz^, 

. 200 nm, Nd = 2.6* 10 18 [cm" 3 ] 

> 150 nm, Nd = 1.0* 10 17 [cm" 3 ] 

: 14 nm, undoped 

= L2 

■ 3000 nm, Nd = 2.6*10 18 [cm' 3 ] 



#1047 
IMEC 



InAs/AlSb/InAs 
Buffer n + InAs, and n + GaAs 
Substrate: n + GaAs 
Barrier thickness = L3 



Ll = 
L2 = 
L3 = 
LA- 
L5> 



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The influence of the epilayer design on the IV-, and CV characteristics has been 
investigated theoretically in some detail. 

The different SBV diodes fabricated by us have been DC characterized with a DC 
parameter analyzer. Most of the measurements have been done on large area diodes, 
100- 2500 |im 2 . A limited number of measurements have been made on small area, 
7 |im 2 , diodes of the #614 type. 

Diodes characterized are one GaAs varactor with an AlGaAs barrier, one diode grown 
on InP with InGaAs and InAlAs epilayers lattice matched to the substrate and two InAs 
diodes with AlSb barriers grown on GaAs substrates. The material from which we have 
fabricated devices are listed in Table 2. DC characterization of the different SBV diodes 
show that the current density is decreased as the barrier height is increased as expected, 
see Fig. 4. The InAs/AlSb material system has a 1.3 eV barrier for indirect transitions 
and = 2 eV for direct. The lowest current density is obtained for InAs/AlSb devices. 
They also show a large variation in their characteristics, a factor of two or more in 
current density over a distance of a few millimeters on the wafer. Whether this variation 
is caused by the epilayers being severely lattice mismatched to the substrate or by some 
growth parameter, is difficult to say. In comparison, diodes of the lattice matched 
InGaAs/InAlAs materials grown on InP substrate as well as those of GaAs/AlGaAs 
show current density variations of only a few percent over the same distance. It can also 
be noted that the breakdown voltage of the InAs and InGaAs diodes is low, about 2 
Volt, in comparison with the GaAs diode. For a varactor to be used in practice the 
breakdown voltage needs to be higher since the pump voltage otherwise could destroy 
the diode. One method of increasing the total breakdown voltage is to fabricate several 
varactor diodes in series. Each of n identical diodes would then only need to sustain the 
total pump voltage divided by n. 




0,2 0,4 0,6 0,8 

Voltage (V) 



1,2 1,4 



Fig. 4 Measured current vs. voltage for SBV diodes of size 25*25 jim 2 in different 
material systems. 



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2.2.2 RF measurements 

Two methods have been developed for measuring the capacitance of these diodes, 
the results will be published elsewhere [14]. One method is similar to the one by 
Tolmunen et. al. [15], the other method uses a 60 GHz coplanar probe to contact a mesa 
diode surrounded by a large ring which is contacted by the two ground contacts of the 
probe. 



2.3 MODELLING DEVICES AND MULTIPLIER PERFORMANCE 

2.3.1 CV and IV modelling 

For calculating the CV-, and IV characteristics, the semiconductor transport 
equations including the energy balance equations for holes and electrons are solved 
simultaneously using a finite-difference scheme in a detailed one-dimensional energy 
transport model for GaAs/Al x Gai_ x As heterojunction structures with 0.0 < x < 1.0. 
The computer program is a modified version of the program developed by H. Hjelmgren 
[3] [4]. The program can readily be modified further for other heterostructures. A non- 
uniform grid-mesh is used to obtain the necessary accuracy and acceptable calculation 
times. Position dependent barrier parameters were implemented following the procedure 
of M. S. Lundstrom and RJ, Schuelke [5]. The model used is based on the paper by R. 
Stratton [6]. The five variables are the electric potential, the quasi-Fermi levels for holes 
and electrons, and hole and electron temperatures. The equation system has a 19 
diagonals band-matrix which is solved by LU decomposition. To improve the 
convergence properties of the program the energy balance equation for holes is not used, 
hence the hole temperature is set to the lattice temperature for all applied voltages. Hot 
electrons and lattice heating are not simulated, hence the five equation model is reduced 
to a three equation (Poisson's and current continuity equations) model. Recombination 
is modelled with Shockley Read Hall (SRH) recombination. The relaxation time was set 
to 3.0 ps, the electron life time to 1.0 ns, and the conduction band-offset to 65 % in 
most simulations for which x < 0.45. The contact resistivity of the two ohmic contacts 
can be set to a value of choice. They are modelled by modifying the boundary conditions 
for the potential. The thermoionic emission current, which limits the current through the 
device, is then calculated from the equation for a reverse biased Schottky diode, using a 
voltage dependent barrier-height obtained from the shape of the conduction band. 
Tunneling current is assumed to be sufficiently small and thus neglected altogether. 
The diode capacitance, C= dQ/dV == AQ/AV is determined numerically from the change 
AQ in the stored electric charge in the semiconductor for an incremental change AV in the 
applied voltage. The integration of charges is carried out to the middle of the barrier for 
single barrier devices. To obtain the depletion capacitance one must correct for the 
influence of the series resistance on the voltage across the depletion layer. 



2.3.2 CV characteristics and conversion efficiencies of modelled structures. 

It is of importance to understand how the CV characteristic can be modified by 
changes to the design of the epitaxial structure. It is obvious that increasing the width of 
the barrier will decrease C max , and that a wider L2 region, see Fig. 1, will allow a larger 
capacitance swing, i.e. make C m i n smaller. Below we will investigate how the doping 
profile of the diode affects the efficiency of a 3*60 GHz tripler. The effect of the plasma 
resonance on the series resistance was not included in the multiplier calculations on 
structures #100-103B, #1 12 and #1 14 at 3*60 GHz. 



Third International Symposium on Space Terahertz Technology Page 121 



The following rules are basic to the understanding of the behaviour of the CV 
characteristic for different choices of doping concentrations, viz. 

i . Doping the barrier will increase C max without influencing the capacitance value at 
large biases, i.e. C m in is not much affected. This phenomenon is studied in structures 
#100-103B. However, too many dopants in the barrier and its blocking capability may 
be destroyed. 

ii. A similiar effect, an increase in Cmax, can be obtained by introducing thin regions 
with high doping concentration adjacent to the barrier on each side. One such example is 
#1 12. This scheme may be less damaging to the barrier's blocking capability, but it may 
also affect material quality in a negative way or make symmetrical growth more difficult. 

iii. A reduced doping concentration in the L2 layer where the depletion occurs, will 
cause a more rapid change in the capacitance vs. bias voltage. It will also lead to a larger 
series resistance. The method cannot therefore be recommended as a stand alone 
measure. 

i v . To avoid an increase in the series resistance according to iii., a tapered doping 
concentration in the L2 region will improve the situation. A comparison between #112 
and #1 14 illustrates this phenomenon. 

Please note that the choice of doping profile will also influence the series resistance. An 
advantageous CV characteristic may offer an undesireably large series resistance. 

The effect of doping the barrier on the CV characteristic is illustrated in Fig. 5 which 
show the results for structures #100-103B. The increase in C ma x is as large as 70 % 
when the doping is increased from 1*10*5 cm~3 to 1*10*8 cm'3. To see any effect at 
all, the doping in the barrier needs to exceed 1*10 17 cm"3. The series resistance remains 
the same of course, since it is essentially determined by the doping in region L2. In Fig. 
6 is shown the conversion efficiency vs. input power (all input power is assumed to be 
absorbed). It is interesting to note that the efficiency actually decreases as C ma x 
increases. This is in agreement with the results presented in paragraph 2.1. 
The effect of a high doping concentration near the barrier interface is illustrated by the 
CV characteristic of #1 12 in Fig. 7. Notice that the CV is almost identical to the one of 
#103. Since the series resistances are also very similar, the efficiencies vs. pump power 
are virtually identical as can be seen in Fig. 6 and Fig. 8. 

Structure #114 has a linearly graded doping concentration in the L2 and L4 regions, 
starting with 1*10*5 cm* 3 near the barrier and increasing to 2*10*7 cm" 3 at a distance 
of 385 nm from the barrier. The ratio C ma x/C m in is increased considerably compared to 
that of a homogenously doped L2 region, but the series resistance is larger than for 
#100-103B. The CV resembles mostly that of #103B. The capacitance swings down 
faster, but have the same C m i n and a small reduction in efficiency. 



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Third International Symposium on Space Terahertz Technology 



4.0 10" 14 



^ 



3.0 10" 



14 _.. 



0.0 10° 



1111 



\ 

"•. \ 



\ 








' ■ ■ ■ ■ I 



#100, 10 \im2 

#101, 10 \irr\2 

#102, 10 \im2 

#103, 10 u,m2 

#103B, 10 u.m2 



l i i i i 



2 3 

Voltage [V] 



Fig. 5 Calculated CV characteristics for symmetric GaAs/Alo.44Gao.56As structures 
#100-103, #103B with different barrier doping. L2 = L4 = 390 nm with 
1.0* 10 17 cm- 3 , L3 = 20 nm, for #100: Nd = 1.0* 10 15 cm-3, for #101: 
Nd = 1.0*10 16 cm" 3 , for #102: Nd = 1.0*10 17 cm' 3 , for#103B: 
Nd = 5.0* 10 17 cm" 3 , for #103: Nd = 1.0*10! 8 cm' 3 . 
For all structures LI = L5= 100 nm, Nd = 3.4*10 18 cm' 3 



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# 



o 

c 

U 

c 
o 



> 

c 

a 




20 30 40 50 60 
Absorbed Power [ mW ] 

Fig. 6 Calculated conversion efficiency vs. pump power for structures #100- 103 and 
#103B at 3*60 GHz. It is the devices with undoped or low doped barrier 
which have the highest conversion efficiency. The series resistance Rslo used 
was between 32.9 and 33. 1 ohm (calculated by the simulation program) for all 
five structures. 



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uu 



o 

c 

(X 
U 



4.0 10" 



3.0 10 14 -\ 



2.0 10" 14 H 



1.0 10" 14 -I 



0.0 10 




2 3 

Voltage [ V ] 



Fig. 7 Calculated CV characteristics for symmetric GaAs/Alo.44Gao.56As structures 
for different doping profiles in the L2 region meant to enhance the Cmax/C m i n 
ratio. 

#112 has L2i = L4i = 386 nm with Nd going from 1.0*10 17 cm* 3 to 
0.8*10*7 cm"3 at the barrier interface, i.e. a weak gradient, L22 = L42 = 

4 nm with Nd =1.0*10 17 cm"^ L3 consists of three pans, L3i = L33 = 7 nm 
Nd = 1.0*10 18 cm-3, L32 = 6 nm Nd = 2.0*10 17 cm' 3 . 

#114 has L2i = IA\ = 385 nm with Nd going from 2.0* 10 17 cm" 3 to 
1.0* 10*5 C m"3 at the barrier interface, i.e. a steep gradient, L22 = L42 = 

5 nm with Nd =5.0*10^ cm" 3 , L3 consists of three parts, L3i = L33 = 6 nm 
Nd = 1.0*10 18 cm" 3 , L32 = 8 nm Nd = 5.0*10 16 cm" 3 . 

LI = L5 = 100 nm with Nd = 3.4*10 18 cm" 3 for both structures. 



Third International Symposium on Space Terahertz Technology 



Page 125 



Fig. 8 



* 



u 

c 

.22 
o 

<o 

c 
o 
'5 

> 

e 
o 
U 



20 



15 - 



10 - 






10 20 30 40 50 60 70 
Absorbed Power [ mW ] 

Calculated conversion efficiency vs. pump power for structures #112 and 
#1 14 at 3*60 GHz. The value of R s \ was 33.8 ohm for #1 12 and 44.2 ohm 
for #114. 



3 . A 750 GHz MULTIPLIER DESIGN STUDY 

A study on different 750 GHz multiplier configurations using GaAs and InAs 
SBV-diodes was performed. The aim of the study was to evaluate theoretically the 
optimum frequency multiplier configuration for generating > 50 |J.W of output power at 
750 GHz, assumed to be enough for pumping a 750 GHz SIS mixer, using a Single 
Barrier Varactor (SBV) diode as the nonlinear element. 

Different important diode parameters were investigated, such as the cutoff frequency 
which is a direct function of the doping level Nj of the low doped epilayers, L2 and L4 
in Fig.l, in the device. The cutoff frequency was calculated for GaAs and InAs SBV- 
diodes of three different sizes, see Fig. 9. In these calculations of Cmax and cutoff 
frequencies, it was assumed that the effective barrier width was three times that of L3 
because of the distance necessary for a lineup of the fermi levels at the interface between 
the low doped layer L2 and the undoped barrier L3. It can be seen that the optimum 
doping concentration is l*10^cm"^ and 2*10^cm"3 for GaAs and InAs devices 
respectively, both having a diameter of 2 ^m. 



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X 

S 

o 

c 
<u 

3 



u 
u 



c 
>» 

Q 




Doping concentration Nde, [ cm-3 ] 

Fig 9 Calculated cutoff frequency vs. doping level Nde f° r GaAs and InAs SBV- 
diodes of different diameters a. Nds = 3.4*l()18cm"3 in LI and L5, Ndb = 
l*10 17 cm- 3 in L3, LI = 100 nm, L5 = 100 nm, L2 varies with Nde, the 
barrier width L3 is 20 nm. 

GaAs: (□): a = 2 (lm., (O): a = 4 |im., (A): a = 20 |im. 
InAs: (■): a = 2 um., (•): a = 4 |im., (A): a = 20 |im. 



Another important parameter to consider is the effect of the plasma resonance 
frequency on the series resistance of the diodes [7]. This is of great importance to 
devices intended for THz frequencies. In Fig. 10 is plotted the calculated series 
resistance for InAs SBV-diodes when the plasmaresonance is taken into account . It is 
found that by using InAs instead of GaAs in the low doped epilayers and in the 
substrate, the plasma resonance frequencies are shifted to higher frequencies. This is 

due to the higher electrical conductivity a of InAs compared to that of GaAs. It can be 
seen in Fig. 10 that a doping Nd in L2 and L4 of about 1*10*7 cm" 3 gives sufficient 
clearence up to a frequency of 2 THz 



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10' 




II III I I I I I II I 



10 ( 



Frequency (GHz) 



Fig. 10 Calculated series resistance vs. pump frequency for an InAs SBV-diode for 
different doping levels Nde in the low doped epilayers L2 and L4. Doping 
concentration Nds is 3.4*10*8 cm" 3 in LI and L5, N D is 1*10*7 cm" 3 in the 
barrier L3, LI =100 nm, L5 = 100 nm, L2 = L4 = 533 nm, L3 = 20 nm, 
diameter a = 3.57 jim. 

(•): Nde is 1*10 16 cm'3., (■): Nde is 1*10 17 cm' 3 ., 
(A): Nd e isl*10 18 cm" 3 . 



The avalanche breakdown voltage for the low doped epilayers limits the maximum 
pump power that can be used for the device. Using the optimum doping concentration 
the breakdown voltage was calculated to be 15.7 V for GaAs and 1.2 V for InAs, and 
having depletion lengths of 480 nm and 100 nm respectively. Due to the very small 
breakdown voltage for the low doped InAs epilayers, L2 and L4 in Fig. 1, the doping of 
the epilayers was reduced slightly to a constant value of 1*10*7 cm -3 which gives a 
breakdown voltage of 2 V and a length of 179 nm for the L2 and L4 epilayers. 
The structure chosen for further study in the InAs/AlSb/InAs material system was the 
following: LI = 100 nm with Nd = 3.4*10 18 cm" 3 , L2 = 150 nm with Nd = 1*10 17 
cm-3, L3 = 14 nm with Nd = 5*10 15 cm-3, L4 = L2> L5 > LI. 

Since the barrier length is choosen to be 14 nm thick, the tunneling current may for such 
a thin barrier not be negligable in reality, especially not at room temperature, why a 
thicker barrier may be more appropriate. The total current through the SBV-diode 
consists of (i) the dc -current, mainly thermionic current, and (ii) the displacement 
current due to the rf-voltage variation over the depletion region. Increasing the dc- 
current compared to the displacement current makes the tripler work more in a varistor 
mode with a reduced efficiency, see Fig. 1 1. 



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It is assumed in the simulations that the same IV and CV characteristic calculated for a 
GaAs based SBV-diode can also be used for InAs based diodes, of a similar design. 
This assumption is due to the fact that the difference in conduction current, see Fig.4, 
between InAs and GaAs diodes at a fixed bias voltage, has been found to have a 
negligible influence on the efficiency, see Fig. 11. 




Input power (mW) 



Fig. 1 1 Calculated conversion efficiency versus input power, when varying the 

calculated conductance, dl/dV, for the device in steps of (dl/dV)* 10 v , where 
y is 0, 1, 2 or 3. Nds = 3.4*10 18 cm"3, Nde =1*10 17 cm" 3 , Ndb = 5*10 15 
cm-3, LI = 100 nm, L5 = 100 nm, L2 = L4 = 150 nm, L3 = 14 nm, diameter 
a = 2^m. (■): y=0.0, (□): y=1.0, (O): y=2.0, (A): y=3.0 



Three different multiplier configurations for generating the desired output power of 
> 50 }iW at 750 GHz have been investigated, see Fig. 12. In the calculations was the 
effect on the series resistance from the plasma resonance included. 
Due to the different frequencies for pumping the 750 GHz SBV-diode multipliers, see 
Fig. 12, the expected maximum available input power differs considerably, as shown in 
the figure. It should be observed that 11 mW of output power from a 250 GHz 
Schottky-varactor diode tripler, see Fig. 12, is achieved at operating conditions close to 
the Schottky diode burnout [9]. Thus a more realistic value of a maximum of 6 mW of 
output power was used for the 250 GHz tripler. In the calculations the maximum 
allowed input power was set to such a value that the rf-voltage Vrf over the device, 
minus the rf-voltage over the series resistance, is less than or equal to the breakdown 
voltage Vbr of the low doped epilayer L2. 



Third International Symposium on Space Terahertz Technology Pagel29 



150 GHz 
75 GHz 



Iz 0(2) — l>(rf) > 75 ° GHz 

<160mW ^— * <20mW ^^>0.05mW 
Schottky SBV 

^-^ 250 GHz ^^ 

83.3 GHz 0( X 3 j — fc(x3 1) 750 GHz 

<60mW V-/ <HmW ^~^>0.05mW 

Schottky SBV 



107GHz 0(x7j > 750GHz 

<36mW >0.05mW 

SBV 

Fig. 12 The different 750 GHz multiplier configurations investigated in this study. 



The conversion efficiency for 750 GHz SBV-diode multipliers using SBV-diodes 
based on InAs and GaAs is shown in Fig. 13. Simulations for two different contact 
resistances Re = 8*10"^ ohmem^ and 1*10"^ ohmem^ were used for the GaAs SBV- 
diodes, in order to evaluate what implications the choice of material has on the efficiency 
of the device. The higher electron mobility in InAs reduces the series resistance of the 
diode, thus giving InAs SBV-diodes an advantage in efficiency for low input powers 
over GaAs SBV-diodes, having the same ohmic contact resistance. This advantage in 
efficiency is even larger when a more realistic ohmic contact resistance, Re, of 8* 10" 7 
ohmem^ is used for the GaAs SBV-diodes. The efficiency of the multipliers increases 
with increasing multiplication rate, see Fig. 13, in principle in the same way for the 
different SBV-diodes. This is due to the fact that at lower frequencies the impedance 
(resistance R v and reactance jX v ) of the variable capacitance increases, i.e. becomes 
larger as compared to R s . The efficiency is proportional to R v //(Rv + Rs). see ref.[10]. 
The cutoff frequency is defined in Eq. 1 . 



Page 130 



Third International Symposium on Space Terahertz Technology 



InAs 
Rc=Rco 



10" 

SlO 1 

Il0°t 

'o : GaAs 

§ io^R^Rc 

c 

•g 10 " 2 

> 

o 10 
U 



10 



GaAs 
: Rc=80 Rco 




I I I I I IS 



■ • • ■ ' 



I I ll 



I I I I 1 1 



10 



-2 



10" 1 10° 10 1 

Input power (mW) 



10' 



Fig. 1 3 Calculated conversion efficiency versus input power for various multiplier 

configurations, see Fig. 12, using InAs and GaAs SBV-diodes as the nonlinear 
element Two GaAs SBV-diodes having different contact resistances Re, 
(=Rco in the figure), are shown in the figure where Rco = 1*10~8 ohmem^. 
Nds = 3.4*10 18 cm' 3 , Nde =1*10 17 cm" 3 , N = 5*10 15 cm" 3 , Ll = 
100 nm, L5 = 100 nm, L2 = L4 = 150 nm, L3 = 14 nm, diameter a = 2 |im. 
(O): GaAs SBV-diode tripler, (A): GaAs SBV-diode quintuples 

(□): GaAs SBV-diode heptupler, (•): InAs SBV-diode tripler, 
(A):. InAs SBV-diode quintupler and (B):InAs SBV-diode heptupler. 



Using a larger diameter SBV-diode means of course a lower impedance, although it 
can handle a larger input power before Vrf exceeds the breakdown voltage Vbr- A larger 
diameter device will also withstand larger thermal heating, caused by the absorbed pump 
power, before efficiency degradation occurs due to the reduced mobility of the electrons. 

The mobility ji of the electrons is proportional to the temperature T of the material, 

where for GaAs: |i ~ T 1 / 2 and for InAs: \i ~ T 3 / 2 [1 1][12]. Thus InAs is more sensitive 
to thermal heating. However, since the breakdown voltage for InAs material is small 
compared to GaAs, the level of the absorbed power is restricted from that point of view. 
The influence of the thermal characteristics of the SBV-diode on the efficiency is not 
taken into account in this study. 

The calculated output power, using the values for the efficiency shown in Fig. 13, is 
shown in Fig. 14. The simulated output power for a state of the an 2.8 fF Schottky- 
varactor diode described in [13], is also shown in Fig. 14 where it can be seen that the 
maximum output power for the Schottky-varactor diode clearly exceeds the output 
power for GaAs SBV-diodes having a contact resistance of 8*10"^ ohmem^ (80Rco), 
for the same input power. However, it should be noted that output powers comparable 
to the Schottky-varactor diode tripler are achieved from the InAs SBV-diode multipliers 
at much lower input power, even though the Schottky-varactor diode is capable of 
generating at least twice the output power compared to the InAs SBV-diode before the 
diode voltage exceeds the breakdown voltage for the device. 



Third International Symposium on Space Terahertz Technology 



Page 131 



It should be noted that the calculations for the Schottky-varactor tripler assume optimum 
match at all frequencies for the tripler, i.e. including the idler. For the SBV-diode tripler 
is optimum match only considered for the input and output frequencies. Thus the 
matching conditions are much easier to obtain for the SBV-diode compared to the 
Schottky-varactor diode, in reality giving the SBV-diode an advantage in efficiency 
compared to the Schottky-varactor diode. Also the unavoidable resistive losses at the 
idler frequency in the Schottky-varactor diode tripler gives the SBV-diode tripler an even 
greater advantage. 

The expected losses in the 750 GHz tripler mount is assumed to be less than 10 dB. 
However it can be seen from Fig. 14 that the only realistic SBV-diodes capable of 
generate ^ 500 |iW, necessary for > 50 jiW of output power at 750 GHz are InAs based 
diodes, even though this is achieved close to or at the breakdown voltage V Dr limit for 
the InAs devices. Although the breakdown voltage V Dr and thereby also the maximum 
allowed absorbed input power for the InAs SBV-diodes are much smaller than for the 
GaAs SBV-diodes, it can be compensated for by connecting several InAs/AlSb 
junctions in series, creating a multiple barrier InAs SBV-diode. 



10 x 
10° 
£ 10 1 

s 
^ io' 2 



S. 10 



a- io 
5 io - 5 



10 



Fig. 14 



-6 



I I I I I I I | I I I I I I I I | I III TTTTT- I I I I I I I I 

ordiod 



Schottky-varactor diode 



InAs 
Rc=Rco 

: GaAs 
Rc=Rco 

[GaAs 
Rc=80 Ri 



10 




IO' 1 10 v 

Input power (mW) 



10' 



Calculated output power versus input power for various multiplier config- 
urations, using InAs and GaAs SBV-diodes and a Schottky-varactor diode as 
the nonlinear element in the multiplier. Two GaAs SBV-diodes having differ- 
ent contact resistances Re, (=Rco in the figure), are shown in the figure where 
Rco = 1*10-8 ohmcm2. Nds = 3.4*10 18 cm"3, Nde =l*10l.7,c m • 
Nb = 5*10 15 cm" 3 , LI = 100 nm, L5 = 100 nm, L2 = L4 = 150 nm, 

L3 = 14 nm and diameter a = 2 (im. 

(O): GaAs SBV-diode tripler, (A) 

(□): GaAs SBV-diode heptupler, (•) 

(A): InAs SBV-diode quintuples (■) 



GaAs SBV-diode quintuples 

InAs SBV-diode tripler, 

InAs SBV-diode heptupler, and 



(ffl): Schottky-varactor tripler with Co = 2.8 fF and R s = 20 ohm [13]. 



p a gel32 Third International Symposium on Space Terahertz Technology 



It should also be noted that the GaAs SBV-diode, Re = 8*10 -7 ohmem 2 , could in 
principle be pumped by a much larger input power than 6 mW, used in Fig. 14, before 
the diode voltage exceeds the breakdown voltage Vbr of the device. 
The dotted line in Fig. 14 marks the minimum 50 |i\V output power limit, which has to 
be exceeded, assuming no losses in the mount. 

As can also be seen in Fig. 14, higher order InAs SBV-diode multipliers can be used for 
generating the necessary 500 jiW of output power. The penalty for using a higher order 
multiplier is more resistive losses due to a larger number of idler circuits. 



4. CONCLUSIONS 

Theoretical work on multipliers show a maximum efficiency for a lower Cmax to 
Cmin ratio than expected. This has been shown both in simplified calculations as well as 
in full harmonic balance simulations. These findings have several implications. First, 
there is no need for a thin barrier. Secondly, a wider barrier reduces the current. This 
also make simulations easier. The C vs. V characteristics have been simulated from 
epilayer parameters and the impact of different doping structures on both the CV and the 
efficiency for the multipier has been presented. The examples presented can be used as 
design rules.. In a case study for a 750GHz multiplier an InAs SBV diode outperforms 
a Schottky diode. 



5. ACKNOWLEDGEMENTS 

Mikael Ekenstedt, Dept. of Physics, Chalmers University of Technology for growing 
wafers #1566 and #1567. This work has been supported by ESA/ESTEC under contract 
7898/88/NL/PB(SC) and The Swedish National Board for Industrial and Technical 
Development (NUTEK). 



6. REFERENCES 

[1] Hans Gronqvist, Erik Kollberg, Anders Rydberg, "Quantum-well and quantum- 
barrier diodes for generating submillimeter wave power", Microwave and Optical 
Technology Letters, Vol. 4, No. 1, pp 33-38, 1991. 

[2] E. Kollberg, T. Tolmunen, M. Frerking, J. East, "Current saturation in sub- 
millimeter wave varactors", to be published in IEEE Transactions on.Microwave Theory 
and Techn., May 1992. 

[3] Hans Hjelmgren, "Numerical modelling of hot electrons in n-GaAs Schottky barrier 
diodes", IEEE Trans, on Electron Devices, vol. ED-37, No. 5, pp 1228-1234, May 
1990. 

[4] Hans Hjelmgren, Erik Kollberg, and Lennart Lundgren, "Numerical simulations of 
the capacitance of forward-biased Schottky diodes", Solid-State Electronics Vol. 34, 
No. 6, pp. 587-590, 1991 

[5] M. S. Lundstrom and R.J, Schuelke, "Numerical analysis of heterostructure 
semiconductor devices," IEEE Trans. Electron Dev., vol. ED-30,p.l 151-1 159, 1983. 



Third International Symposium on Space Terahertz Technology Page 133 



[6] R. Stratton, "Diffusion of hot and cold electrons in semiconductor barriers", Phys. 
Rev., vol. 126, pp 2002-2014, 1962. 

[7] K.S. Champlin and G. Eisenstein, "Cutoff frequency of submillimeter schottky- 
barrier diodes," EEEE Trans, on Microwave Theory and Tech., vol. MTT-26, pp. 31 - 
34, 1978. 

[8] P.H. Siegel, A.R.Kerr and W. Hwang, "Topics in the optimization of millimeter- 
wave mixers," NASA Technical Paper 2287, 1984. 

[9] N.R. Erickson, "Very high efficiency frequency tripler for 100-300 GHz," Proc. 
of the 10th Int. Conf. on Infrared and Millimeter Waves, pp. 54-55, 1985. 

[10] P. Penfield Jr. and R.P. Rafuse, " Varactor applications ", Massachusets Institute 
of Technology, Cambridge, Massachusets, USA, The MIT Press, 1962. 

[1 1] S.M. Sze, "Physics of Semiconductors," John Wiley & Sons, pp. 28-29, 1981. 

[12] Landolt-Bornstein, "Numerical data and functional relationships in science and 
technology, Group III: Crystals and solid state physics," Volume 17 Semiconductors, 
Springer- Verlag Berlin, pp.577, 1982.' 

[13] A. Rydberg, B. N. Lyons and U.S. Lidholm, "On the development of a high 
efficiency 750 GHz frequency tripler for THz heterodyne systems," To be published in 
IEEE Trans, on Microwave Theory and Techn., May 1992. 

[14] H. Gronqvist, S. Nilsen, A. Rydberg and E. Kollberg, "Characterizing highly 
efficient millimeter wave Single Barrier Varactor multiplier diodes", to be presented at 
The European Microwave Conference, Helsinki, Finland, 1992 

[15] T. Tolmunen, S. Nilsen, O. Boric, M. Frerking and E. Kollberg, "Accurate 
characterization of varactors with fF capacitance", Conference Digest, 16 th International 
Conference on Infrared and Millimeter Waves, Lausanne, Switzerland, August 26-30 
1991, pp 214-215 



Pagel34 Third International Symposium on Space Terahertz Technology 






Effect of Cooling on the Efficiency 
of Schottky Varactor Frequency 
Multipliers at Millimeter Waves 

Jyrki Louhi 1 , Antti Raisanen 1 , Neal Erickson 2 



1 Helsinki University of Technology, Radio Laboratory, SF-02150 Espoo, Finland 

2 Five College Radio Astronomy Observatory, University of Massachusetts, 
619 Lederle Graduate Research Center, Amherst, MA 01003, USA 



Abstract 

The efficiency of the Schottky diode multiplier can be increased by cooling the 
diode to 77 K. The main reason for better efficiency is the increased mobility of the 
free carriers. Because of that the series resistance decreases and a few dB higher 
efficiency can be expected at low input power levels. At high output frequencies 
and at high power levels the current saturation decreases the efficiency of the 
multiplication. When the diode is cooled the maximum current of the diode 
increases and much more output power can be expected. There are also slight 
changes in the I — V characteristic and in the diode junction capacitance, but 
they have a negligible effect on the efficiency of the multiplier. 

1 Introduction 

It is well known, that cooling a Schottky diode mixer improves its sensitivity, i.e. 
reduces the mixer noise temperature. This is mainly due to the sharper I — V 
characteristic at cryogenic temperatures, only partly due to the smaller series 
resistance and lower metal losses in the mixer mount. In satellite applications 
the heterodyne receiver is readily cooled passively to temperatures of 110. ..150 
K. Also, a space qualified 80 K cooler is available. This makes it very reasonable 
to consider the effect of cooling on the frequency multiplier performance. This is 
especially important at submillimeter waves, where not enough power is available 
from ordinary all-solid-state frequency multipliers. 



Third International Symposium on Space Terahertz Technology 



Page 135 



2 Diode model and effect of cooling 

A simple equivalent circuit of the Schottky diode contains three components: non- 
linear junction capacitance, nonlinear junction conductance and series impedance 

[1]. 



WW 




Ci (V) 



V 

Figure 1: Simple equivalent circuit of the Schottky diode. 

Capacitance 

The basic model for the junction capacitance of the Schottky diode is 

Co 



Cj(V) = 



y/l-V/fa' 



(1) 



where <£« is the built-in potential (about 1 V) and Co is diode capacitance, when 
the voltage over the junction is zero. For very small submillimeter wave diodes 
the edge effect must be included in the diode model as [2] 



C 0) . ^.(i+'-OT 



w 



(V) 



2-r a 



w(V) = 




\q-N D 



(2) 
(3) 



where A is the anode area, e, is the dielectric constant of the semiconductor, 
w(V) is the length of the depletion region, r is the anode radius, q is the charge 
of an electron, No is the doping density in the semiconductor, k is Boltzmann's 
constant and T is the temperature. In these models the junction capacitance is 
very high near the contact potential <£w Physically this is impossible, and a bet- 
ter model for junction capacitance must be calculated by using the drift-diffusion 
model [3]. In any of the models, the primary mechanism for the efficiency of the 
multiplier, the degree of capacitance nonlinearity, is not temperature dependent. 
Thus, cooling has no effect on the diode's inherent capability to generate har- 
monics. In the two simple models the only temperature dependent factor is 0^. 



Page 136 



Third International Symposium on Space Terahertz Technology 



When the diode is cooled from 300 K to 77 K, the contact potential fa increases 
by about 0.1 V [4]. Because fa varies only slightly when the diode is cooled, the 
same operation point can be reached if the bias potential Vfcuw is also increased 
as much as fa. In all, the effect of cooling on the junction capacitance is so small 
that it has an almost negligible effect on the multiplier efficiency. 



U. 



o 
o 

c 

(0 

*> 

+-t 

o 
to 

a 

10 

o 




0.6 0.8 1 

Voltage [V] 



1.2 



Figure 2: The junction capacitance at temperatures 300 K (solid line) and at 77 
K (dashed line). 

Series impedance 

When the nonlinearity of the epitaxial layer above the plasma resonance is not in- 
cluded, the series impedance of the submillimeter wave Schottky diode is modeled 

as [5] 



Z.(u) = Z*(v) + Z^{u) + Z. kin (u) + R e 



1 



Z^{u>) = 



1 + ;u// u, » u d 

+ ] — 
4-r [l+ju//u/, w d 



_ P**b 



1 



w.) - ^.^-.^y/yzg^, 



(4) 
(5) 

(6) 
(7) 



where R* is the contact resistance (about 1 fi), p is resistivity, t e (e//) is t e — u>(Vw«), 
t e is the thickness of the epitaxial layer, b is the radius of the chip and 8 t is the 
skin depth in the substrate given by 






(8) 



Third International Symposium on Space Terahertz Technology Pagel37 



where fi is permeability of GaAs. Scattering frequency u t and dielectric relax- 
ation frequency uj are 

"> = -r~, (9) 

«- = ~V' ( 10 ) 

where m* is the effective carrier mass and \i, is the carrier mobility. The resistivity 
is 

P=— ~ , (11) 

q-n-n, 

where n is the concentration of the free electrons in the conduction band. 

In a semiconductor, the concentration of the free carriers n and the mobility of the 
carriers /z, are the most important temperature dependent factors in equations 
given above. In GaAs the donor binding energy Ed is so small and the concen- 
tration of donors Nd is usually so high that the concentration of the free carriers 
n is equivalent to Nd at all temperatures, where the diode should be used. At 
a very cold temperature, below 10 K, the concentration drops, because there is 
not enough thermal energy to ionize electrons to the conduction band, and so the 
resistivity of GaAs becomes high. At very high temperatures the concentration 
of the intrinsic carriers is higher than Nd, and thus n is also higher than No- 

In GaAs the mobility of the free carriers can be calculated from the mobilities of 
the various scattering processes by using the Matthiessen rule 

i = E^ (12) 

/*. Hi 

In GaAs the most important scattering processes are the ionized impurity scatter- 
ing, acoustic-mode scattering and polar-optical scattering. At room temperature, 
the polar-optical scattering dominates. When GaAs is cooled, the mobility in- 
creases until the mobility of the polar-optical scattering and the mobility of the 
impurity scattering are equal. At that temperature, mobility \i, has a maximum, 
and when the diode is cooled more the mobility decreases. When Nd is rather 
low (1 • 10 16 cm -3 ) the optimum temperature is low (~ 50 K) and the mobility 
greatly increases [6]. At very high doping concentration (2-10 17 cm~ 3 ) the optimum 
temperature is higher (~ 150 K) and the mobility increases only a little when the 
diode is cooled to 77 K. 

When considering the effect of cooling on the series impedance of the Schottky 
diode, it is simplest to consider first its effect on the DC resistance and then the 
effect on the series impedance at high frequencies. When the diode is cooled to 
77 K, the mobility of electrons increases and thus the resistivity of the epitaxial 
layer decreases, which also decreases the DC resistance of the diode. When the 
doping concentration of the epitaxial layer is low, the DC resistance decreases 
significantly. (For diode UVA 6P2 the measured decrease is about 4.5 fi, from a 



Page 138 Third International Symposium on Space Terahertz Technology 



value of 10.5 ft to 6 ft; the calculated values agree very well, see Figure 3) When 
the doping rate is higher the decrease of the resistance is not as large. (For diode 
UVA 2T2 the calculated decrease is about 3.5 ft, from a value of 12 ft to 8.5 ft) 
When considering the effect of the decreased series resistance on the efficiency of 
the multiplication, it must be noticed that the resistance of the epitaxial layer Z ep i 
is a function of the thickness of the layer. In an efficient reactive multiplication, 
the voltage over the depletion region spends a substantial part of the pump cycle 
in the low voltage region, where the contribution of Z^ in Z, is large, but a small 
part of the pump cycle in the high reverse voltage region, where the contribution 
of Z^ in Z, is small. When the diode is now cooled, the decrease of the series 
resistance is smaller than the decrease of the DC resistance, but still the decrease 
of resistance has a very strong positive effect on the efficiency of multiplication. 

At high frequencies the series impedance of the Schottky diode is no longer purely 
resistive, because of the plasma resonance and the skin effect. When the diode is 
cooled, the plasma resonance frequency 




w p = ,/u t -u; d = W — (13) 



does not change, because it is independent of the electron mobility fi t . Because u>, 
and u>4 are temperature dependent, the Q-factor of the resonance is also temper- 
ature dependent, and when the diode is cooled to 77 K the Q is increased (Figure 
3). Because the mobility in the substrate changes only very little when the diode 
is cooled, the impedance of the substrate Zmi, and the impedance of the skin effect 
are not changed significantly. 



Third International Symposium on Space Terahertz Technology 



Page 139 



10 a E 



10 a 



CD 
CO 



l i 



T 1 — I I I I I I 



I I I 




J I 1 I I 1 1 1 1 I I I I 1 I I 1 



0" 



10 « 

Frequency [Hz] 



10 



13 



Figure 3: The series resistance of diode UVA 6P2 at temperatures 300 K (solid 
line) and 77 K (dashed line). 



& 



i 1 — i — i i m i r— t — i — i i i t f 




10 ia 
Frequency [HzJ 



10" 



[h P 



Figure 4: The series reactance of diode UVA 6P2 at temperatures 300 K (solid 
line) and 77 K (dashed line). 



Page 140 



Third International Symposium on Space Terahertz Technology 



I-V characteristic 

For a Schottky diode the I — V characteristic is assumed to be [4] 

B = flycoth^) 



q± I N D 



(14) 
(15) 

(16) 



where R** is modified Richardson's constant, h is h/2ir and h is Planck's constant. 

There are two important factors of the I — V characteristic for the efficiency of the 
frequency multiplication: the turn-up point of the I — V curve, and the steepness 
of the I — V curve beyond that. When the Schottky diode is cooled, the possible 
voltage range where the multiplication is mainly reactive, increases, and thus the 
maximum efficiency can also increase. For a cooled diode the shape of the I — V 
characteristic is also sharper, and therefore the resistive multiplication is slightly 
more effective. 



XlQ- 6 



C 
<D 

C 
C 
3 
U 




0.5 
Voltage [V] 



1.5 



Figure 5: The current- voltage characteristic at temperatures of 300 K (solid line) 
and 77 K (dashed line). 



Third International Symposium on Space Terahertz Technology Page 141 



Current saturation 

At a low electric field the electron drift velocity vj is directly related to the electric 
field £ as 

v d = H. • S. (17) 

When the electric field increases the drift velocity also increases until the velocity 
reaches a maximum value v mox (= 2.2 -lO 5 m/s at about 3.2 kV/cm in an intrinsic 
case). In that situation the electron conduction current 

i e = A ■ n - q ■ fi t • £ (18) 

must be replaced by the maximum current 

imax = A-n-q-Vna*. (19) 

This current saturation causes a very significant decrease in the efficiency of the 
multiplier at high power levels and also when the output frequency is high, because 
the junction capacitance cannot be pumped with optimum current. The current 
saturation seems to be the most important factor for a submillimeter wave fre- 
quency multiplier, when the efficiency of the multiplication is considered. When 
the diode is cooled, the maximum drift velocity increases [6] and because of that 
the maximum electron current also increases. Therefore, when the diode is cooled 
the effect of the current saturation is less significant. This increases the efficien- 
cy especially at high power levels^ at high frequencies, and in the case of a high 
multiplication factor. 

The current saturation may be modelled by strongly current dependent series 
resistance R,(i) above the maximum current. Kollberg et al. have presented the 
following model [7]: 

R.{i) = R,(DC) • a • i 6 , (20) 

where a is a parameter, depending on the maximum current of the diode i max . The 
meaning of the R,(i) is to modify the current waveform approximately as required 
by causing a very strong increase in the series resistance when the current of the 
diode is higher than the maximum current i max . The parameter a has been fitted 
empirically to the measured results only in one case and must be estimated for 
other diodes and frequencies. There seems to be no physical background for this 
model, but so far no better model has been proposed. 

3 Analysis of multipliers 

At millimeter waves Schottky varactors are often driven into conduction, which 
is only nearly optimal. In this case, the usefulness of classical theories [8] is poor 
and harmonic balance analysis [9] should be used. One form of the harmonic 
balance analysis is the multiple reflections technique, where the multiplier circuit 
is divided into linear and nonlinear subcircuits, which are then analyzed in the 
frequency and time domain. 



Page 142 



Third International Symposium on Space Terahertz Technology 



Doubler for 160 GHz 

Let us first consider the effect of cooling on a two diode balanced doubler for 160 
GHz, because we have also experimental results for it [10]. 

Table 1: Parameters used for UVA 6P2. 





Co 


A 


t. 


N D 


M 


^max 


300 K 


21 


33 


1.0 


3.5-10 16 


0.61 


44 


77 K 


21 


33 


1.0 


3.5-10 16 


1.40 


66 




fF 


fiva. 2 


fiva. 


cm" 3 


m/s 


mA 



The two diode construction has been analyzed both at 300 K and at 77 K. First, 
the efficiency has been calculated with optimum embedding impedances. These 
results have been plotted in Figure 6 (solid lines). Here the efficiency has only a 
very poor correlation with the measurements (0 and X) because of the VSWR, 
which is mainly caused by the fact that the embedding impedances are optimized 
for high input power. When the doubler is then analyzed by using the optimum 
embedding impedances for high input power at all input power levels, the corre- 
lation is much better, especially when 0.5 dB losses in the input and 0.8 dB losses 
in the output have been taken into account. 



Table 2: Experimental output power versus temperature and input power (two 
diodes). 







Input power 






10 


33 


50 


100 


mW 


Temperature 


300 K 


1.6 


9.0 


13.9 


22.0 


mW 


223 K 


1.9 


10.4 


16.3 


26.7 


mW 


77 K 


2.2 


12.8 


18.7 


30.7 


mW 



In order to understand better the agreement between the theory and experiment, it 
is worth separating the effects of the decreased series impedance and the increased 
current handling capability due to the cooling. First, if the current saturation is 
omitted in the theoretical analysis, the effect of cooling is as follows. At low 
input power levels when the multiplication is purely reactive, the decreased series 
impedance causes a clear increase in the efficiency due to smaller losses in the 
series impedance. According to simulations, the increase of the efficiency in the 
above case at low input power levels is about 1.5 dB. However, when the input 
power per diode is large (i.e. > 10 mW), the multiplication efficiency tends to 
decrease with the increased input power due to the resistive multiplication. This 



Third International Symposium on Space Terahertz Technology 



Page 143 



u 

c 



tu 



80 

70 
60 
50 
40 
30 
20 
10 





-1 — 


: 1 


1 — 






_ 










_ 


- jT 










- 


1 / ' 
/ / 

- / / /< 

II' 

J, ' 


o o 




X 




- 


1 












- 










- 




_1 


-- J 1 


» 







10 20 30 40 
Input power per diode [mW] 



50 



Figure 6: The efficiency of the 160 GHz doubler at 300 K and at 77 K (above), 
when using optimum impedances (solid line) and impedances optimum for high 
power (dashed line). Measurement results, when 0.5 dB input losses and 0.8 dB 
output losses have been taken into account, at 300 K (o) and at 77 K (x) have 
also been plotted. 

is because the voltage swing reaches the conduction region during every cycle. 
The smaller the series impedance, the lower the input power needed to reach this 
conduction, and thus, resistive multiplication. Therefore, the gain due to the 
smaller series impedance is smaller at high input power levels than at low power 
levels. According to the simulations, the efficiency increase due to the smaller 
series impedance in the multiplier described above is only 0.5 dB at 50 mW input 
power per diode. 

When the current saturation is taken into account, but not the series resistance, 
the positive effect of cooling is seen only at high power levels. This is because 
the junction capacitance can be pumped at 77 K more effectively than at 300 K. 
At small power levels the saturation, of course, does not play an important role. 
According to our simulations, the higher current handling capability of the cooled 
diode 6P2 improves the efficiency by 1 dB at 50 mW input power per diode. 

These two effects of cooling together, the decreased series impedance and the 
increased current handling capability, explain the experimentally verified 1.5 dB 
increase in the multiplication efficiency of all power levels and therefore give some 
kind of a proof of the current saturation in the diode at high input power levels. 
Due to the higher efficiency at high input power levels, the maximum output power 
is also increased by the same amount, which helps in pumping the following stage 
in the multiplier chain producing submillimeter wave frequencies. 



Page 144 Third International Symposium on Space Terahertz Technology 



Multipliers for 1 THz 

When constructing multiplier chains for 1 THz, a reasonable choice is first to 
double the output frequency of a powerful W-band Gunn oscillator and then to 
follow by a tripler and a doubler or by a doubler and a tripler. The latter choice 
does not only depend on the varactor diodes but also on the technology to build 
fine mechanical multiplier mounts. 

In order to get some understanding of how much power could be available at 
1 THz, the choice of a tripler to 500 GHz and doubler to 1 THz has been made 
because this allows comparison with experimental results up to 500 GHz [10]. The 
tripler for 500 GHz and the doubler for 1 THz can be analyzed the same way as 
the doubler for 160 GHz, but now current saturation plays a very important role. 
Because the presented model for current saturation has only poor correlation to 
the physics, the results for high frequency multipliers should be considered only 
qualitatively. 

Some general aspects can still be presented. First, when the diode is cooled, 
the maximum drift velocity increases, which also increases the maximum current. 
Second, when the output frequency is high, the changes during the voltage swing 
are very fast. In that situation, the current needed for optimum multiplication 
is very high, and then current saturation plays a very important role by greatly 
decreasing the efficiency. By cooling, the maximum current should increase and 
the efficiency of the multiplication may increase by a few dB. Third, when the first 
or second stage multiplier is cooled, the maximum input power for the last stage 
multiplier increases, and so also the maximum output power for 1 THz increases. 
Our simulations have indicated an increase of about 7 dB from 100 /iW to 500 
/xW in the optimum situation. 

4 Conclusions 

Cooling of a Schottky varactor multiplier increases its efficiency by as much as 
a few dB. Because of the smaller series impedance the efficiency of frequency 
multiplication increases by 1-2 dB at small input power levels. At large input 
power levels the efficiency increases by 2-10 dB due to the higher current handling 
capability of the diode. A cooled multiplier can be readily used in satellite ap- 
plications, where the receiver is cooled to 50. . .150 K. The positive effect of the 
cooling should be utilized especially in submillimeter wave multipliers when the 
output power necessary cannot be reached in any other way. 

Even though the model of a Schottky diode is already rather complex, it should 
be studied more. The main reason for a poor model is that current saturation is 
poorly handled. Much more work must be done to model the saturation exactly. 
Also much more experimental work on cooling multipliers is needed, before all the 
effects of the cooling can be understood. 



Third International Symposium on Space Terahertz Technology 



Page 145 



References 



[1] Raisanen A.V., Sironen M.: Capability of Schottky-diode multipliers as local 
oscillators at 1 THz. Microwave and Optical Technology Letters, vol. 4, no. 
1, 1991, p. 29-33. 

[2] Copeland J.A.: Diode edge effect on doping-profile measurements. IEEE 
Transactions on Electron Devices, vol. ED-17, no. 5, 1970, p. 401-407. 

[3] Hjelmgren H., Kollberg E., Lundgren L.: Numerical simulations of the ca- 
pacitance of forward-biased Schottky-diodes. Solid-State Electronics, vol. 34, 
no. 6, 1991, p. 587-590. 

[4] Kollberg E.L., Zirath H., Jelenski A., Temperature-variable characteristics 
and noise in metal-semiconductor junctions. IEEE Transactions on Mic- 
rowave Theory and Techiques, vol. MTT-34, no. 9, 1986, p. 913-922. 

[5] Crowe T.W.: GaAs Schottky barrier mixer diodes for the frequency range 
1-10 THz. International Journal of Infrared and Millimeter Waves, vol. 10, 
no. 7, 1989, p. 765-777. 

[6] Ruch J.G., Fawcett W.: Temperature dependence of the transport properties 
of Gallium Arsenide determined by a Monte Carlo method. Journal of Applied 
Physics, vol. 41, no. 9, 1970, p. 3843-3849. 

[7] Kollberg E., Tolmunen T., Frerking M., East J.: Current saturation in sub- 
millimeter wave var actors. Proceedings of the 2nd International Symposium 
on Space Terahertz Technology, 1991, p. 306-322. 

[8] Penfield P., Rafuse R.P.: Varactor Applications, Cambridge, Mass., The MIT 
Press, 1962. 

[9] Siegel P.H., Kenr A.R., Hwang W.: Topics in the Optimization of Millimeter 
- Wave Mixers, NASA Technical Paper 2287, 1984. 

[10] Erickson N.: High efficiency submillimeter frequency multipliers. IEEE MTT- 
S International Microwave Symposium Digest, vol III, Dallas, 1990, p. 1301- 
1304. 



Page 146 Third International Symposium on Space Terahertz Technology 



-5/3.-23 



Superlattice Barrier Varactors* 

C. Raman, J. P. Sun, W. L. Chen, G. Munns, 

J. East and G. Haddad 

Solid State Electronics Laboratory 

University of Michigan, Ann Arbor, Michigan 

Abstract 

SBV (Single Barrier Varactor) diodes have been proposed as alternatives to Schottky 
barrier diodes for harmonic multiplier applications. However these show a higher current 
than expected. The excess current is due to X valley transport in the barrier. We will 
present experimental results showing that the use of a superlattice barrier and doping 
spikes in the GaAs depletion regions on either side of the barrier can reduce the excess 
current and improve the control of the capacitance vs. voltage characteristic. 

The experimental results consist of data taken from two types of device structures. 
The first test structure was used to study the performance of AlAs/GaAs superlattice 
barriers. The wafer was fabricated into 90 micron diameter mesa diodes and the resulting 
current vs. voltage characteristics were measured. A 10 period superlattice structure 
with a total thickness of approximately 400 A worked well as an electron barrier. The 
structure had a current density of about one A/cm 2 at one volt at room temperature. The 
capacitance variation of these structures was small because of the design of the GaAs 
cladding layers. The second test structure was used to study cladding layer designs. 
These wafers were InGaAs and InAlAs layers lattice matched to an InP substrate. The 
layers have n + doping spikes near the barrier to increase the zero bias capacitance and 
control the shape of the capacitance vs. voltage characteristic. These structures have a 
capacitance ratio of 5:1 and an abrupt change from maximum to minimum capacitance. 
The measurements were made at 80 K. Based on the information obtained from these two 
structures, we have designed a structure that combines the low current density barrier 
with the improved cladding layers. The capacitance and current-voltage characteristics 
from this structure are presented. 



- 'This work was supported by the Center for Space Terahertz Technology under NASA Contract No. 
N AG W- 1334 and by the URI-ARO Program Contract No. DAAL03-87-K-0007. 



Third International Symposium on Space Terahertz Technology Page 147 



Introduction 

Varactor diodes are an important component of harmonic multipliers operating above 
100 GHz. These multipliers are the primary source of power in the submillimeter wave 
frequency range, where the diode predominantly used is a Schottky barrier device. How- 
ever, a multiplier based on Schottky diodes suffers certain disadvantages. The circuit is 
complex, with higher order conversion requiring matching at all frequencies-the input, 
the output, as well as idler frequencies. The varactor must also be biased to increase the 
voltage swing and prevent current flow over the barrier in the forward direction. These 
problems can be overcome by using heteroj unction based varactor structures. Implemen- 
tation of such novel diode structures is motivated by the symmetry of their capacitance 
vs. voltage characteristic which permits odd harmonic conversion without the added 
complexity of even harmonic idlers and bias circuitry . 

The Single Barrier Varactor 

One possible structure is the single barrier varactor, shown in Figure la. The cor- 
responding energy band diagrams under thermal equilibrium and bias are shown in lb 
and lc. The analogue of the metal-semiconductor electron energy barrier found in the 
Schottky varactor diode is the conduction band offset between the GaAs region and the 
wider bandgap Al x Ga 1 _ r As layer. This energy barrier acts to inhibit electron transport 
through the structure and ideally should be as large as possible for the device impedance 
to be purely reactive. When one side is biased relative to the other, the GaAs region 
on the anode side is depleted of electrons and becomes positively charged, similar to the 
semiconductor region of a Schottky varactor under reverse bias. Electrons accumulate 
on the cathode side of the structure, forming a charge separation across the Al^Ga^^As 
region. The relationship between the stored charge and the applied voltage is non-linear, 
resulting in a non-constant device capacitance. Moreover, since the structure is symmet- 
ric, reversing the sign of the applied voltage merely interchanges the roles of depletion 
and accumulation regions and does not affect the device capacitance. Therefore the 
capacitance-voltage characteristic has even symmetry. Early attempts at implementing 
such varactors were limited in efficiency by high leakage current densities 1 . Details of 
the leakage current analysis are described elsewhere in these proceedings 2 . 

By replacing the single heterostructure by a superlattice barrier, we propose to reduce 
the carrier transport through the device by increasing the effective barrier height seen 



Anders Rydberg, Hans Gronqvist, and Erik Kollberg, Millimeter and Sub- Millimeter Wave Multi- 
pliers Using Quantum Barrier Varactor (QBV) Diodes, IEEE Electron Device Letters, Vol. 11, No. 9. 
Sept. 1990, pp. 373-375. 

2 H. Hjelmgren, J. East and E. Kollberg, "Thermionic Emission Current in a Single Barrier Varactor,", 
these proceedings. 



Page 148 Third International Symposium on Space Terahertz Technology 



by an electron. A doping profile modification can improve the capacitance vs. voltage 
characteristic; however, a trade-off is seen between maintaining a high barrier height and 
a good C-V profile. 

The Superlattice Barrier Varactor 

Although the SBV presents a symmetric C-V characteristic, its efficiency is degraded 
due to the high leakage current associated with the device because a purely reactive 
multiplier is more efficient than a resistive one. The true thermionic emission energy 
barrier seen by an electron in the GaAs regions is not the T to T energy level offset but 
the considerably lower T to X energy level difference, as seen in Figure 2a. By means of 
a scattering process an electron incident on the barrier can pass through the Al x Gai_ x As 
region into the X valley of the barrier material. Thermionic emission over this X level 
requires less kinetic energy than emission over the T level and consequently a smaller 
bias voltage is needed to turn on the current. 

The X valley transport can be suppressed by replacing the single heterojunction 
by a series of barriers interspersed with quantum wells, i.e., a superlattice. Such a 
superlattice appears in Figure 2b. The well regions are thin, resulting in quantum 
mechanical confinement. The energy mini-bands are shifted upward considerably with 
respect to the T point in bulk GaAs. A simple calculation indicates that the energy 
increase can be on the order of an electron volt. Unlike the SBV, there is no longer a 
continuous X valley current path since the well energies are higher than the X levels. 
The superlattice structure achieves a larger effective barrier than that of the SBV and 
suppresses the leakage current. 

A thin, highly doped layer between the superlattice and each adjacent N~ region can 
be used to modify the capacitance. The so-called 5-doped regions (see Figure 3) are thin 
and contain a large amount of charge so that at zero bias they remain mostly undepleted. 
Flat band effects are minimized and the zero bias capacitance is increased. Varying the 
bias slightly causes almost no change in the depletion width and the capacitance remains 
constant. However, beyond a threshold voltage, the entire 8 region becomes depleted 
and the lightly doped N~ region begins to deplete rapidly, causing a sharp drop in the 
capacitance from its zero bias value. 

Advantages and Disadvantages 

In designing superlattice varactors, two material systems were considered: GaAs/AlAs 
and InGaAs/InAlAs lattice matched to InP. Structure I, seen in Figure 4, shows a 10 pe- 
riod, 20 A / 20 A GaAs/ Al As superlattice barrier varactor chosen for examination of the 
barrier properties. The MBE grown wafer was processed on the front side by photolithog- 
raphy. Contacts were made to the front and back by evaporating a Ni/Ge/Au/Ti/Au 



Third International Symposium on Space Terahertz Technology Page 149 



sequence and annealing at 405 degrees Celsius. Mesa diameters of 5 to 90 //m were then 
chemically etched. The measured room temperature I-V curve in Figure 5 demonstrates 
the effectiveness of the superlattice in keeping the current to a minimum. At a bias of 
1 volt the current density is about 1 A/cm 2 , compared with about 150 A/cm 2 for the 
•single barrier varactor. However, the ratio of maximum to minimum capacitance is 
insufficient for any significant harmonic conversion. 

The second structure, a 40 A / 40 A InGaAs/InAlAs superlattice, was designed with 
doping spikes added, as outlined in Figure 6a. Figure 6b shows the C-V data, taken at 
SO K to reduce the parallel conduction current 3 . A close agreement is observed with the 
capacitance characteristic predicted by a self-consistent quantum mechanical and Poisson 
solution for the charge and potential distribution throughout the superlattice varactor. 
When a voltage of approximately 0.2 volts is applied, the capacitance drops sharply to 
about one-fifth of its zero bias value, corroborating the theoretical expectation. Thus 
doping profile modifications allow good control of the capacitance-voltage characteristic. 

The final structure utilized the low leakage properties of the GaAs/AlAs superlattice 
and incorporated the doping modifications that had been tested in the InGaAs/InAlAs 
system. In structure III the same superlattice as Structure I (with one well layer removed 
from the end for symmetry) was grown on a new varactor wafer which incorporated a 
doping spike of sheet density lxlO 12 cm -2 and a more lightly doped N" region than the 
previous GaAs/AlAs wafer. Figures 7 and 8 present the room temperature I-V and C-V 
data from 90 micron diameter mesas fabricated on this wafer. Compared with structure 
I, the leakage current density has increased. This current degradation can be attributed 
to the following mechanism: the effective electron thermionic emission barrier is reduced 
by band bending at the barrier edge due to the high electron density in the doping spike. 
The capacitance and parallel device conductance were simultaneously measured on a 
HP4275A LCR meter, where the bias was varied until the conductance exceeded the 
measurement capability of the machine. As seen in Figure 8, the C-V profile exhibits a 
swing ratio of close to 4. However, the zero bias capacitance was larger than the value 
predicted by the quantum mechanical analysis by about a factor of 2. The reason for 
this is not clearly understood at the present time. 

Despite the fact that the current levels in the final structure were higher than ex- 
pected, the superlattice varactor represents improvement over the single barrier varactor. 
Figure 9 compares the current densities for the two devices at low and high voltages and 
temperatures. For operation at low temperature or for small voltages the superlattice 
barrier fares better than the SBV. 



3 J. P. Sun, W. L. Chen, J. East and G. Haddad, C-V Characteristics of Quantum Well Varactors, 
Proceedings of the 1991 International Semiconductor Devices Research Symposium. 



Page 150 Third International Symposium on Space Terahertz Technology 



Conclusions 

We have demonstrated a superlattice barrier varactor in the GaAs/AlAs system which 
promises to be useful in odd harmonic generation due to its reduced leakage properties 
when compared with the SBV. Moreover, modification of the capacitance by means of 
an appropriate doping profile has been demonstrated in the InGaAs/InAlAs material 
system. A compromise appears, however, between minimizing leakage current levels in 
the device and achieving a large maximum to minimum capacitance swing ratio. We 
hope to begin RF testing of the superlattice barrier varactors in the near future. 



Third International Symposium on Space Terahertz Technology 



Page 151 



a) 



N + GaAs 



yV" GaAs 



N~ GaAs 



T 
i Al x Gai_ T As 



N+ GaAs 



b) 



c) 



+ + + 
+ + + 



r 



FIGURE (1): (a) Single Barrier Varactor , (b) band diagram under 
equilibrium, (c) band diagram under bias. 



Page 152 



Third International Symposium on Space Terahertz Technology 



a) 



Gamma 
Level 



X 
Level 



Electron sees 
lower barrier 




Electron 



Confined Gamma Energy Levels 
In The Well Regions 



b) 



X 



X 



X 




Electron 



FIGURE (2): (a) X valley transport in Single Barrier Varactor, 
(b) Suppression of X transport by superlattice. 



Third International Symposium on Space Terahertz Technology 



Page 153 




SPACER 
LAYER 



SUPER- 
LATTICE 



DOPING 

SPIKE 

N+ 



FIGURE (3): Superlattice with doping spikes 



3000 A 

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Ten Period 
Superlattice 



FIGURE (4): Structure I: GaAs/ Al As superlattice varactor 



Page 154 



Third International Symposium on Space Terahertz Technology 



xxxxxx 



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Third International Symposium on Space Terahertz Technology 



Page 155 



4000 A 



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Voltage (V) 



FIGURE (6): (a) Structure II: InGaAs/InAlAs superlattice varactor with 
doping spikes, (b) Structure II: G-V characteristic 



Page 156 



Third International Symposium on Space Terahertz Technology 



SUFERUTTICE IV (DELTA-DOPED): 90 MICRONS 



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GAAS/AUS DEOA-OOFED S/t C-V: SO MICRON MESA DIAMETER 



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Third International Symposium on Space Terahertz Technology 



Page 157 



J (Amps/sqcm) 





At V = 0.1 volts 


At V = 0.4 volts 


SBV 


1.1 @300K 
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FIGURE (9): Comparison of improvements 



Page 158 Third International Symposium on Space Terahertz Technology 

A NEW FABRICATION TECHNIQUE FOR BACK-TO-BACK 
S/3-33 . VARAC^RDZODES N93-277 3 9 

/ (& &£)*t / ^- P eter Smith, Debabani Choudhury, Suzanne Martin, and M.A. Frerking, 

John K. Liu, and Frank A. Grunthaner 



.V 



Center for Space Microelectronic Technology 

Jet Propulsion Laboratory/California Institute of Technology 

4800 Oak Grove Drive 

Pasadena, CA 91109 



Abstract: A new varactor diode process has been developed in which much of the 
processing is done from the back of an extremely thin semiconductor wafer laminated to a 
low-dielectric substrate. Back-to-back BNN diodes were fabricated with this technique; 
excellent DC and low-frequecy capacitance measurements were obtained. Advantages of 
the new technique relative to other techniques include greatly reduced frontside wafer 
damage from exposure to process chemicals, improved capability to integrate devices 
(e.g., for antenna patterns, transmission lines, or wafer-scale grids), and higher line yield. 
BNN diodes fabricated with this technique exhibit approximately the expected capacitance- 
voltage characteristics while showing leakage currents under 10 mA at voltages three 
times that needed to deplete the varactor. This leakage is many orders of magnitude better 
than comparable Schottky diodes. 



Introduction: 

Planar varactor diodes * are being developed in place of whisker-contacted devices in order 
to improve the performance and ruggedness of spacebome submillimeter-wave heterodyne 
receivers; at the same time thin heterostructure layers are being used to improve diode 
performance, making processing more demanding. Such devices could be more useful if 
integrated into relatively large arrays that could potentially be used for communications 
systems. It is expected that such back-to-back multiplier diodes with Schottky contacts and 
heterostructure barriers can be made to operate reasonably efficiently at frequencies over 
one terahertz2>3,4 j^ m i s paper we present results from devices in which the isolation 
was performed from the back. This technique simplifies processing, greatly increasing 
yield and providing a much lower dielectric constant, and thus low loss, environment for 
antennas and other circuit patterns. 

Conventional multiplier diode isolation techniques have a number of problem areas. 
Isolation implants are commonly used, but the removal of masking materials from the 
wafer often presents difficulties. An alternative technique for isolating the active devices is 
to perform a mesa etch, but connection of the contacts to the top of the mesa is problematic. 
One approach attempted here requires relatively difficult planarization processing that can 
potentially damage the thin barrier layer. Metal step coverage may be also a problem with 
this approach. Air bridging can also be used with mesa isolation, but this also exposes the 
top of the semiconductor material to a larger number of process steps. Some inactive 
semiconductor material, with its associated high dielectric constant, is generally left in place 
with all of these approaches. 

We have developed an alternative processing technique that promises to be simpler and 
more robust. In this process, no inactive semiconductor material is left, and the front of the 
wafer is exposed to the absolute minimum of processing possible for a front- side back-to- 



Third International Symposium on Space Terahertz Technology Page 159 

back diode process. In our work, the remaining process steps were completed with the 
devices laminated to a 3 mil quartz substrate. This quartz is the same as is typically used 
for the crossed-field waveguide multiplier filter structures for which the devices were 
intended. Clearly, there is a wide lattitude in substrate material, which adds a great deal of 
flexibility to the design of submillimeter-wave components. The total number of steps is 
relatively low, improving yield. Since thinned 1 or lifted off 5 devices never need to be 
handled off of the substrate material, relatively large scale integration can potentially be 
achieved with this process. 

Initial fabrication runs have been successful. Back-to-back BNN diodes have been 
fabricated using the new technique and then measured for DC and low-frequency 
capacitance characteristics. Eight micron mesas with 1.5 and 3.75 micron wide Schottky 
metalization showed good C-V and outstanding I-V characteristics. While the C-V pulses 
were approximately two to three volts wide (full-width half-maximum), leakage currents 
were as low as 50 nA with 10 volts between pads. Mesas as small as 1 micron were 
successfully patterned. 

Device Fabrication: 

A. Semiconductor layer structure 

Details regarding appropriate BNN layer structures have been addressed before2>3 A The 
general approach for a GaAs-based BNN diode, from the top surface down, is to include: 
(1 -optional) a thin GaAs cap layer, (2) an AlGaAs layer that is sufficiently thick to preclude 
tunneling but sufficiently thin to allow a large capacitance per unit area - 15 to 20 nm of 
Alo.45GaQ.55As is typical, (3) near the AlGaAs, a highly doped region in order to ensure 
that the high capacitance mentioned above is achieved at zero voltage, (4) a moderately . 
doped GaAs dnft/varactor region in which all of the doping can be depleted with little 
parasitic conduction to the metal pads, and (5) a highly doped region that provides a low- 
resistance path between the two Schottky pads. 

The structure used for these devices is as follows: (1) a 2 nm GaAs cap, (2) a 15 nm 
AlQ.45Gao.55As barrier, (3) a 3 nm GaAs spacer followed by SxlO^/cm^ silicon planar 

doping, (4) a lxl0 17 /cm 3 by 125 nm drift region, and (5) a 5xl0 18 /cm 3 by 900 nm 
conducting base. A 600 nm undoped Alo.45Gao.55As layer for use as an etch stop layer 
was located just below the active device layers, although much thinner layers have also 
been successfully used. The layer structure is shown schematically in Figure 1. 
Capacitance-voltage measurements between large capacitance pads on another area of the 
same wafer gave the data shown in Figure 2. 

The process is shown in Figure 3. We processed GaAs wafer pieces about 1.7 cm on a 
side that were laminated onto quartz pieces 76 microns (3 mils) thick by 2.5 cm in 
diameter. The GaAs wafers were initially 510 microns (20 mils) thick. 

The Schottky contacts are defined by exposing AZ 5214 with a standard image reversal 
technique and then evaporating and lifting off Ti/Pt/Au. While we did not do a surface 
isolation, the devices should show less leakage current and particularly less parasitic 
capacitance if etching (or implanting) were done through (or into) the delta-doped layer. 
Etching by using the metal as a mask has been done successfully but not on the wafers 
processed in this work. 



Page 160 



Third International Symposium on Space Terahertz Technology 





Undoped A^ 4S Ga 0S As 
1 50 Angstroms 


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r ,«,12, 2 




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Figure 1: Epitaxially grown GaAs/AlGaAs layer 
for BNN diodes used in this study. Wafer layers were 
grown with molecular beam epitaxy. Doping is silicon. 

B. Device Processing 



Figure 2: Capacitance-voltage 
characteristics measured at 1 MHz 
using large dots. 





Saadng waftr 



(b) 





Figure 3: (1-a) The Schottky contacts are deposited, and a thin isolation through the delta-doped region is 
etched optionally, (2-a) front-side passivation is deposited, (3-b) the wafer is mounted face-down on a quartz 
wafer with wax for thinning (this wafer must be suitably thin to serve as the final substrate for the device 
and may be mounted in turn on a sturdier substrate), (4-b) conventional lapping and then selective thinning 
to an AlGaAs etch stop layer located just below the active layer, (5-c) lithography for mesa isolation is 
performed from the back of the wafer (at this point the wafer is sufficiently thin that the front-side metal can 
be viewed from the back through the substrate), and (6-c) the wafer is etched, (7-d) thin Si3N4 or Si02 is 
deposited in order to passivate the back and, if necessary, opened for bonding pads, and (8-d) the thin quartz 
wafer is diced. 



Third International Symposium on Space Terahertz Technology Page 161 



The GaAs wafers were glued face down to a 75 micron thick quartz wafer, 2.5 cm in 
diameter, with UV-curing glue. The GaAs wafers were then lapped down to a 100 micron 
thickness using 5 micron, 1 micron, and finally 0.3 micron grit. Polishing was done with 
a silica colloidal suspension and pad. Selective thinning to the etch stop layer (5, above) 
was done using a 95/5 solution of ammonia and peroxide *. The peroxide/ammonia etch 
undercut the edges of the wafer two to three hundred microns, two to three times the 
vertical etch distance. Little damage of scale larger than pinholes of a micron or two in 
diameter could be seen over most of the wafers processed. A brief dip in HC1 and water 
was used to remove the remaining oxide layer. 

It should be noted that the wafers were remarkably rugged. The single cracked wafer 
observed so far occured during the lapping operation, which in our lab is performed using 
a relatively violent vibration table. The wafer that broke lost some small portions of the 
edge, but the area on which the GaAs was epoxied proved to be extremely tough despite 
cracks through the quartz. The wafer was processed to completion despite the cracks, with 
no allowances made to ease the normal rigors of microelectronic processing (spinning, 
contact lithography, etc.). 

Mesas were defined by standard positive photolithography and dry etching. Wet etching 
was unsuccessful using either photoresist masks (excessive undercutting for the smaller, 1 
micron mesas) or nitride masks (adhesion problems). The mesas were aligned to the 
frontside metal with IR backlighting, although the semiconductor is thin enough to permit 
aligning with optical backlighting if available. 

Additional process steps have been done, including backside passivation with more ECR 
nitride and etching of contact holes with a CF4/02 plasma. Clearly, the ability to deposit 
more metal in order to further reduce loss and to make MIM capacitors with the frontside 
metal and backside nitride is very attractive. Also, we have tried thinning the glue with 
acetone prior to spinning; this results in a much thinner layer (roughly 10 microns) between 
the GaAs wafer and the quartz substrate. 




Figure 4: Backlit photograph of central portion of device. The small dark rectangle in the center is the 
approximately 4x16 micron micron mesa. The larger, dark portions with the thinner protrusions (2 
micron fingers) are the Ti/Pt/Au pads. The remaining area is quartz. 



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Third International Symposium on Space Terahertz Technology 



The ECR nitride process is then repeated on the back, with windows opened with CF4/02 
RIE to the frontside metal for contacting. Other MMIC-style processing, including plating, 
MIM capacitor formation, etc. could be performed at this point. 

Figure 4 shows a photograph of a completed device. 

Electrical Measurements 

Devices were checked for DC and capacitance characteristics after processing. Figure 5 
shows the 1 MHz capacitance of a 3.75 x 8 micron device. While the peak capacitance is 
lower than expected from the capacitance data shown in Figure 2, it is thought that the 
difference is probably due to the fact that the processed wafer piece is from the edge of the 
MBE wafer. Figure 6 shows the DC current leakage before backside passivation. The 
measured current is many orders of magnitude lower than would be observed with 
comparable Schottky diodes, leading us to believe that the BNN is potentially superior as a 
multiplied power source. 



60 t 

^ 50 

^ 40 
u 

I 30 
o 
§. 20 

CO 

° 10 






-I H- 1 1 



-3 -2 



-10 12 3 4 5 

Voltage (V) 



Figure 5: Measured C- V characteristics of a device with back-to-back 3.75 x 8 micron BNN diode. 



IF 



























































.- 






ID. OO 
/dlv 













O 






























/ 






















/ 












































SO. OO 



















-io. OO 



2. QOD/dlv 



< V> 



IO. OO 



Figure 6: Measured DC leakage current for back-to-back 3.75 x 8 micron diode. Traces for positive and 
negative voltages were taken by sweeping away from zero volts. Higher leakage currents can be measured 
instantaneously if the voltage is not slowly swept; similar behavior commonly observed in MESFETs 
leads us to believe that the method used above is the most relevant. 



Third International Symposium on Space Terahertz Technology Page 163 

Conclusions 

We have demonstrated a new process for back-to-back diode fabrication using a BNN 
structure. The process should make integration of submillimeter-wave diodes much easier 
while simultaneously reducing RF losses through the elimination of all non-essential high- 
dielectric semiconductor. The BNN diodes constructed with this process show a strikingly 
low leakage current. 

Acknowledgements 

The research described in this paper was performed at the Center for Space 
Microelectronics Technology, Jet Propulsion Laboratory, California Institute of 
Technology and was sponsored by the National Aeronautics and Space Administration, 
Office of Aeronautics, Space, and Technology, and by the Army Research Office. 

References 

1 W.L. Bishop, E.R. Meiburg, R.J. Mattauch, and T.W. Crowe, "A Micron Thickness 
Planar Schottky Diode Chip for Terahertz Applications with Theoretical Minimum 
Capacitance," 1990 IEEE MTT-S International Microwave Symposium Digest, p. 1305. 

2 U. Lieneweg and U. Maserjian, "Harmonic generation in the near-millimeter-wave 
range by thin-MOS structures," presented at Sixth International Conference on Infrared and 
Millimeter Waves, Miami, FL, 1981. 

3 U. Lieneweg, T. Tomunen, M. Frerking, and J. Maserjian, "Design of Planar varactor 
Frequency Multiplier Devices with Blocking Barrier," submitted to IEEE MTT Special 
Issue on Terahertz Technology, 1991. 

4 E. Kollberg, T. Tolmunen, M. Frerking, and J. East, "Current Saturation in 
Submillimeter Wave Varactors," submitted to thte IEEE MTT Special Issue on Terahertz 
Technology, 1991. 

5 E. Yablonovich, T. Gmitter, J.P. Harbison, and R. Bhat, "Extreme Selectivity in the 
Lift-Off of Epitaxial GaAs Films," Appl. Phys. Lett. vol. 51, p. 2222 (1987). 

6 Norland Optical Adhesive 61, Norland Products Inc., P.O. Box 145, New Brunswick, 
NJ 18902. 



Page 164 Third International Symposium on Space Terahertz Technology 



/&>&* N9 3 -27 7 40 



v •■ 



A 200 GHz TRIPLER USING SINGLE BARRIER VARACTOR 



Debabani Choudhury, Margaret A. Frerking 
and Paul D. Batelaan 

Jet Propulsion Laboratory 
California Institute of Technology 
Pasadena, California 91109, USA 



ABSTRACT 

/>' The GaAs Schottky varactor diode is the non-linear device most commonly 

used for submillimeter wave harmonic generation. Output power adequate to serve as a 
local oscillator source for SIS tunnel junctions has been demonstrated with whisker- 
contacted GaAs Schottky varactor multipliers in waveguide mounts up to about 800 GHz. 
In this paper, we present results for a tripler to 200 GHz using a new multiplier 
device, the single barrier varactor (SBV). This new varactor has a potential advantages 
such as stronger non-linearities or special symmetry, which make it attractive for 
submillimeter wave frequency multiplication. 

The performance of a tripler using a SBV over a output frequency range 
from 186 to 207 GHz has been measured in a crossed waveguide mount. The theoretical 
performance of the device has been calculated using large signal analysis. A comparison 



Third International Symposium on Space Terahertz Technology Pagel65 

of theoretical and measured results and a discussion of various losses in the mount and 
the varactor have also been presented. 



INTRODUCTION 



Heterodyne receivers are used for high spectral resolution shorter- 
millimeter and sub-millimeter wave astrophysics and earth remote sensing 
observations. Local oscillator, mixer and antenna are the critical components in a 
receiver. One approach to provide sub-millimeter power is to use the combination of a 
high-power millimeter-wave source with a harmonic multiplier for higher frequency 
generation. Frequency multipliers use a non-linear device to generate harmonics of the 
input frequency from a fundamental oscillator. Although the Manley-Rowe relations 
show that an ideal harmonic genarator with 100% efficiency is possible with a varactor, 
real multiplier circuits are limited by loss in the device and circuit and by impedance 
matching limitations at the input, output, idler and harmonic frequencies [1,2]. As the 
circuits become smaller with increase in frequency, impedances and losses become more 
difficult to control. 

To achieve the full capability of the diode, appropriate embedding 
impedances must be provided by the multiplier mount. The impedances at the input and 
output frequencies must be set to maximize coupling power into or out of the device. In 
higher order multipliers, current flow at the intermediate harmonics (i.e. the idler 
frequencies) will enhance harmonic conversion. Therefore, the diode must be terminated 
with a lossless reactance at these frequencies. The embedding impedances are provided by 
the multiplier mount. Nonlinearities symmetric about zero bias will generate only odd 



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Third International Symposium on Space Terahertz Technology 



harmonics, greatly simplifying the multiplier mount design. For instance a tripler 
mount for a symmetric device will be equivalent to a doubler mount for a device without 
symmetry. Similarly, a quintupler mount will be equivalent to a tripler mount, both 
requiring one idler. 

This paper presents the theoretical and experimental results of a 200 GHz 
tripler using a single barrier varactor (SBV) as the nonlinear device. 



MULTIPLIER DEVICE 

The single barrier varactor diode used as the multiplier device in our 
experiment, was developed at the Chalmers University of Technology [3,4]. These 
Chalmers devices were fabricated with the epitaxial GaAs/Alo.7Gao.3As/GaAs material 

grown as indicated in Fig.1. It is inherently a symmetric device. The Alo.7Gao.3As barrier 



19980 A 



{ 



4000 A 

.5330 A 
530 A . 



530 A 
5330 A 




Whisker 



t*^«K««^««& 



*mmmm%m* 



jTOTJTwr 



HMMlitMMHtUtMMiMU 



*V'V 



<i' 



AuGe/Ni/Au 



GaAs 



As 







GaAs 



,GaAs Substrate 
(n-doped) 

u_ AuGe/Ni/Au 



Fig.1 : Schematic of the Chalmers Device 



Third International Symposium on Space Terahertz Technology p a ge 167 

which blocks the current flow is in the center having a thickness of 213 A. On either side 



of the barrier, there is an undoped GaAs spacer having a thickness of 53 A. GaAs depletion 



region (n=1x10i? cm-3 ) on either side has a thickness of 5330 A. Top and bottom 

a 
contacts are formed on highly doped GaAs regions (n=3.4x10 18 cm-3 ) US j n g 1000 A 

O 

AuGe, 200 A Ni and 1600 A Au. The top contact is made with a whisker and the bottom is 
a large area ohmic contact. The Alo.7Gao.3As barrier will to a large extent prevent 

electrons from passing through the structure. Thus the conduction current through the 
device is very small. For moderate voltages, the conduction current is essentially caused 
by thermionic emission. The width of the depleted part of the moderately doped epitaxial 
layer will vary with bias voltage, thus forming a voltage dependent capacitance C(V). 
When the diode is biased in the forward direction, the depleted region will appear on one 
side of the barrier, and the depletion capacitance of the device will decrease with 
increasing voltage. Since the diode is symmetric, a reverse bias will in the same way 
cause a decrease in the capacitance value of the device. Hence, the maximum capacitance 
is obtained for zero voltage and is determined by the thickness of the Alo.7Gao.3As 

barrier. The minimum capacitance which occurs for maximum bias voltage, is 
determined by the doping concentration and the extension of the moderately doped drift 
region. For an appropriately designed device, similar capacitance swing with voltage as 
for the Schottky-varactor diode is expected [5]. 

The losses due to the series resistance may be larger in the SBV diode than 
in Schottky varactors, since the maximum current i m ax= CdV/dt will occur for V(t) = 0, 

i.e. when both dV/dt and C are maximum and n-doped drift regions on both sides of the 
barrier are undepleted and contribute to the series resistance. In addition, for small area 
device, the ohmic contacts exhibits higher resistance than Schottky contacts. However 
for a Schottky varactor tripler, the idler current at the second harmonic will degrade 



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Third International Symposium on Space Terahertz Technology 



the tripler performance, since any finite reactance termination will cause power losses 
in the series resistance. For the SBV tripler, this particular problem is virtually non- 
existent [4]. 

The Chalmers devices tested here, have a mesa height of about 2.5 microns 
and area of 5x5 micron 2 . In order to evaluate the dc characteristics, the device has been 
mounted in a coaxial mount as shown in Fig.2. The S-parameters are measured using a 
HP 851 OB Network Analyzer. The K-connector provides 50 ohms up to the whisker to 
allow accurate de-embedding of the mount [6]. The equivalent circuit of the diode 
mounted in a co-axial mount is shown in Fig.3. Chalmers device was measured to have a 
dc series resistance of 7 ohms. The measured C-V and l-V characteristics for the 5x5 
micron 2 Chalmers device are shown in Fig.4. The measured maximum capacitance is 




Fig.2 : Schematic diagram of the device mounted in a coaxial mount 



Device 



C(V) 



r s Lw K-Connector 

-'NA/* — 9 — ono 1 I 



#. ^«(V) J^Cj 



Fig. 3 : Equivalent circuit of a device in a coaxial mount 



Third International Symposium on Space Terahertz Technology 



Page 169 




•6 



-4-2024 

DC Bias Voltage (V) 




-4-2 2 4 

DC Bias Voltage (V) 



Fig.4 : Measured C-V and l-V characteristics of Chalmers device 

65.6 fF and minimum capacitance is 12.4 fF. The figure of merit of the diode, which is 
its cut-off frequency, is given by, 

f c; L_{_J ^} 

27tK s v^min ^max 

Chalmers device has a cut-off frequency of 1200 GHz. The diodes are found to be damaged 
when the dc voltage exceeded about 6 V. 



LARGE SIGNAL ANALYSIS 



The critical step in the multiplier analysis is to solve the voltage and 
current waveforms of the nonlinear device which is pumped and biased in an arbitrary 
embedding network. A common solution of this nonlinear problem is to use a type of 
harmonic balance technique. Time-domain current and voltage solution are sought which 



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Third International Symposium on Space Terahertz Technology 



satisfy the diode conditions and frequency-domain solutions are sought which satisfy the 
external circuit equations. In this work, a modified nonlinear program based on Siegel, 
Kerr and Hwang [5] has been used in order to calculate the tripling efficiency of the 
Chalmers single barrier varactor diodes. Fig. 5 shows the equivalent circuit of a 






v B ~± 



c <%>. 



9<»d>/ »d 



1 



2e(2fp) 



2>(3lp) 



Ze(n»p) 



Fig.5 : Eduivalent circuit of a multiplier 



multiplier. Harmonic triplers with 186 GHz,192 GHz, 200 GHz, 207 GHz output 
frequencies are calculated. We have optimized the impedance at the third harmonic 
frequency. The idler and the higher harmonics are set to open circuits. Impedance up to 
12th harmonic are analyzed. In the analyses, the measured C-V and l-V characteristic 
shown in Fig. 4 have been used. Since the series resistance is important in the device 
performance, the calculations are carried out for a range of resistances, 5 ohm, 10 ohm, 
15 ohm and 20 ohm. 

Fig.6 presents efficiency versus the input power for a SBV tripler to 192 
GHz taking series resistance of the device as a parameter. In order to quantify the effect 
of current flow in the device on the multiplier performance, we have calculated the 
performance for a device with 5 ohm series resistance with no current. The theoretical 
efficiency is found to degrade from 45% to 20% when the measured current is included. 
Higher device series resistance degrades the tripler performance significantly as the 



Third International Symposium on Space Terahertz Technology 



Page 171 



so T 

3" 45 .. 



192 GHz Output 




R»=5 Ohma, msaa CV only 



Ra=5 Ohms, meat IV.CV 

... , R»=10 Ohma,maaa IV.CV 

Ha=15 Ohma, mmat IV.CV 

7»— -- — — "*— " R»=20 Ohma, meaa IV.CV 

1 1 1 



10 20 30 40 50 60 

Input Power at Diode (mW) 



70 



80 



Fig.6 : Calculated tripling efficiency for Chalmers device 

series resistance of 20 ohm results in about a factor of four worst performance than a 
series resistance of 5 ohm. 

200 GHz WAVEGUIDE MOUNT 

To achieve optimum performance of the device, it must be provided with 

the appropriate circuit embedding impedances. The impedances at the input and output 
frequencies must be set to maximize coupling power into or out of the device. In higher 
order multipliers, current flow at the intermediate harmonics i.e. the idler frequencies 
will enhance the harmonic conversion. Therefore, the diode must be terminated with a 
lossless reactance at these frequencies. 

An output of the large signal analysis, used to optimize the device, is the 



C-3 



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Third International Symposium on Space Terahertz Technology 



required embedding impedance. In fig. 7, the real part of the optimum impedance is 



120,. 
100 

_ 80 . 



sa 



60 . 

40 ■ 

20 . 





64 GHZ 



90 mW hput powar 




10 20 30 

XI Raal Unpadanca (Ohm*) 



"20 Ohms 
■ IS Ohma 
-10 Ohma 
-4 Ohms 



5 - 



SO 

45 ■ ■ 

40 

IS 

30 

2S 

20 

15 

10 

5 





192 GHz 



50 tnW Input 




10 



20 



30 



X3 Real Impadanca (Ohma) 



Fig. 7 : Input and output circuit optimum embedding impedances calculated 
by large signal analysis 



shown on the horizontal axis and the imaginary part on the vertical axis, parameterized 
by the input power for both the input frequency (Ri t Xi) and the output frequency (R3, 

X3). Impedances are plotted for different series resistances of the device. Input power 

increases from mW to 50 mW. At low input power the real part is same as the device 
series resistance. The input imaginary impedance is the impedance corresponding to the 
maximum capacitance at input frequency. As the input power increases, the device 
capacitance decreases increasing the impedance. The real impedances needed are in the 
range from 7-30 Ohms. The imaginary impedances range from 40-100 Ohms for the 
input circuit and from 15-42 Ohms for the output circuit respectively. 

The embedding impedances are provided to the single barrier varactor 
(SBV) device by a crossed waveguide mount. In addition the mount distributes the power. 
A schematic drawing of the crossed waveguide mount is shown in Fig.8. The single 
barrier varactor device is mounted spanning the output waveguide. The output waveguide 
is actually oriented perpendicular to the plane of the paper. Power at the input frequency 
travels down the input waveguide. A low pass filter consisting of Au metallization on the 



Third International Symposium on Space Terahertz Technology 



Page 173 



quartz substrate couples the input power from the waveguide to the whisker contacted 




Fig.8 : Schematic diagram of the device in the mount 

SBV device located at the output waveguide. An E-plane tuner and a backshort at the input 
waveguide provide adjustments to optimize the embedding impedance at the input 
frequency. The output waveguide is cutoff at the input frequency preventing propagation 
down it, thereby confining the input power to the vicinity of the SBV device. The tripled 
power is coupled out the output waveguide. The embedding impedance at the output 
frequency is adjusted by varying the whisker length and by a movable backshort. The low 
pass filter prevents the output frequency from traveling to the input waveguide. A 
scanning electron micrograph of the device in the mount is shown in Fig. 9. 



EXPERIMENTAL RESULTS 



The set-up for 200 GHz tripler measurements is shown schematically in 
Fig. 10. A 60-70 GHz klystron is used as the pump source. The input power is monitored 



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Third International Symposium on Space Terahertz Technology 




Fig. 9 : Scanning Electron Micrograph of the device in the mount 




Out Fin 



Power 
Meter 



In Fin 



"^ 




Waveguide 
Switch 



Power 
Meter 



mm-Wave source 

Klystron 

60 - 70 GHz 

100 mW 



Fig. 10 : Test setup for the 200 GHz tripler measurement 



by a Anritsu power meter, calibrated to give the power at the input flange. The reflected 
power is measured using a directional coupler coupled to a second power meter. The third 
harmonic output power is measured by a third powermeter. We determine the loss at the 
third harmonic in the waveguide from the output flange to the powermeter by a 
substitution technique. The observed loss in the WR4 output waveguide is 0.032 dB/A., 



Third International Symposium on Space Terahertz Technology 



Page 175 



consistent with the resistive losses corresponding to the metal conductivity of 2x1 7 
mho/m. The flange-to-flange efficiency is defined as the ratio of the power at output 
flange to the power available at the input flange. Using various whisker lengths, the 
efficiency and output power were measured between 1 86-207 GHz. Measurements were 
taken with three different whisker lengths, 6 mil, 8.4 mil and 11 mil. It was seen that, 
the Chalmers 25 micron 2 device, contacted with a 8.4 mils long whisker gives best 
tripler performance. The measured efficiency versus input power for 186 GHz, 192 
GHz, 196.5 GHz and 201 GHz output frequencies are shown in Fig.11 for a 8.4 mil long 



8.4 mil whisker 




10 20 30 

Input Power (mW) 



186 GHz 

-■ 192 GHz 

- — 196.5 GHz 
-•—201 GHz 



Fig. 11 : Measured efficiency versus input power plot for the tripler 



whisker contact. The best performance has been achieved at 192 GHz, giving an 
efficiency of more than 2% at 40 mW input power. This is similar to the results 
demonstrated by Rydberg et al. using the same device [3]. 

To compare the experimental results to the performance predicted by 
large signal analysis, the loss in the multiplier mount is assesed. Loss arises from 
several mechanisms. At the input frequency, the finite conductivity of the waveguide and 



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Third International Symposium on Space Terahertz Technology 



dielectric loss in the filter are very small. The primary loss mechanism is due to the 
impedance mismatch. By measuring reflected power at the input, less than 0.2 dB loss 
due to mismatch was observed, over the frequency range tested. 

At the output frequency, the impact of finite conductivity is higher. In 
addition, losses due to imperfections in the backshort are critical. Other mechanisms 
include the impedance mismatch and higher harmonic generation. 

Some of these loss mechanisms at the output frequency have been modeled 
using Hewlett Packard's Microwave Design System (MDS) package. Fig. 12 shows the 



no loss 
'fl sigma=2e7 

Bl! 




50 100 

Backshort Position (mils) 



150 



Fig. 12 : Calculated transmission from the diode to the output flange of 
the tripler mount 



calculated transmission from diode in the tripler mount to the output flange of the mount 
as a function of backshort position. When there is no loss, the transmission is 100% at 
resonant positions of backshort. If a finite conductivity of 2x1 7 mho/m, the measured 
conductivity of the WR4 output waveguide is included, the peak transmission reduces to 
about 90%. If in addition, the backshort has a 10% loss, the transmission reduces to 



Third International Symposium on Space Terahertz Technology 



Page 177 



about 60%. For a backshort loss of 25%, the transmission goes down to 40%. The losses 
reduce the height of the peaks and the sharpness of the resonances out. In addition, the 
antiresonant backshort positions donot give zero percent transmission. 

The measured results are plotted in Fig. 13, which shows the relative 





Loss in Output Waveguide 


1 t- 




Intensity 

o o o o o 




A A 


£ 0.4. 
1 0-3. 
"3 0.2 . 
" 0.1- 




A/\s 


■ 




. 





50 100 150 




Backshort Position (mils) 



Fig. 13 : Plot of measured output power as a fumction of output backshort position 



output power as a function of output backshort position. This looks qualitatively similar 
to the theoretical results. The measured first peak match the theoretical results with 
10% backshort loss, while the valley and second peak are closer to the 25% backshort 
loss. 

Based on these observations, the multiplier mount loss budget is 
presented in Table-I. In the output circuit, the loss due to the finite conductivity is 
estimated as 1 dB. Loss due to backshort is 3 dB. Impedance mismatch loss and loss due to 
higher harmonics are not known. Therefore, the loss in the output circuit is estimated to 
be more than 4 dB. Input circuit loss is estimated to be less than 0.2 dB. Using these loss 
values, 0.2 dB at input and 4 dB at output, the measured flange-to-flange efficiency is 



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Third International Symposium on Space Terahertz Technology 



Table 





Loss 


Output Circuit 




Finite Conductivity 


1 dB 


Backshort Lose 


3dB 


Impedance mismatch 


? 


Hlaher harmonics 


? 


Total 


> 4dB 


Input Circuit 




Impedance Mismatch 


0.2 dB 


(Reflected power) 





corrected to determine the efficiency at diode, which is plotted in Fig. 14. Superimposed 
of diode efficiencies are the theoretical efficiency calculated from the measured CV and IV 
curves for series resistances 10 ohm, 15 ohm and 20 ohm. At low input power the 
measured efficiency follows the theoretical efficiency for a series resistance of about 12 
ohm. The measured dc series resistance is 7 ohm. At 192 GHz, the series resistance is 



Rs=10 Ohms 




. .»— --— — * •• 

Rs=15 Ohms 



Rs=20 Oh ma 



Meas ® 192 GHz 



10 20 30 40 50 

Input Power at Diode (mW) 



60 



Fig. 14 : Tripling efficiency at diode versus input power plot 



Third International Symposium on Space Terahertz Technology Page 179 



expected to be somewhat higher due to skin effect. At higher power the efficiency falls 
off. This fall off corresponds to the power at which the device starts drawing significant 
current. This suggests that the impact of the current flow in the device on the multiplier 
performance is not well understood. 



DISCUSSION AMP. CONCLUSION 

The single barrier varactor diode has been shown to be able to provide 

more than 5% efficiency as a 200 GHz tripler. About 2% flange-to-flange tripling 
efficiency is obtained using the crossed-waveguide tripler mount for symmetric devices. 
The multiplier mount has a 0.2 dB input circuit loss due to the impedance mismatch, 
introduced by the reflected power. A total loss of more than 4 dB is estimated at the 
output circuit. This includes the 1 dB loss due to finite conductivity of the waveguide and 
3 dB loss due to the backshort. 

Development of devices with lower leakage current will significantly 
improve the tripler performance. Results can be further improved by reducing the 
output circuit 4 dB loss by improving the imperfect backshort. 



ACKNOWLEDGEMENTS 

The research described in this paper was performed by the Center for 
Space Microelectronics, Jet Propulsion Laboratory, California Institute of Technology 
and was sponsored by the National Aeronautics and Space Administration, Office of 



Page 180 Third International Symposium on Space Terahertz Technology 

Aeronautics, Space and Technology. Authors would like to thank Prof. E.Kollberg for 
kindly supplying the SBV devices and Mark Natzic for whisker contacting the devices into 
the multiplier mount. Authors also would wish to thank T.J.Tolmunen for various useful 
discussions and Hans Grdnqvist for giving some valuable information about the device. 



REFERENCES 

[1] T.J.Tolmunen and M.A.Frerking.'Theoretical Efficiency of Multiplier Devices', 

Second International Symposium on Space Terahertz Technology, 197- 211, 
(1991). 

[2] P.W.Penfield and R.P.Rafuse.'Varactor Applications', The MIT Press, Cambridge, 
MA, (1962). 

[3] A.Rydberg, H.Gronqvist and E.Kollberg, 'Millimeter- and submillimeter-wave 
Multipliers using Quantum Barrier varactor (QBV) Diodes', IEEE Electron 
Device Letters, 373-375, vol.11, No. 9, September, (1990). 

[4] E.Kollberg and A.Rydberg.'Quantum-Barrier-varactor Diodes for High- 
Efficiency Millimeter-wave Multipliers', Electronics Letters, 1696-1698, 
Vol.25, No.25, (1989). 

[5] O. Boric, T.J.Tolmunen, E.K.Kollberg and M.A.Frerking.'The Anamolous 
Capacitance of Quantum Well Double-Barrier Diodes', submitted to International 
Journal of Infrared and Millimeter Waves (1992). 

[6] M.Frerking and J.East.'Novel Varactors', submitted to the Proceeding of the IEEE 

(1991). 
[7] P.H.Siegel, A.R.Kerr and W.Hwang, 'Topics in the Optimization of Millimeter- 
wave Mixers', NASA Tech. Paper 2287, (1984). 



Third International Symposium on Space Terahertz Technology Page 181 

A Submillimeter Tripler Using a Quasi-Waveguide Structure 

Neal R. Erickson and German Cortes— Medellin -</^=^3 R 

Five College Radio Astronomy Observatory 

Department of Physics and Astronomy /fro •^'Ky' 



University of Massachusetts V Q O ^ & %t t** 

Amherst. MA 01003 ** ^ *> • 7 4 I 



Abstract 

A new type of frequency multiplier structure is being developed which is suitable for 
application at frequencies above 1 THz. This structure preserves some of the properties of 
waveguide for mode control, yet is not truly single mode. The device resembles a sectoral 
horn, with a varactor diode mounted near the throat. Input and output coupling are 
through the same aperture, requiring a quasi-optical diplexer. Initial tests are directed at 
building a tripler at 500 GHz, for comparison with waveguide structures. The diplexer is a 
blazed diffraction grating with appropriate focusing optics. Model studies show that the 
impedance match to a varactor should be good, and initial tests of the beam patterns of the 
prototype indicate that optical coupling efficiency should be very high. The structure also 
has the potential for use as a fundamental mixer, or as a third harmonic mixer. 

Introduction 

As the operating frequencies of receiver systems shift toward ever higher frequencies, 
it has become apparent that there is no clear upper limit to the application of single mode 
waveguide [1,2]. However, the machining and assembly problems increase rapidly above 
~300 GHz, and it seems clear that by some frequency of ~lTHz, the cost of waveguide 
components will be too high for use in most systems. This limitation is purely a practical 
one; the loss of waveguide does not seem to be a serious limit since submillimeter parts use 
very short waveguide runs. However, the loss is high enough to be of some concern. Cube 
corner mixers have been used in the higher frequency range, but suffer from a low beam 
efficiency and a rather high embedding impedance. This paper suggests a new type of 
mounting structure to replace waveguide at these frequencies, which combines some of the 
advantages of both waveguide and quasi-optical structures. 



Page 182 Third International Symposium on Space Terahertz Technology 

An ideal mounting structure for a >lTHz multiplier should have the following 
characteristics: 

1. Ease of fabrication, both in the machining and the assembly with the diode. 

2. Good mode control at all frequencies involved. 

3. Resistive losses should be low. 

4. Impedance level should match varactors, which tend to have a low real part and 
require series inductance for matching. 

5. The structure should be readily suited for whisker contacted diodes. Shape or size of 
the chip should not be critical, and whiskers should not be too short. 

6. If the device is optically coupled, the ports should be linearly polarized. 

7. Beam width should be reasonably narrow to ease the design of coupling optics. 

The structure being studied appears capable of satisfying all of these points, although 
tests are still in progress. The relative ease of fabrication of the 500 GHz prototype 
indicates that a scaled device at well over 1 THz should be practical. Additionally the 
needed coupling optics have been developed, which appear capable of separating the input 
and output beams with low loss as well as coupling them to a source and load. 

Quasi— waveguide Mount 

The structure as shown in Fig. 1, is essentially a pair of parallel metal plates 
separated by less than A/2 at the highest frequency of interest, with two intersecting 
sidewalls to guide the beam. The sidewalls intersect at 90° in the prototype, although other 
angles may be equally good or better. The varactor is mounted between the top and bottom 
plates, spaced from the vertex by less then 0.71A at the highest frequency. With these 
constraints, the structure can couple only to a mode with the electric field uniform and 
perpendicular to the plates, and the mode pattern in the H plane has single maximum. To 
produce a convenient output beamwidth, the top and bottom plates are made nonparallel so 
that the separation becomes a few wavelengths at the opening. The best situation for ease of 
design would be to make the planes parallel at the diode, and then change the angle to begin 
the flare, as in a typical waveguide— horn interface. However, this makes the structure 
impractical to build at >lTHz, so the approach taken was to maintain a continuous flare, 
which is slow enough to not greatly perturb the embedding conditions at the diode. If this 
flare becomes too fast, the embedding impedances may change and the evanescent higher 
modes at the diode are not sufficiently cut off before the separation becomes great enough for 



Third International Symposium on Space Terahertz Technology 



Page 183 



them to propagate. The relatively large spacing of the plates (compared to reduced height 
waveguide) makes mounting a diode and whisker easier, and reduces the losses. The wide 
side wall spacing minimizes their contribution to the loss except near the vertex. 

A model study was done to determine the embedding impedances in this structure. A 
coaxial probe was introduced up to the effective terminal where the diode would be 
contacted, and a "contact whisker" of various lengths and shapes extended to the opposite 
wall. Various vertex distances within the constraint of the maximum spacing were also 
tested. The general result is that any antenna within this structure acts as an inefficient 
radiator so that it maintains the character of a lossy transmission line, with the loss 
increasing with frequency. Thus it shows resonant behavior, which is pronounced when the 
plate spacing is as wide as used here. The diode terminal impedance circles the Smith chart, 
at a high value of p, with approximately periodic behavior dependent on the physical length 
of the whisker, rather than the plate spacing. The match initially appears quite poor at all 
but the highest frequencies, but it is found that through the choice of whisker length and 
shape, it is possible to achieve a considerable range of impedances at three harmonically 
related frequencies. By adding a short circuited transmission line in series with the contact 
whisker (with two adjustable parameters, length and impedance), it is possible to produce an 
even wider range of values. The net result is that a reasonable match may be made to the 
impedance of a varactor, over a bandwidth of 5%, at all three frequencies required for a 
tripler, including a low resistance at the idler. The best configuration occurs with the 



WALL 



COAXIAL TUNER 




COAXIAL 
DC BIAS PORT 

EXPANDED 
SIDE VIEW 
(E PLANE) 



DIODE 



.7X 
MAX 




DIODE AND 
ANTENNA 



DIRECTION 
OF BEAM 



TOP VIEW 
(H PLANE) 



Fig. l. Top view and side view cross section of the quasi— waveguide tripler. 



Page 184 Third International Symposium on Space Terahertz Technology 

maximum plate separation and vertex distance allowable, but there is still some freedom in 
the choice of whisker parameters (length, diameter and shape). No other flare angles were 
tested, so this remains as another possible adjustment. While the bandwidth of this 
particular choice of geometry is limited, it seems good enough to evaluate the potential of 
the device. 

Based on this design, a prototype tripler has been fabricated for an output frequency 
of 500 GHz. Initial tests seemed best at a frequency where comparison with the results for 
waveguide mounted devices is possible, while this frequency is high enough to permit a 
realistic assessment of the fabrication difficulties. The structure is built as a split block, 
with the bottom plane and side walls in one part, and a flat plate forming the other half. 
The diode in this device is biased through a coaxial filter designed to present a short circuit 
at all three frequencies involved. As in typical mixers and multipliers, the diode chip is 
mounted on the end of the coaxial filter, forming much of the final section. The diode 
chosen is U.Va. type 2T2 with Cj(0) = 6fF, R s = 12ft, and V b = 11V. The flare angle of 
the plates is 9° , over a total length of 2.3 cm, so that the opening aperture is 3.6 mm. The 
whisker is mounted on the end of a short circuited coaxial section providing the needed 
reactances at the input and output. 

While the bottom half of the mount was made by electroforming over the corner of a 
cube, it could also easily be machined, except for the vertex itself, where a small radius has 
little effect. Probably the most difficult machining area is in the coaxial bias filter, which 
would be impractical at substantially higher frequencies. An alternative is to use a very thin 
capacitor for rf bypass between the diode and the bottom plane; this would also require a 
thin diode. A bias wire can then connect to a feedthrough at the vertex. 

Coupling Optics and Diplexer 

A quasi— optical device is only of value if it can be coupled efficiently with optics. In 
this case the input power is likely to be derived from a waveguide mounted varactor 
multiplier with a feed horn on the output, which may be approximated by a Gaussian beam 
waist. The output load will almost certainly be an antenna coupled mixer but the details of 
the pattern of such a mixer are presently unknown. We can only assume that a Gaussian 
beam is suitable. 



Third International Symposium on Space Terahertz Technology 



Page 185 



Beam patterns for this device are expected to be those of a uniformly illuminated 
aperture in the E plane and one sector of that due to a square array of four antennas in the 
H plane. The E plane pattern has a moderate sidelobe level reducing its coupling efficiency 
to about 85% to a Gaussian mode, while the H plane pattern is well tapered with no 
sidelobes, and thus couples with high efficiency. The beam is unusual in that the phase 
centers for the two planes are far apart. The H plane originates essentially at the vertex, 
while the E plane center is at the physical aperture. Thus the optics must be very 
astigmatic. In addition, the beams at the input and output are very different in size, 
particularly in the E plane. 






L 
01 
3 
O 

a. 



<u 




■'A 






O 

a. 



ex. 









rrrr 


1" 










// 












-a 


















-4 


- 




// 


\ 








- 


-6 






i 
1/ 


\ 

\ 
\ 








- 


-B 


















• 




11 
1 


\ 
\ 








• 


la 


■ 




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li 


\ 
\ 








• 


12 


. 




1 
1 


\ 
\ 








• 


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"* 




It 
II 


\ 

V 








— 


It 
18 


• 


\ 


'/ 


\ 

\ 
i 
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- 


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llll 




llllllllllllllllllllMIU4lllll.il 


I1U. 


uu 


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Azimuth CDegreesJ 



Fig. 2. Cuts through the principle planes of the beams in the E plane 
(azimuth) and the H plane (elevation). Solid line is at 164 GHz, dashed 
line is at 492 GHz. Beams are for the tripler without additional optics. 



Page 186 



Third International Symposium on Space Terahertz Technology 



Tests have been performed using the varactor diode as a video detector at both 
frequencies, with sources at 164 GHz using a doubled Gunn oscillator, and at 492 GHz using 
the same oscillator multiplied by six. E and H plane cuts are shown for the prototype in 
Fig. 2, for both frequencies of interest. Full contour maps show no additional features out of 
the principle planes. These patterns confirm the theoretical predictions, but the H plane at 
the output frequency is off center by about 9° . This is due to a small asymmetry in the 
whisker location or shape, but otherwise the beam shape is as expected. 

Frequency separation in the submillimeter may be done in several ways, but is 
particularly easy for a tripler because of the large frequency ratio. While a perforated plate 
high pass filter would work well [3], an easier device to fabricate is a diffraction grating. 
With the correct grating period, the input frequency can be below the onset of diffraction so 
that the grating behaves as a simple mirror, while the output can be scattered in a very 
different direction in the first order. A particular advantage of this mode of operation is 
that the input signal is well isolated from the output, which makes measurement of the 
output power easier since filters are not needed. The efficiency of this scattering can be 
made very high through the correct choice of reflection geometry and the blaze angle of the 
grooves. The electric field must be perpendicular to the ruling direction for high efficiency. 
A convenient configuration is with the grating tilted by 45° relative to the two beams, 
reflecting the input through 90° and the output by 45° . This requires a blaze angle to the 
grooves of 22.5° and a period of 0.86 mm. These optics are shown in Fig. 3. There is no 



» output\ 

\ BEAM N 



TRIPLER 



INPUT 
BEAM 




CYLINDRICAL 
LENS 



CYLINDRICAL 
BLAZED GRATING 



Fig. 3. Diplexer and optics to separate and focus the beams. The grating 
is cylindrical in the plane out of the figure. 



Third International Symposium on Space Terahertz Technology 



Page 187 



scattering loss at the input, while the theoretical scattering into the two possible unwanted 
orders at the output totals about 5%. Gratings may be curved in one dimension without loss 
of function, so a cylindrical grating with a radius of curvature of 5.3 cm is used to eliminate 
the the very rapid divergence in the H plane. For a highly curved surface such as this one at 
an off— axis angle, there is higher loss at the edges due to the projected tilt of the grooves 
relative to the polarization vector. 

These optics have been tested with the prototype and show essentially the intended 
function. The grating efficiency is in fact very high and a scan through two orders shows 
only 3% of the power in zeroth order relative to the desired first order. The spurious second 
order is exactly backscattered and is unmeasurable, but is predicted to be the same as the 
zeroth order. The focusing action is very good in the H plane, producing a beam at the 
input requiring only one further cylindrical focusing mirror or lens (in the other plane). At 
the output frequency the beam can be made fairly symmetric with just the one mirror. One 
additional complication is due to the off axis cylinder; the focal length of such a mirror 





8 




6 




4 


r\ 




0> 


2 


Q 


8 


w 


-2 


N 


-4 


<E 


-6 



I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I I ' I I I I I I 




I I I I I I I I I I I I I I I I I I I I I ■ ■ I ■ I ■ I ■ ■ I ' ■ ■ ' I ■ ■ ■ ■ I 



-25 -28 -15 -18 -5 



18 15 28 25 



El CDeg) 



3.8 p- 




-3.8 



-8 -6 -4-2 8 2 

El CDeg5 



Fig. 4. Beam pattern measured at the line focus of the cylindrical grating 

at (a) 164 GHz and (b) 492 GHz. Contour interval is 3 dB. The focal distance 

is different for the two frequencies. 



Page 188 Third International Symposium on Space Terahertz Technology 

depends on the incidence angle (in the plane of Fig. 3). Because the beams are broad in this 
dimension, this is quite noticeable in the beam patterns, particularly for the input since this 
beam is the widest, and the most off— axis. This effect is apparent in the contour map of the 
input beam, shown in Fig. 4a, as measured near the refocusing point about 50 cm away. The 
problem is much less apparent in the output beam, shown in Fig. 4b. The easiest cure is to 
make the grating a different shape, but a periodic ruling can only be made on a few surfaces. 
It appears that a conical grating can satisfy the requirements, if it represents only a small 
distortion of a cylinder, but this solution remains to be tested. Optics beyond this point are 
still being designed, but appear to be straightforward mirrors or lenses. 

Conclusions 

A new type of mounting structure is proposed for use in a frequency tripler, which is 
relatively easily fabricated for frequencies above 1 THz. Beam patterns are suitable for 
efficient coupling to an input source, although the optimum optical system remains to be 
designed. The present use of a blazed diffraction grating appears to be an excellent means to 
separate frequencies, with high input-output isolation. The behavior of the device seems 
less suitable for use as a doubler, although this has not been explored in detail. 

A second use might be as a mixer mount for a higher performance submillimeter 
mixer than cube corner mixers. The application would be much more narrow band, but 
offers more optimal impedances and a greatly reduced side lobe level. It might also be 
practical to use for a third harmonic mixer, extending the frequency range of mixers using 
multiplied sources. A recent analysis of a third harmonic mixer for 1 THz indicates that a 
properly designed mixer should be quite competitive [4], but this analysis has not been 
extended to the impedance environment presented by this structure. 

References 

1. B.L.A. Rydberg, B.N. Lyons, and U.S. Lidholm, "Multipliers for THz heterodyne 
systems," Proceedings of the Second Int'l. Conf. on Space THz Tech., pp. 212-217, 1991. 

2. N.R. Erickson, "Low noise 500—700 GHz receivers using single— diode harmonic mixers," 
Proceedings of the First Int'l Symposium on Space THz Tech., pp. 339-408, 1990. 

3. P.H. Siegel, R.J. Dengler, and J.C. Chen, "THz dichroic plates for use at high angles of 
incidence," IEEE Microwave and Guided Wave Lett., Vol. 1, pp. 8-9, 1991. 

4. N.R. Erickson, "Low noise submillimeter receivers using single— diode harmonic mixers," 
to be published in Proceedings of the IEEE, Nov. 1992. 



Third International Symposium on Space Terahertz Technology p a q e 189 

A 380 GHz SIS Receiver using N 9 3 ">•% 7 JZ, 

Nb/A10 x /Nb Junctions for a Radio Astronomical /&~ &\ 

Balloon-borne Experiment : PRONAOS /£>0£3® 

P. Febvre* + , P. Feautrier**, C.Robert* J.C. Pernot**, /° &' 

A. Germont*,M. Hanus**,R. Maoli*,M. Gheudin*,G. Beaudin*, P. Encrenaz** 

*Observatoire de Paris-Meudon, DEMIRM - URA 336 

5, Place Jules Janssen 92195 Meudon - France 

** Ecole Normale Superieure, Laboraxoire de Radioastronomie, 

24 rue Lhomond 75005 Paris - France 

+ Now at Jet Propulsion Laboratory 

MS. 168-314 

4800 Oak Grove Drive 

Pasadena, California 91109 USA 



ABSTRACT 

y ^ The superheterodyne detection technique used for the spectrometer instrument of the 
PRONAOS project will provide a very high spectral resolution (Av/v = 1(H). The most critical 
components are those located at the front-end of the receiver : their contribution dominates the total 
noise of the receiver. Therefore it is important to perform accurate studies for specific components, 
such as mixers and multipliers working in the submillimeter wave range. 

Difficulties in generating enough local oscillator (L.O.) power at high frequencies make SIS 
mixers very desirable for operation above 300 GHz. The low L.O. power requirements and the low 
noise temperature of these mixers are the primary reason for buiding an SIS receiver. 

This paper will report the successful fabrication of small (< 1 pm 2 ) Nb/Al-Ox/Nb junctions 
and arrays with excellent I-V characteristics and very good reliability, resulting in a low noise 
receiver performance measured in the 368/380 GHz frequency range. 



Page 190 Third International Symposium on Space Terahertz Technology 

I - INTRODUCTION 

Observations from a stratospheric balloon are unobstructed by the atmosphere which is 
opaque at submillimeter and far-infrared wavelengths from the ground. For this reason, a 
submillimeter balloon-bome observatory is being developed under the responsibility of the "Centre 
National d'Etudes Spatiales" (CNES), the French Space Agency. 

It consists of a stabilized gondola supporting a 2 meter diameter telescope, associated 
alternately with an infrared multiband spectrometer or a submillimeter heterodyne spectrometer 
(SMH). This last instrument will be used to simultaneously detect the 368 GHz O2 line and the 
380 GHz H2O line in the interstellar medium. It is scheduled to fly in fall 1994 using a 1,000,000 
m 3 balloon at an altitude of 37 km. 

Receivers using SIS tunnel junctions have shown better sensitivities than Schottky diode 
receivers operated at millimeter and submillimeter wavelengths. Theoretically, sensitivities 
approaching the quantum limit can be achieved [1]. 

Up to about 300 GHz the most sensitive receivers use waveguides and superconducting RF 
tuning circuits integrated with the SIS junctions [2,3,4,5,6,7,8]. Above this frequency, two options 
appear to be available. The first possibility is to design a waveguide mixer (with full-height or 
reduced-height waveguide) using two tuners (i.e. generally a backshort and an E-plane tuner) 
[9,10]. A DSB receiver noise temperature of 150 K at 345 GHz has been reported with this design 
[9]. Another possibility is to use a quasioptical SIS mixer, which is very promising above 500 GHz 
where very small waveguides are very difficult to machine [11,12,13,14]. This design is 
compatible with tuning elements. 

Finally we have chosen for our first experiments a waveguide design because it is better 
understood than open- structure mixers. 

II - RECEIVER DESCRIPTION 

A block diagram of our submillimeter wave heterodyne spectrometer is shown in figure 1. 
Rotation of a flat mirror set allows the calibration of the receiver by commuting the incoming beam 
from the telescope between a hot and a cold load. Due to the short wavelengths, a quasioptical free 
space propagation is adopted [15]. A Mach-Zehnder type diplexer is used for the 374 GHz local 
oscillator signal injection into the SIS mixer. The L.O. source consists of a phase-locked 93.5 GHz 
Gunn diode oscillator combined with two varactor diode doublers connected in series. The 
intermediate frequency (I.F.) is chosen at 5.85 GHz to allow the simultaneous detection of the O2 
line in the lower band at 368 GHz and the H2O line in the upper band at 380 GHz. The I.F. output 
feeds a specially designed cooled low-noise HEMT amplifier with a gain of 30 dB. A noise 
temperature of 18 K has been achieved at 5.85 GHz over a 700 MHz bandwidth at a temperature of 



Third International Symposium on Space Terahertz Technology 



Page 191 



27 K [16]. The signal is then amplified at room- temperature and coupled to the acousto optical 
spectrometer (AOS) subsystem with a resolution of 800 kHz in a 800 MHz bandwidth. 



telescope 



368 -380 GHz 



Calibration! r.f 
device 



Quasi-optical 

diplexer 
— T" 







374 GHz 

SIS Mixer 

& 

Integrated I.F. 

matching circuit 



I.F. 

— ► 



HEMT 
amplifier 



Liquid He cryostat 



5.85 
GHz 



I „i _. _ _ _. _ i. _ ii 



374 
GHz 



r 

Output 



Second 
Doubler 



^187 
■ GHz" 



First 
Doubler 



93.5 
GHz 



I 



Gunn 
oscillator 



Phase-Lock 
Loop 



Figure 1: Block diagram of the receiver front-end 
used for PRONAOS 

III - 380 GHz SUB MILLIMETER RECEIVER FRONT-END 

III-l - SIS junctions fabrication procedure 

We report here the fabrication process of Nb/Al-AlOx/Nb junctions with very sharp I-V 
curves and a gap voltage for one junction about 2.9 mV at 4.2 K. A high gap voltage is known to 
be necessary for good results at high frequencies ( above 300 GHz ). It is the reason why NbN 
junctions are promising for very high frequencies (above 500 GHz ). The smallest junction area 
achievable with our technology without deterioration of the I-V curve is 0.9 |im2. Our process has 
already been described in a previous paper [17]. Some parameters have changed since this article to 
obtain the desired junction area for the 380 GHz mixer. 



The fabrication process is described on figure 2 .The Nb/Al-AlOx/Nb trilayer is deposited 
on the whole substrate without breaking the vacuum in order to have a good barrier interface (see 
fig. 2-a). The diameter of this substrate is one inch, and the thickness is 95 ± 5 fim. It is made of 
fused quartz and is polished on one side. During the deposition the substrate is attached to a copper 
heat sink cooled by a closed water circuit at 20 °C. The vacuum is made by a cryopump with a 
background pressure typically under 5. 10"6 Pa. The Nb and Al films are sputter deposited by a DC 
magnetron at an argon pressure of 1.1 Pa. The Nb base electrode (170 nm thick) and 



Pagel92 Third International Symposium on Space Terahertz Technology 

counterelectrode (100 nm thick) are evaporated at a rate of 1.9 nm/sec. The Al film ( 10 nm ) is 
deposited with an oscillating substrate table at a rate of 0.2 nm/sec and is oxidized by introducing Ar 
+ 10% 02 into the chamber for 20 to 30 min at 60 to 1000 Pa. 

A positive photoresist is deposited and patterned to define the RF filter with an etching 
technique. Nb and Al films are etched by reactive ion etching in SF6- Nb is etched with a 10 seem 
SF6 flow at 0.7 Pa using 60 W of power. The corresponding etching rate is 200 nm/min. Al is 
etched at lower pressure and higher power with an etching rate of 10 nm/min; the ST 6 flow is 
5 seem, the pressure is 0.3 Pa and the power is 80 W. Under these conditions, the etching is 
dominated by a mechanical action rather than a chemical effect like in the plasma etching method. 
We observed that a CF4 gas does not etch Al even at low pressure. RIE with Ar has not been 
selected, because it produces too much damage on the resist (with Ar, it is only a mechanical 
etching process). 

After removing the remaining photoresist in acetone (see fig. 2-b), a new resist layer is 
deposited to define the junction area (see fig.2-c). This is the critical point of the process which 
limits the smallest area achievable by this technology. Our mask aligner uses a 400 nm UV source 
and is limited to 0.8 jxm resolution. In practice, it is impossible to define a diameter smaller than 1 
|j.m ( ie. an area smaller than 0.9 jirn^ ). This resist is used to protect the upper layer of Nb etched 
by RIE under the following conditions : 20 seem of SF^, 6 seem of O2, a pressure of 0.7 Pa and a 
power of 60 W. If the etching rate (100 nm/min) is lower than for the trilayer etching (see fig. 2-a), 
these conditions provide sloped edges which are easier to insulate without microshorts in the next 
step. The etch stops at the AI2O3/AI barrier, because the etching rate of Al is very low with SF6/O2. 
We use laser end point detection to avoid overetching ( it is necessary to have a sufficient thickness 
of resist for the SiO lift-off ). 

Once the upper Nb etched then a 300 nm layer of SiO is evaporated to insulate the junction 
perimeter (see fig.2-d). The excess SiO is removed in acetone (lift-off). Then, the junctions in 
series are connected together by a 300 nm layer of Nb sputter deposited with a rate of 1.3 nm/s 
through another resist stencil. The excess Nb is finally lifted-off in acetone. Different experimental 
investigations have been made to optimise each parameter. For example, the stresses in Nb films 
have been minimized by changing the Ar pressure during the sputtering step. The stresses are 
evaluated by optical interferometry. The Nb edge is another parameter we have studied. We 
succeeded in obtaining sloped edges with a reasonable selectivity by using a mixture of SF6 and O2 
at low pressure for the RIE. Finally, anodisation spectroscopy was an useful method to investigate 
the quality of the interfaces Nb/Al and to understand the diffusion problem of Al into Nb; such a 
diffusion process gives poor quality junctions . 



Third International Symposium on Space Terahertz Technology 



Page 193 



2- 



Then, the individual junctions ( 400 junctions per substrate of 1 inch diameter) are cut with a 
dicing saw and cooled in liquid helium at 4.2 K to test their I-V characteristics. It is possible to test 
6 junctions in one run. The junctions are connected with spring contacts on gold pads evaporated at 
the ends of the R.F. filter. With this technique, we can contact the 6 junctions very quickly without 
problem of series resistance on Nb surface. 



(a) 




V//////////////////////M/7A 



\i^^^^^^ 



(b) 



Base electrode 



V////////////7777?7t 



xwwwvwwwwwv 



(c) 



(d) 



Resist 



72. 



KWVW\V^VVVW\\S\M 



SiO 



X 






*■'*■].■! .' ' . 



m^K\^\\^\\\N\S\V^5^ 



Nb contacts 



(e) 




^^^^^^ 



Figure 2: Fabrication process of Nb/AI-AIOx/Nb junctions 

(a) Nb/AI-AIOx/Nb deposition. Definition of the base electrode by photolithography, (b) 
Etching of the trilayer. (c) Etching of the upper electrode, (d) Self-aligned deposition of a 
SiO insulating layer, (e) Nb interconnection layer. 



Figure 3 gives an example of a typical I-V curve of an array of 2 junctions in series. The 
area of each junction is 0.9 nm^, so the effective area of the array is about 0.45 jjirA 




Pagel94 Third International Symposium on Space Terahertz Technology 



Figure 3 

20 uA / div 
i 



2 mV / div 

A eff = 0-47 p.m2 

RN = 150 ft 

R 300 K = 182 fl 



III-2 - Mixer design 

a) General features 

The SIS mixer block is based on the Ellison design [9]. It includes an electroformed 
integrated dual-mode Potter hom [18] transformed by a circular to rectangular transition into a third- 
height reduced waveguide [19] to increase R.F. bandwidth and decrease the characteristic 
impedance at 150 Q . Superconducting coils (to suppress the Josephson Current), an I.F. matching 
circuit and junction DC bias are integrated in the mixer block in order to facilitate the installation of 
the SIS mixer in the laboratory cryogenerator or in the flight cryostat . This also allows better 
reproducibility of mixer performance due to the optimization of the mixer mount for the SIS 
junctions. Dimensions of the waveguide are 700 (im x 120 u.m and two contacting tuners (i.e. a 
backshort and an E-plane tuner placed at Xg/2 towards the feedhom in front of the junction) provide 
a large range of embedding impedances to the SIS junctions (see figure 4). 



b) Mixer configuration 

A low-pass microstrip filter designed on Touchtone [20] is fabricated by photolithography 
on a 0.1 mm thick fused quartz substrate; its rejection is about 20 dB at 374 GHz. The metallization 
is made of Nb like the SIS junction and this 1.8 mm long 0.2 mm wide substrate is only put down 
in the mixer block channel on a thin silicon grease film for a better thermal contact. Mechanical 
support is provided by this silicon grease film when cooled at 4 K and by the 25 \xm gold wires 
contacting the filter to ground and the LF. output. This assembly allows numerous tries of different 
junctions without breaking substrates. The LF. output gold wire is fixed with silver glue on the 
low-pass filter at one end and directly on the LF. matching circuit at the other end. This matching 
circuit formed on Duroi'd (£r=10.2) supports the junction DC bias too. This avoids the sudden 
impedance change of a SMA connector, increases the LF. bandwith and decreases the LF. losses. 
The DC bias includes two 10 kQ chip resistors (to prevent junction from being destroyed by voltage 
spikes) followed by an insulated wire soldered at X/4 of a X/2 stub (see figure 5) to provide 
approximately an open circuit at the LF. frequency of 5.85 GHz on a 700 MHz bandwidth. The 



Third International Symposium on Space Terahertz Technology Page 195 

25 u.m gold wire is the first part of the I.F. matching circuit, then a length of a micTostrip line 
provides a real impedance transformed into 50 Q by a X/4 line (figure 5). 




Figure 4 



Miniature connectors 
(for bias) 



I.F. output 




-§fj e 



Pocket for 

superconducting 

coil 



s 



figure 5 



Page 196 Third International Symposium on Space Terahertz Technology 

The 1.8 cm diameter superconducting coils have been designed to produce 310 Gauss with 
a current of 1 A. Each one is made of about 1200 turns of Niobium-Titanium superconducting wire. 
Indeed, for circular junctions of surface S, the magnetic field suppressing the Josephson current is 
given by : 

2,23 . IP' 11 



B(Gauss) = = 

d.VS (for one flux quantum ) 



with : d = 2Xl+w where: 
Xl = London penetration depth of Nb (m) 
w = width of insulator between the two superconductors (m) 

The area of the smallest junctions fabricated in the laboratory is about 1 \xm 2 . So, with a 
pessimistic value of the London penetration depth (400 A), B = 255 Gauss ; the real value should 
be lower. The coils are small, because the flight cryostat was specified for a smaller Schottky 
mixer. Moreover, some constraints about the optical axis were already fixed in the flight cryostat. 
The mixer block is a Faraday cage for these coils against electromagnetic spikes even if any external 
magnetic field can penetrate into it . 

IV - LABORATORY MEASUREMENT BENCH 

Results shown further have been obtained on a laboratory bench with a 4 K cryogenerator 
including two closed circuits of helium. The first one is a classical CTI 1020 compressor including 
two stages at 50 K and 12 K. The second one is a Joule-Thomson expansion pumping on the 12 K 
stage to reach 3 to 4 K on the "4 K" stage. Temperature can be quickly changed and stabilized by 
varying the return helium pressure of the 4 K helium circuit [21]. A teflon corrugated window is 
used on the room temperature shield for the quasioptical RF input. The heat flux entering the 
cryogenerator is then reduced with an IR filter. It's a 80 Jim thick (one wavelength at 374 GHz) 48 
mm diameter crystalline quartz plate mounted on the 50 K stage shield. Then a 0.8 mm thick 
fluorogold window, 38 mm diameter, filters the far IR 50 K blackbody radiations. The SIS mixer is 
on the 4 K stage at the focus of a cold corrugated teflon lens cooled by the same stage. 

Mechanical contacting tuners are operated by vacuum feedthroughs and are manually 
movable with micrometer drives when measuring receiver performance. Each electrical wire, I.F. 
cable or tuner drive is thermalized at 12 K and 50 K to exhaust heat flows. Some miniature 
connectors are used for the DC bias. A four points measurement of the I-V curve releases us from 
any series resistance. 

The I.F. output of the SIS mixer is connected to a semi-rigid cable followed by a coupler, an 
isolator and the HEMT amplifier. This low-noise amplifier is installed on the 12 K stage, its output 
cable is thermalized at 50 K before going out of the cryogenerator (see figure 6). The coupler is 



Third International Symposium on Space Terahertz Technology 



Page 197 



used to inject an additive noise at the I.F. mixer output to know its match relatively to 50 Q. A 
preliminary calibration without mixer allows us to calculate approximatively the mixer temperature 
Tm and its conversion losses LM- 

The socket ot the superconductive coils is installed on the 12 K stage to have a better thermal 
contact between superconducting and copper wires and to prevent a heating of the 4 K stage. The 
L.O. and signal injections are achieved by a quasi-optical diplexer. The coupling ratio for the L.O. 
is higher than 90 %. 



Corrugated n 
teflon window ^ 



For additive 
noise injection 



Towards 
room-temperature 
amplifiers 
Copper wire A 




. Towards 
\ micrometer 
drive 



380 GHz SIS mixer 
(with I.F. matching circuit) 



Quartz window: * Fluorogold window: 

thickness: one wavelength thickness: one wavelength 



Figure 6 



Page 198 Third International Symposium on Space Terahertz Technology 

V - LOCAL OSCILLATOR VARACTOR DIODE DOUBLERS 

The structure of each doubler has already been described in a previous paper [22]. The 
maximum efficiency found for the first doubler was about 18 % for an incident power of 15 mW 
with a 5P8 diode of the University of Virginia. The input frequency was 91.6 GHz and the output 
power was higher than 6 mW with a 50 mW input power. These results haven't been found again 
with the other doubler block at 93.5 GHz. They were due to a very good coupling between the 
diode and the waveguide by the whisker. More commonly, we can reach 3 to 4.5 mW with a good 
reproducibility and with an input power of 50 mW at 93.5 GHz. A typical curve of our last results 
is shown on figure 7. 















93,5-187 GHz Doubler 




Output Power 


(5P8 diode) 








(mW) 


Efficiency 


<; . 






_ Q<?n 


4,5 - 


■l*' 


Efficiency "'■--. ry^ _ 


- 8% 


4 - 


■ 1 

$ 

- * 
■ 
* 


£l * * - 


- 7% 


3,5 - 


i 

■ 




■ 6% 


3 - 


" i 

■ 

* 


PDutput Power 


- 5% 


2,5 - 


- 9 






2 - 


1 

■ 
1 

■ > 




■ 4% 


1,5 - 


u / 




- 3% 


1 - 


i ^/ 




- 2% 


0,5 • 


■ / 


Whisker length - 


- 1% 


■ J§r 


289 microns 




o ntffrT., .. • ■ 


iiit 


_ r&z. 


10 20 


30 40 50 60 70 




Input Power (mW) 



Figure 7 



Concerning the second doubler, its input power (approximatively the output power of the 
first doubler) is low and consequently its efficiency is relatively low. Indeed, we can see on 



Third International Symposium on Space Terahertz Technology 



Page 199 



figure 7 that the efficiency of the first doubler is lower than 4 % at 187 GHz for an input power of 
about 5 mW. 

Partly due to the much higher frequency, we can foresee that the second doubler will not 
produce so much power. Such a local oscillator cannot be used for a Schottky mixer. Nevertheless, 
some diodes whose the maximum efficiency is obtained for a 3-4 mW input power like bbBNN 
diodes could provide sufficient power to pump a Schottky mixer [23]. The best output power 
obtained at 374 GHz is approximatively 30 p.W with the bolometer hom put directly across from the 
second doubler horn, i.e. an efficiency lower than 1%. Two types of diodes have been tested, 2T8 
and 2T9, they come from the University of Virginia and we can see on the following figure 8 that 
the 2T8 diode provides more power than the 2T9 diode. This is partly due to its smaller capacitance 
(4 £F versus 8 fF for the 2T9 diode). 











Output Power 
(microwatts) 


187-374 GHz Doubler 




30 -i 








25 - 








20 - 








15 - 




2T8 diode / 
whisker length: / 
146 microns / 




10 - 






♦ 


5 - 






2T9 diode 

whisker length: 

150 microns 


f) t 








1 


2 3 
Input Power (mW) 


ii| i i i i 1 

4 5 



Figure 8 



Page 200 Third International Symposium on Space Terahertz Technology 

To prevent the second doubler diode from being destroyed by voltage spikes and due to the 
low input power, it has been short circuited in direct current instead of being reverse biased for an 
optimum efficiency, this diminishes the output power. Moreover, input and output backshorts of 
each doubler have been fixed or soldered which still damages performance. So the output power at 
374 GHz is about 13 (iW. Other measurements have been made with a quasi-optical bench 
composed of two corrugated lenses which is approximately the bench used for the measurements 
of the SIS mixer. The output power is then 10 jiW. This local oscilltor signal is powerful enough, 
even to pump four SIS junctions in series. 

VI - RESULTS 

Different types of junctions have been tested with 2, 3 or 4 junctions in series coming from 
the same wafer. The best results obtained with each substrate are summarized in table I. The L.O. 
frequency is 374 GHz, the I.F. center frequency is 5.85 GHz. Measurements have been made with 
a 285 MHz I.F. bandwidth filter, we used the Y-factor method with 2 loads at 77 K and 295 K. 

Q We can firstly point out the good match between calculated and measured values of the 
magnetic field suppressing the Josephson current Ij. The product B(Ij = 0) x D is reported on the 
following table II (for one flux quantum), where B(Ij = 0) is the magnetic field suppressing the 
Josephson current and D the diameter of one junction. This product should be constant for junctions 
fabricated on the same wafer according to the previous formula of III-2-b: d is a parameter 
depending only on the oxidation time of aluminium in AI2O3. We see that B(lj=0)x D is nearly 
constant to within about 10 %, this comes from the uncertainty of the junction areas. We can also 
deduce the London penetration depth of our niobium films which is about 600 A. 

□ Nevertheless, the Josephson Current is not always completely suppressed with one flux 
quantum, because the areas of the junctions in series are slighdy different. The Josephson current 
for each of the couple of junctions in series of one substrate is reported on figure 9. 

The relative difference of the magnetic field suppressing the Josephson Current of each 
junction taken individually is about 5 to 10 %, that means a relative difference of area between the 
two junctions of 10 to 20 %. Such a difference is in good agreement with the accuracy of 
photolithography to define small junction areas. For this reason, the current densities and the coRnC 
products are not exactly the same for the different junctions of the table I. This corresponds to the 
uncertainty of the value of the junction area. 

Q The measurements of the required L.O. power are deduced from a preliminary calibration 
of the L.O. output power as a function of the first doubler self-biased voltage. The required power 
depends on the square of the number of junctions in series; four junctions in series should require 



Third International Symposium on Space Terahertz Technology 



Page 201 



about four times as much power as two junctions in series. We observed a 3.7 dB difference 
between expected and measured values which corresponds mainly to the R.F. mismatch at the 
374 GHz frequency since we measured the incoming L.O. power. And we can see that the 
difference of the conversion losses for these junctions is 3 dB, this point confirms the first one. 



Junction 


E380-1-8-2 


E380- 1-6-5 


E380- 1-8-4 


E380-1-4-1 


E380-1-8-6 


Diameter (jim) 


1.1 


1.5 


1.1 


1.9 


1.1 


Number of 

junctions in 

series 


2 


3 


2 


4 


2 


Effective 
surface (|J.m 2 ) 


0.47 


0.59 


0.47 


0.71 


0.47 


Rn(Q) 


143 


137 


150 


113 


143 


coRnC 
at 374 GHz 


9.5 


11.4 


10 


11.3 


9.5 


j c (A/cm 2 ) 


4600 


3600 


4600 


4200 


4400 


L.O. power 
(HW) 


? 


? 


7 


7.5 


0.8 


Magnetic field 
applied (Gauss) 


175 


255 
(2 flux quanta) 


175 


192 
(2 flux quanta) 


185 


DSB receiver 
temperature (K) 


1200 


470 


360 


525 


310 


Mixer noise 

temperatureTM 

(K) 


? 


200 


195 


225 


155 


Conversion losses 
(dB) 


? 


11 


9,1 


11,8 


8,8 


Transmitted I.F. 
power 


between 10 
and 40 % 


= 90% 


= 90% 


= 98% 


= 97% 


Contribution of 
amplifier to noise 


>70% 


57% 


46% 


57% 


50% 


table I 


Junction 


E380-1-8-2 


E380- 1-6-5 


E380-1-8-4 


E380-1-4-1 


E380- 1-8-6 


B(Ij = 0)xD 
(Gauss x (im) 


193 


191 


193 


182 


203 



table II 



Page 202 



Third International Symposium on Space Terahertz Technology 



□ Relatively high conversion losses result in a contribution of 50 % for the HEMT 
amplifier in the receiver noise temperature. These conversion losses include intrinsic conversion 
losses increased by RF quasioptical injection, RF and I.F. mismatches, RF filter and I.F. matching 
circuit losses. Differences of receiver noise are mainly due to miscellaneous conversion losses. 
Indeed, some different effective areas of junction have been tested and the couple of tuners don't 
enable to completely tune out the junction capacitance because the coRnC product is high (>8). So 
the excess of conversion losses corresponds to a higher RF mismatch. 

Some typical curves of different measured junctions are shown on figure 10. 



dc Josephson Current 
for each junction 
(microamperes) 
15 



Junction E380 -1-8-2 

Seff = 0.47 microns A 2 

Normal resistance: 143 ohms 








Substrate with 
2 junctions in series 



10 to 20 Gauss difference 




50 



100 150 200 250 300 

Magnetic field (Gauss) 



figure 9 



Remark : Exact values of magnetic fields haven't been measured but calculated with current flowing 
through the coils. Error is around ±5 %. 



Third International Symposium on Space Terahertz Technology Page 203 

The three curves shown below are some experimental curves digitalized by our data 
acquisition system of different arrays of SIS junctions in series fabricated on the same wafer with 
nearly the same normal resistances (about 150 Q). Fot this reason, the current densities are of the 
same order of magnitude for each array. 

Static I— V curves tor 2.3 and * SIS junctions in series 




V (mV) 

figure 10 



10 12 I* 16 18 



We can see on figure 1 1 the dc characteristic of the junction E380-1-8-6 (a). Also shown on 
this figure is the I-V curve of the same junction pumped with the 374 GHz L.O.. The width of the 
photon assisted step is 2.h.VL.o7e where h is the Planck constant , e is the electron charge and vl.o 
is the frequency of the local oscillator. 



60 - 



40 



20 




V (mV) 



figure 11 



10 



Page 204 Third International Symposium on Space Terahertz Technology 

The static impedance at the bias point (approximatively 4 mVrthe middle of the first photon 
step) is about 500 Gl so the I.F. circuit has been designed to match this impedance, we assumed that 
it is close to the impedance at 5.85 GHz (Ri.r). The normal resistance is 143 Q and the range of the 
quotient Ri.f./Rn has been found to be contained between 3 and 4.5, the value of Ri.f. being 
adjusted by varying the L.O. power. We can see in table I that the I.F. match for the last four 
junctions is good which validates our assumption. 

On the contrary the high receiver noise temperature measured for the first junction was due 
to a poor I.F. match, the I.F. impedance being unknown at that time. 

Some I-V curves with and without suppression of the Josephson current are plotted on the 
following figure 12. We can observe 3 Shapiro steps due to the coupling of the L.O. power with 
the Josephson current. when it is not suppressed. The width of these steps is exactly one half of the 
quasiparticle step due to the L.O. power. These sharp steps partly explain the instabilities observed 
when the Josephson current is not. completely suppressed. 











JUNCT 


ON E380-186. T=4.38 K 


80 - 






60 - 




/ 


40 - 




2.h.i//e *s 




2.h.u/2.e 


\ Y 


20 - 






; J 


*~ V 


VITHOUT 

JOSEPHSON CURRENT 
VITH 


- 


- ~ 




. i i i 1 i i i 1 i i ..... , 



s 

V (mV) 



10 



12 



figure 12 



Some dependances of different parameters are shown on the following figures. We can see 
on figure 13 the influence of the magnetic field to the noise receiver. 

The noise temperature begins to increase for a magnetic field lower than 170 Gauss which 
corresponds to a residual Josephson Current of about 1 (iA providing an additive Josephson noise 
coming with instabilities of the I.F. output power. 



Third International Symposium on Space Terahertz Technology 



Page 205 



DSB receiver 
noise temperature 

(K) 
450 t 



400 -- 
350 -- 
300 -- 



Junction E380 -1-8-6 

Seff = 0.47 microns A 2 

Normal resistance: 143 ohms 



250 -- 



200 



Bias current: 10 microamperes 

Physical junction temperature: 4.41 K 

I.F. Bandwidth: 285 MHz 



— i 1 1 1 1 1 r- 

120 140 160 180 

Magnetic field 
(Gauss) 



200 



220 



240 



figure 13 

The noise temperature is plotted as a function of bias current on figure 14. The receiver 
noise temperature remains lower than 330 K with a relative variation of bias current of 20 % which 
is adequate for our balloon-borne experiment where there is no remote control of the DC bias; all the 
other parameters remained unchanged. 

At last, the L.O. frequency was varied from 345 to 385 GHz (see figure 15) the receiver 
noise temperature is higher at lower and higher frequencies than 374 GHz. This is partly due to the 
narrow RF bandwidth of the Potter hom. We can point out that the receiver temperature is below 
380 K in the frequency range from 355 to 385 GHz. 



Influence of temperature was only observed with the junction E380- 1-6-5. With other 
junctions, the mixer noise temperature has not decreased by cooling more the junction; this is 
certainly due to a poor thermal contact with the silicon grease film. 



Page 206 



Third International Symposium on Space Terahertz Technology 



DSB receiver 

noise temperature 

(K) 
460 -r 




Junction E380 -1-8-6 

Seff = 0.47 microns A 2 

Normal resistance: 143 ohms 



Physical junction temperature:4.32 K 

I.F. Bandwidth: 285 MHz 

Magnetic Field - 210 Gauss 



9.5 10 

Bias current 
(microamperes) 



10.5 



11 



11.5 



figure 14 



DSB receiver 

noise temperature 

(K) 

410 1.. 

3901 



370 + 

350 

330 



Junction E380 -1-8-6 

Seff = 0.47 microns A 2 

Normal resistance: 143 ohms 



Bias current = 9.1-10.6 microamperes*' 

Physical junction temperature :4.3 K 

I.F. Bandwidth: 285 MHz 



310 



— I — 
350 



— I — 
355 



-+- 



+ 



+ 



■-«a« 



345 



360 365 370 

L.O. Frequency (GHz) 



375 



380 



385 



figure 15 



Third International Symposium on Space Terahertz Technology Page 207 

V - CONCLUSION 

Some Nb/Al-Al203/Al SIS junctions with small areas and sharp I-V curves have been 
successfully fabricated, dc measured and integrated in the mixer. The smallest area achievable with 
our process is about 0.9 (im^. Arrays of two junctions with this area have been made, the effective 
area is then around 0.45 (jm^. They are very stable according to some repeated thermal cycles: more 
than 15 cycles have been completed between room temperature and 4 K temperature and no change 
has been detected. This reliability is essential for space applications. 

The 380 GHz SIS mixer was designed with an integrated I.F. matching circuit and two 
integrated superconducting coils; it has been tested over a 40 GHz L.O. bandwidth. The best 
receiver noise temperature (310 K DSB) has been measured with an array of a couple of junctions 
in series having an effective surface of 0.47 |im2 and a normal resistance of 143 Q.. The L.O. 
frequency was 374 GHz. The relatively high conversion losses (8.8 dB) reveal a R.F. mismatch. It 
could be decreased by using junctions with lower capacitances (i.e. areas) and lower normal 
resistances. Then the fabrication of SIS junctions with higher current densities is planned. The 
lowest mixer noise temperature is around 155 K and some new junctions with lower normal 
resistances should also reduce it So we are optimistic for the following. 

The 374 GHz L.O. source has been made with a fundamental InP Gunn Oscillator at 
93.5 GHz followed by two GaAs varactor doublers in series. This subsystem provides enough 
power to drive the SIS mixer even with 4 junctions in series but a more powerful first multiplier 
will be necessary to produce more power at higher frequencies (above 500 GHz) for the future. 

A 6 GHz low-noise H.E.M.T. amplifier has been specifically designed for cryogenic 
applications, it meets fully the specifications and will be used in connexion with the SIS mixer. The 
contribution of the amplifier to the system noise is about 50 % due to the high conversion losses. 
We hope that the new junctions will decrease this contribution. 

Acknowledgments : We would like to thank Gilles Ruffle for his valuable aid and support, 
Andre Deschamps for the data acquisition system and Olivier Perrin for the design of the doubler 
blocks. We are especially grateful to Serge Lebourg and Jean Morin for their help on the mechanical 
realizations for the measurement bench. We also wish to thank Veronique Serpette (Observatoire de 
Paris) for the numerous photolithographies of the I.F. matching circuits. In addition we thank Marc 
David for his assistance and support on cryogeny. Thanks also to Albert Brel, Annick Gassais and 
Francoise Gadea for their technical help. We would also like to thank Matthew Carter and Jacques 
Blondel of IRAM (Institut de Radioastronomie Millimetrique) for useful discussions. 
Finally we are greatly indebted to William R. McGrath for his careful reading and numerous 
comments on this article. 

This work is supported by the Centre National d'Etudes Spatiales (CNES) and the 
C.N.R.S. (URA 336) 



Page 208 Third International Symposium on Space Terahertz Technology 

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[2] S.K. Pan, A.R. Kerr, MJ. Feldman, A.W. Kleinsasser, J.W. Stasiak, R.L. Sandstrom and W.J. Gallagher "A 

85-116 GHz SIS receiver using inductively shunted edge junctions", IEEE Trans. MTT, Vol.37,N 3, March 1989 

[3] A.R. Kerr, S.K. Pan: "Some recent developments in the design of SIS mixers" Jnt. J. of Infrared and Millimeter 

Waves, Vol.1 1,N° 10,1990 

[4] R. Blundell, M. Carter and K.H. Gundlach:"A low-noise SIS receiver covering the frequency range 215-250 GHz, 

InU. of Infrared and Millimeter Waves, Vol 9,1^4,1988 

[5] BiM. Ellison and R.E. Miller: "A low-noise 230 GHz SIS receiver", InU. of Infrared and Millimeter Waves, 

Vol.8,pp 609-625June 1987 

[6] H.H.S. Javadi, W.R. McGrath,S.R. Cypher, B. Bumble, B.D. Hunt and H.G. Leduc:"Performance of SIS mixers 

at 205 GHz employing submicron Nb and NbN tunnel junctions",Digest of the 15th International Conference on 

Infrared and Millimeter Waves,December 1990 

[7] D. Winkler, W.G. Ugras, A.H. Worsham and D.E. Prober, N.R. Erickson and P.F. Goldsmith: "A full-band 

waveguide SIS receiver with integrated tuning", IEEE Trans, on Magnetics, ^1.27^*2, March 1991 

[8] J.W. Kooi, M. Chan, T.G. Phillips, B. Bumble and H. G. Leduc:"A low-noise 230 GHz heterodyne receiver 

employing 0.25 p.m 2 Area Nb/AlOx/Nb tunnel junctions", 2"d International Symposium on Space Terahertz 

Technology, Jet Propulsion Laboratory, California Institute of Technology, Pasadena, Feb. 1991 

[9] B.N£llison, P.L.Schaffer, W.Schaal, D.Vail, R.E.MiUer : "A 345 GHz SIS receiver for radio astronomy ", IntJ. 

of Infrared and Millimeter Waves, Vol. 10, N° 8, 1989 

[10] C.E. Honingh, M.M.T.M. Dierichs, H.H.A. Schaeffer, T.M. Klapwijk and Th. de Graauw:"A 345 GHz 

waveguide mixer with two mechanical tuners using an array of four Nb-Al-AbC^-Nb SIS junctions", 2 n d 

International Symposium on Space Terahertz Technology, Jet Propulsion Laboratory, California Institute of 

Technology, Pasadena, Feb. 1991 

[II] M. Wengler, D.P. Woody, R.E. Miller, T.G. Phillips:"A low noise receiver for millimeter and submillimeter 
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[12] T.H. Buttgenbach, R.E. Miller, M.G. Wengler, D.M. Watson, T.G. Phillips:"A broad-band low-noise SIS 

receiver for submillimiter astronomy", IEEE Trans. MTT, Vol. MTT 36, pp 1720-1726, Dec. 1985 

[13] X.Li, P.L. Richards, F.L. Lloyd, "SIS quasiparticle mixers with bow-tie antennas", Int. J. of Infrared and 

Millimeter Waves, Vol. 9, pp 101-103, 1988 

[14] J. Zmuidzinas, H.G. Leduc, "Quasi-optical slot antenna SIS mixers", to be published in IEEE Trans. MTT, 

1992 and 2 n{ * International Symposium on Space Terahertz Technology, Jet Propulsion Laboratory, California 

Institute of Technology, Pasadena, Feb. 1991 

[15] PJ.Goldsmith, " Quasi-optical techniques at millimeter and submillimeter wavelengths", Infrared and Millimeter 

Waves vol 6 :System and components, KJ.Button (editor), Academic Press, New York, p 277-343, 1982 . 



Third International Symposium on Space Terahertz Technology Page 209 

[16] CRobert, M.Gheudin: " A 6 GHz HEMT low-noise cooled amplifier for a radioastronomical submillimeter 

heterodyne receiver ", 15th International Conference on Infrared and Millimeter Waves, Conference digest, pp. 127- 

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[17] PJeautrier, J.Blondel, MHanus, J.Y.Chenu, P.Encrenaz, M.Carter :"Low noise 80-1 15 GHz quasiparticle mixer 

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[19] Mixer block constructed by Radiometer Physics, Meckenheim, Germany 

[20] Touchtone CAD Software, Eesof 

[21] J.CMarechal, J.CPernot, PJ-Encrenaz: "A 2K closed cycle cryogenerator", Conf. URSI, Granada, Sept. 1984. 

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p 210 Third ^ternational Symposium on Space Terahertz Technology 

N93-2? 743 

//~3-5 A low noise 410-495 heterodyne two tuner mixer, 

/ GpOo^l using submicron Nb/Al 2 3 /Nb Tunneljunctions 

y G. DE LANGE*, C.E. HONINGH', M.M.T.M. DIERICHS*, R.A. 

PANHUYZEN*, H.H.A. SCHAEFFER", T.M. KLAPWUK # , H. VAN DE 
STADT*. M.W.M DE GRAAUW'. # Universitv of Groningen. Ntjenborgh 4. 9747 
AG Groningen; "Space Research Organisation of the Netherlands. Landleven 12. 
9747 AD Groning en. 



A 410-495 GHz Heterodyne receiver, with an array of two Nb/Al 2 0-/Nb 
tunneljunctions as mixing element is described. The noise temperature 
of this receiver is below 230 K (DSB) over the whole frequency range, 
and has lowest values of 160 K in the 435-460 GHz range.The 
I calculated DSB mixergain over the whole frequency range varies from 
>^-11.9 ± 0.6 dB to -12.6 ± 0.6 dB and the mixer noise is 90 ± 30 K. 

Introduction 

SIS-mixers are currently being used as heterodyne receivers up to submillimeter 
wavelengths [1][2]. Two different types of mixers are used to achieve low noise 
receivers: quasi-optical mixers, with a fixed tuned broadband planar antenna, and 
waveguide mixers, with one or two tuning elements. 

In this paper a two tuner waveguide mixer designed for the 400-500 GHz range is 
described. The used mixing elements are two different arrays of two Nb/Al 2 3 /Nb 
tunneljunctions. This mixer is the second frequency step towards our goal to make 
a THz receiver (350-490-750-1000 GHz). At these high frequencies the dimensions 
of a waveguide structure become very small (100-400 jim) and it is unclear how far 
losses due to surface irregularities deteriorate the mixer performance. 
The laboratory tests of the mixer show that in the 400 to 500 GHz frequency range, 
waveguide mixers can still achieve low noise temperatures (as low as 160 K DSB), 
and the flat response over the band indicates that the current design can be scaled 
up to higher frequencies. 



Third International Symposium on Space Terahertz Technology Page 211 



The behaviour of SIS-mixers is well described by theory [3][4]. An extensive 
comparison between theory and measurements has been performed at 345 GHz [5] 
and this comparison is started in the 410-490 GHz range. First results show a good 
agreement between calculated and measured IV-curves, from which the 
electromagnetic environment of the junction is deduced. 

This paper describes the design of the mixer, the results from the two different 
arrays of junctions and some preliminary analysis of the mixer performance. 

Receiver design 

The mixer block design for 400-500 GHz is a scaled version of the 345 GHz mixer 
described by Honingh et al.[6]. 

The mixer system is placed inside an Infrared Laboratories HD 3 cryostat. The 
signal and LO-power enter the cryostat via a 1 mm thick HDP window of 3 cm 
diameter. On the 77 K radiation shield a 200 nm thick quartz plate serves as heat 
filter. 

A diagonal horn with an aperture size of 4;5 mm, a length of 12 mm and a flare 
angle of 11 degrees is used. Laboratory tests of this horn showed a good gaussian 
beam-coupling (side lobes <-15db), equal beamwidth in E-,D- and H- planes and 
a low cross-polarisation (<-15 dB). In front of the horn a F=31.4 mm HDP lens is 
used. The lens is mounted in a holder which can be directly mounted on the 
mixerblock. The mixerblock is cut in OFHC. The full height waveguide with 
dimensions 0.44*0.22 mm, has a cut off frequency of 340 GHz. The waveguide 
system has two moving shorts as tuning elements, each with a quarter wave choke 
section to improve the quality of the short. In order to suppress the Josephson 
currents in the junction, a coil with 10.000 turns of 0.1 mm Nb wire is placed 
around the horn, in front of the mixerblock. The IF-chain consists of a Radiall 
R443533 T-bias, a Pamtech LTE 1290 isolator and a Berkshire Technologies L-1.4- 
30H IF-amplifier. 



Thomson care l natron 
410-495 GHZ 



Radloneter Physics 
76-82 (jhz fiunn-t 
douto I er * 
tr I p I er 



]:-CD"l 




® Mixer 

© Pan tech I so I a tor 

® Berkshire technologies 

Lll 1.4-30H,4 0dB Amp I I f I r . ( 1 -2 GHZ) 
® Radial I R443533 T-blas 

© I'-'Ti .4 HDP lens. 

© c'OOvhi Ouartz Window. 77K. 

© Inn H.n.P. window. 30 OK 

© Mlteq AM 3A 3 0db anpllfir CI -2 GHZ) 

© Ml teq AM 2A 2 0db anp I I fir < 1 -2 GHZ) 

O Tuner dr I ves 

O Narda Directional coupler. 

QRadlall Directional coupler 

O HP 43ba Power neter 



HP Vectra 




Stanford lock- in 



o 



K&.L F I Iter Q 
1 .4 GHZ, 100MHZ 



© 



HP . Powersensor 

8481a 

8484a 



Third International Symposium on Space Terahertz Technology 



Page 213 



SIS-junctions 

The fabrication of the used junction arrays, is described elsewhere [7]. Up to now 
two different types of arrays have been used. The arrays differ in junction area, 
current density, RF-filterstructure and gapvoltage. An overview is given in table 1. 



Junction array 


Q34 


Q35 


A (1 junction) (/xm 2 ) 


0.8 


2 


I c (A/cm 2 ) 


7000 


12.000 


R n (n) 


100 


22 


RF-filter 


Chebychev 


YaX 


Gap-voltage (mV) 


2.7 


2.4 



Table 1. Overview of the 2 different junction arrays. 

The lower gapvoltage of junctions Q35 is caused by a higher oxygen background 
pressure during sputter deposition of the trilayer. 

Measurement set-up 

A schematic diagram of the measurement set-up is given in Fig. 1. The use of two 
coherent sources, in combination with a spectrum analyser allow to analyse the 
mixer performance at the LO and the upper and lower sideband frequencies. In the 
measurements the carcinotron acted as LO-source. 

The noise temperature of the mixer was measured by using the well known "Y- 
factor" method. The measurements were corrected for the loss of the beamsplitter 
(15 Mm mylar, 95% transmission). 



Results 

The DSB noise temperature of the receiver is shown in Fig. 2 for two different 
junction arrays. Array Q34 (Rn= 100 n, A =0.8 /im 2 , Chebychev RF-filter) has noise 
temperatures below 230 K over the whole 410-495 GHz range. The lowest value 
measured was 160 K at 445 GHz. Array Q35 (Rn=22 n, 2 Mm 2 ,1/4 k RF-filter) has 
noise temperatures between 260 K to 220 K in the frequency range 445-495 GHz. 
The noise temperature increases sharply below 445 GHz, due to leakage of the RF- 
filter. 



Page 214 



Third International Symposium on Space Terahertz Technology 



SRON 410-490 GHZ 2-tuner mixer 

Q35 Rn-22 Ohms, 1/4 Xfilter, 2gm 2 
Q34 Rn-100 Ohms, Chebychev filter .8 urn 2 
2-junction array 



800 


DSB Noise Temperature 






□ 










700 




junction 

■■■□■■ Q35 




600 








A Q34 




500 








400 








300 








200 


□. 

" A- A.. A ..A. 

a -aaa- 


D 
.A- 


•□■•□ ■•□■ ^ .A-A--& 
A . A a A 

AA" 


100 
n . 


I I I I 


i. 


i i i i 



400 410 420 430 440 450 460 470 480 490 500 



Frequency (GHz) 



State University Groningen 
SRON Groningen, The Netherlands 



Fig. 2 



Third International Symposium on Space Terahertz Technology 



Page 215 



1200 



SRON 490 GHZ 2-tuner mixer 
Tuned and instantaneous bandwidth 
Q34 Rn-100 Ohms ,Chebychev-filter 

DSB Noise Temperature 




400 410 420 430 440 450 460 470 480 490 500 



Frequency (GHz) 



State University Groningen 
SRON Groningen, The Netherlands 



Fig. 3 



Page 216 



Third International Symposium on Space Terahertz Technology 



The instantaneous bandwidth of junction Q34 is shown in Fig. 3. Here the frequency 
was changed without adjusting the tuners, only the pump power was adjusted for 
optimum H/C response. 



The unpumped and pumped IV-curves of the two arrays at two different frequencies 
are shown in Fig. 4. Array Q34 has a low leakage current, a gap voltage of 17 mV 
(1 junction) and a well defined photon step above the gap. The gap voltage of array 
Q35 is 2.4 mV and it is clearly observed that if the LO-power is radiated on the 
junction the gap voltage decreases due to heating. The shape of the photon step 
above the gap also indicates heating effects occur in the junction. 



Pumped IV-curvea 410-490 GHz 
junction Q34 



DC current (uA) 



100 




Pumped IV-curves 430-490 GHz 
junction Q35 

DC current (uA) 



50- 



4 • • 10 a 
Bias voltage (mV) 




a 4 • 

Bias voltage (mV) 



Fig. 4 Pumped and unpumped IV-curves of the two junction arrays 



Fig. 5 shows the IF power output at two LO-frequencies, when a hot or cold load 
are placed in front of the junction. The smooth curves indicate that the Josephson 
current is suppressed sufficiently. The structure (at 3 mV) in the IF-output seen at 
495 GHz is due to the fact that the second photonstep from the negative bias 
voltage range "creeps" into the first photonstep at the positive voltage range. 



Third International Symposium on Space Terahertz Technology Page 217 

Hot/Cold response 410 GHz Hot/Cold response 495 GHz 



IF-output (uW) 




2 4 • t 10 12 

Bias voltage (mV) 



IF-output (uW) 




Biaa voltage (mV) 



Fig. 5 IF-output with hot and cold signal 

Analysis 

In the analysis of the noise temperatures, the receiver is divided into three 
elements: the RF-input, the mixer and the IF-output. A schematic diagram of the 
whole receiver is given in Fig. 6. Each of these elements contributes to the total 
receiever noise and gain. 



1 



Trf,Grf 



mix 



,g 



mix 



RF-input 



Mixer 



T/f,Gjf 



IF-output 



Fig. 6 Schematic representation of the noise and gain 

contributions in the receiver 

For the analysis of the contribution of the IF chain (T-bias, isolator, IF-amplifiers), 
the shotnoise of an unpumped junction is used. With the known shotnoise of an 
unpumped junction a Y-factor measurement on the IF-chain is performed. The total 
power at the end of the IF-chain is given by (1). Here r is the reflection coefficient 
between the 50 fi line and the junction. P^cP^ and P IF are the noise powers 



Page 218 



Third International Symposium on Space Terahertz Technology 



coming from the junction, the isolator load and the IF-amplifier. G^G^, and G T _ 
bias are the gains from the various components. The noisepower from an unpumped 
array of two junctions is given by (2), where B is the bandwidth, R dyn is the dynamic 
resistance of the array and V is the voltage over the entire array. 

PlF 



Pout — {Pjunc^ + -Pjso/(1 — T) + 



*- r isol*- r T— bia: 



■)G[FGi so iGT-bias (1) 



Pjunc = £</?> Rdyn = l^eB coth( : ^ ; )I(V)R dyn 



8 



8 



Ak B T 



(2) 



IF-noise of unpumped junction 
IF-output (uW) 




4 a 8 10 12 

Bias voltage (mV) 




T-Bias 



Isolator 



amplifier 



Schematic Diagram of junction and IF- 
chain 



Fig. 7 Measured and calculated 

noisepower output of an unpumped 

array of two junctions 

Fig. 7 shows the experimental and fitted curves. The values for the gain and noise 
contributions of the IF-chain are: G, F =88.4 ±0.1 dB, Tj F =4.8 ± 0.2 K. 
The gain and noise contributions of the RF input (beamsplitter, HDP-window, 
quartz-filter, lens, horn, waveguide and tuners) are difficult to estimate. Several 
elements were measured seperately, but reflections at the horn waveguide transition 
and losses in the waveguide and thf two tuners are difficult to find. The total gain 
and noise at the RF-input are: Grf=0.77 ±0.1 and ^=56 ± 20 K. 



Third International Symposium on Space Terahertz Technology 



Page 219 



The mixer noise and gain are now calculated with (3) and (4), where S? ut is 
measured power difference at the IF-frequency and 5P in is the difference in input 
power from the hot and the cold load. T^ is the measured total receiver noise 
temperature. 

SPout 



{Jmix — 



GRFGrF^Pi 



(3) 



n 



Tmix = T rec GRF — TrfGrf — 



Tip 



(4) 



mix 



The gain and noise of the mixer are, just as the receiver noise temperature, nearly 
constant over the 400-500 GHz band. Typical values for the contributions in the 
receiver DSB gain and noise are: G mix = -12.5 dB ± 0.6 dB and T mix = 90 ± 30 K. 
For a complete calculation of the mixer performance it is necessary to know the 
embedding admittances at the LO and the upper and lower sideband frequencies. 
These admittances are found by fitting a calculated pumped IV-curve to a measured 
pumped IV-curve. An example of the quality of this fit at two different frequencies 
is shown in Fig. 8. 



Pumped IV-curve 410 GHz 
G-0.67 B-0.42 (norm, to 1/100 Mho) 

DC current (uA) 



Pumped IV-curve 490 GHz 
00.60 B-0.31 (norm, to 1/100 Mho) 

DC current (uA) 



too - 



so 











oatautatad 

■aaauraa' 




- 







100 ■ 



•0 











aaaaured 




- 







IS 



Bias voltage (mV) Bias voltage (mV) 

Fig. 8 Measured and calculated IV-curves. The embedding parameters 
are shown in the header 



Page 220 



Third International Symposium on Space Terahertz Technology 



It is observed that the quality of the fit is good, except for the discrepancy near the 
gap, which is due to heating and difficult to model. The derived embedding 
admittances indicate that the tuning elements are able to compensate the junction 
capacitance. Unfortunately the sideband admittances are not yet calculated and a 
full analysis cannot be performed at the moment. The result of a calculation of the. 
mixergain, under the assumption that the sideband admittances are equal to the 
LO-admitance, is shown in Fig. 10. 



Calculated Gain 410 GHz 
G-0.67 B-0.42 (norm, to 1/100 Mho) 

Gain 













0.12 


= /l 


■■— urad 




0.1 


■ n\ 








0.0t 


■ \ \ 




0,08 


■ 1 \ 




0.04 


■ I \ 




0.02 


■ I \ 




n 


. J. \l^c\- . . 



0,6 1 1,8 2 

Bias voltage (mV) 



2.8 



Calculated gain 490 GHz 
00.60 B-0.31 (norm, to 1/100 Mho) 



Gain 



0.08 



0.0* 




0.04 - 



0,02 



0,8 1 1,8 2 

Bias voltage (mV) 



2.8 



Fig. 10 Measured and calculated mixer gain 

In this figure the calculated and measured gain are normalized to each other. One 
observes a big discrepancy between the measured and calculated gain, which 
indicates that the LO and sideband frequencies differ significantly. 
In both the calculated and the measured gain, some fine structure on the first 
photonstep region is observed, indicating that the calculation method is working 
properly, but the input parameters are wrong. 



Summary 

Measurements were performed in the 400-500 GHz range with a two tuner 
waveguide mixer. The measured (receiver) noise temperatures are amongst the 
lowest values measured at these frequencies. The results show that an array of two 



Third International Symposium on Space Terahertz Technology Page 221 

junctions is suitable in achieving a low noise receiver in the 400-500 GHz range. It 
is also found that a qualitatively "bad" junction with a low gap-voltage can still serve 
as a low noise mixer element. The preliminary comparisons between theory and 
measurement show a good agreement between calculated and measured pumped 
IV-curves. Gain calculations indicate that the measured noise temperatures are not 
fully DSB measurements. Further analysis is needed to determine the USB and LSB 
gain and noise contributions. 

Acknowledgements: 

This work was supported by ESA under contract No. 7898/88/NL/PB(SC), the 
Stichting Technische Wetenschappen and the Stichting voor Fundamenteel 
Onderzoek der Materie. 

References 

1 J.Zmuidzinas, H.G. LeDuc, "Quasi-Optical Slot Antenna SIS Mixer", Proceedings 
of the Second International Symposium on THz Technology. 

2 C.K. Walker, M.Chen, P.L Shafer, H.G. LeDuc, J.E. Carlstrom, T.G. Carlstrom, 
T.G. Phillips, "A 492 GHz SIS Waveguid Receiver for Submillimeter Astronomy", 
Int J. of IR and Millimeter Waves 1992. 

3 J.R. Tucker, M.J. Feldman, "Quantum Detection at Millimeter Wavelengths" Rev. 
Mod. Phys 57, 1055 (1985) 

4 C.A. Mears, Qing Hu, P.L. Richards, A.H. Worsham, D.E. Prober, A.V. Raisanen, 
"Quantum Limited Quasiparticles Mixers at 100 GHz", IEEE Trans. Magrt, vol 27, 
2, 1991 

5 C.E. Honingh, G. de Lange, M.M.T.M Dierichs, H.H.A. Schaeffer, J. Wezelman, 
J. v.d. Kuur, Th. de Graauw, T.M. Klapwijk, "Comparison of Measured and 
Predicted Performance of a SIS Waveguide Mixer at 345 GHz",. these proceedings 

6 C.E. Honingh, unpublished results 

7 M.M.T.M. Dierichs, unpublished results 



Page 222 Third International Symposium on Space Terahertz Technology 

^3~£3 N9 3 -2? 744 

/£0£)32^~ Double Dipole Antenna SIS Receivers at 100 and 400 GHz 



\* 



A. Skalare* »', H. van de Stadt", Th. de Graauw", R. A. Panhuyzen"**, 
M. M. T. M. Dierichs'" 



* Dept. of Applied Electron Physics, Rannvagen 6, 
Chalmers University of Technology, G5teborg, Sweden. 

** Space Research Organization of the Netherlands (SRON), 
Landleven 12, 9747 AD Groningen, the Netherlands. 

* " " Dept. of Applied Physics and Materials Science Center, University of Groningen, 
Nijenborgh 4, 9747 AG Groningen, the Netherlands 



Abstract 

Antenna patterns were measured between 95 and 120 GHz for a double dipole antenna 
/ ellipsoidal lens combination. The structure produces a non-astigmatic beam with low 
side lobe levels over that whole band. A heterodyne SIS receiver based on this concept 
gave a best noise temperature of 145K DSB at 98 GHz. Measurements were also made 
with a 400 GHz heterodyne SIS receiver, using a double dipole antenna in conjunction 
with a hyperhemispherical lens. The best noise temperature was 220 K DSB at 402 GHz. 
On-chip stubs were used to tune out the SIS junction capacitance. 



Introduction 

We here describe two SIS heterodyne receivers, both using double dipole antennas 
[1,2,3,4] placed on the back plane of a thick dielectric lens. In both cases, the antenna 
consisted of two half wave dipoles, connected by a stripline with the SIS mixer at the mid 
point, Fig.l . The quartz chip with the antenna was mounted on the back plane of a thick 
quartz. lens, and was backed by a quarter wave thick quartz slab and a reflector, Fig.2 . 
Some early low frequency scale model measurements of this structure can be found in 
[4]. 

Both receivers were designed in similar but not identical ways, and will be described 
separately. 



100 GHz Receiver Design 

The size of the dipole antenna was chosen to give a center frequency of 100 GHz. The 
11mm diameter quartz lens was polished to an ellipsoidal shape, designed to produce a 



Third International Symposium on Space Terahertz Technology Page 223 



diffraction limited main lobe. The polishing tool was a brass rod with an ellipsoidal hole 
at one end, machined with a numerically controlled milling machine. 

As shown in Fig.3 the mixer fixture pinched the lens between a copper back plate and 
two flanges, which fitted into two grooves in the quartz. With the help of a small amount 
of vacuum grease, this provided excellent 4K cooling of the lens and the mixer chip. 

The mixer itself was a series array of two Nb-Al/AlOx-Nb SIS junctions with a normal 
state resistance of 34 ohms. The size of each junction was 2 square microns. No attempts 
were made to tune out the parasitic capacitance of the junctions. 

The mixer chip was contacted by a flexible strip transmission line, Fig.4 , which was 
soldered to a 85 mil output co-ax. The strip was cut to high accuracy from a 
Kapton/copper laminated sheet, to give a characteristic impedance close to 50 ohms. As 
shown in Fig.5 , the strip line would work well even for intermediate frequencies of up 
to 8 GHz, where resonances in the lens fixture begin to appear. 



100 GHz Measurements 

The antenna pattern of the dipole/lens fixture was measured at room temperature with 
a bismuth bolometer in place of the SIS junctions, Fig.6 . Both the E- and H-plane beam 
profiles were of high quality over the whole 95-120 GHz band, in good agreement with 
earlier scale model measurements [4]. 

Y-factor measurements with the SIS mixer in a 4.2K helium cryostat gave a best receiver 
noise temperature of 145K DSB at 98 GHz, Fig.7 . The intermediate frequency amplifier 
was a 1.5 GHz cooled Berkshire HEMT, with a noise temperature of 4K (from the 
manufacturers data sheets). We believe that the noise performance can be improved 
considerably by the use of on-chip tuning structures, but have not yet implemented this 
in the 100 GHz receiver. 



400 GHz Receiver Design 

The 400 GHz receiver differed from the 100 GHz one in a few ways, namely : 

1. We chose to use a hyperhemisperical lens, mainly because it can be manufactured 
to higher tolerances than an ellipsoidal one. 

2. The antenna was a scaled down version of the 100 GHz structure. The center 
frequency was chosen to 310 GHz, which puts the upper frequency limit of the antenna 
itself somewhere around 430 GHz (one octave bandwidth). 

3. The SIS parasitic capacitances were tuned by on-chip stubs, as can be seen in 
Figs.8 and 9 . A series array of two junctions, each of 3 square microns, was used. 

4. We used two different IF amplifier chains, one with a 1.5 GHz FET with 
approximately UK noise temperature, the other with a 4 GHz Berkshire Technology 



Page 224 Third International Symposium on Space Terahertz Technology 

HEMT with 4K noise temperature. 

5. The Josephson effect was suppressed by a superconducting coil. 

400 GHz Measurements 

The video response of the receiver over the range 100-600 GHz was studied with a 
Fourier transform spectrometer, Fig. 10 . Two peaks are visible in the diagram, one just 
above 200 GHz and one close to 400 GHz. The approximate positions of the peaks could 
be predicted from a simple circuit model, where each junction was represented by a 
capacitance of 165 fF in parallel with its normal state resistance. 

Two initial Y-factor measurements with the same mixer chip are shown in Fig. 11 . The 
best performance was at the lower end of the available local oscillator band, with a 
lowest noise temperature of 220K DSB. 

Summary 

The dipole / ellipsoidal lens configuration was investigated in terms both of antenna 
pattern and of matching to SIS junctions. The pattern measurements showed low side 
lobe levels, and a non-astigmatic beam over the whole band 95-120 GHz. The best 
receiver noise temperature was 145K DSB at 98 GHz, a value we believe will be 
improved with the use of integrated tuning structures. 

The initial measurements with the other receiver, in which a double dipole antenna is 
combined with a hyperhemispherical lens, yielded a best noise temperature of 220K 
DSB at 402 GHz. 

The Kapton laminate strip should function well up to 8 GHz as an intermediate 
frequency connection to the mixer chips. 

Acknowledgements 

The authors extend their gratitude to Prof Dr Ir T. M. Klapwijk for his advice and for 
useful discussions, to Mr. H. Schaeffer for his technical support, and to Mr. G. de Lange 
and Ms. C. E. Honingh for their assistance in the noise temperature measurements. The 
work presented here was supported financially by the European Space Agency (ESA), 
through contract 7898/88/NL/PB(SC). 

References 

[1] P. T. Parrish, T. C. L. G. Sollner, R. H. Mathews, H. R. Fetterman, C. D. Parker, 
P. E. Tannenwald, A. G. Cardiasmenos, "Printed Dipole-Schottky Diode Millimeter 
Wave Antenna Array", SPIE Millimeter Wave Technology, Vol. 337, 1982, pp.49-52 



Third International Symposium on Space Terahertz Technology Page 225 



[2] W. Chew, H. R. Fetterman, "Printed Circuit Antennas with Integrated FET 
Detectors for mm- Wave Quasi-Optics", IEEE Trans. Microwave Theory Tech., Vol. 
MTT-37, No. 3, 1989. 

[3] J. A. Taylor, T. C. L. G. Sollner, D. D. Parker, J. A. Calviello, "Planar Dipole-fed 
Mixer Arrays for Imaging at Millimeter and Sub-Millimeter Wavelenghts", Proc. of the 
1985 Int. Conf. on IR and mm-Waves, 1985, pp.197-188. 

[4] A. Skalare, Th. de Graauw, H. van de Stadt, "A Planar Dipole Array Antenna 
with an Elliptical Lens", Microwave and Optical Tech. Letters, Vol.4, No.l, Jan. 1991. 



Page 226 



Third International Symposium on Space Terahertz Technology 



.4 



^ 



.25 



Fig.l : The geometry of the 100 GHz double dipole antenna. The dimensions are in units 
of wavelength at the design frequency. 





Fig. 2 : The antenna chip is placed between a quartz lens and a quarter wavelength thick 
quartz slab with a reflector. 



Third International Symposium on Space Terahertz Technology 



Page 227 



Front Plate 
(Copper) 



Quartz 
Lens 



PTFE 
Screw 



Back Plate 
(Copper) 



Flanges 



Cooling 
Strap 




Side View 



Back View 



Fig.3 : The fixture that holds the lens and the mixer chip. The diameter of the lens in 
this figure is 11mm. 



85 mil co-ax 

(50 Q) Solder 



Strip Line 



Contacting Tabs 
(Copper foil) 



Mixer Chip 




Fig.4 : The flexible Kapton laminate strip line used for the DC & IF connections. The 
tabs at the end of the strip are glued to the contact pads on th< chip with silver paint. 



I 

00 



CM 1 R u 



1 U FS H 06.737 O 
riAnCH 5 1992 SHORT. 



-1,8004 O 20,997 pF 
CHIP RES. 



Cor 
Del 



Gnt 
Hid 




START .130 OOO 000 GHz STOP 20.000 000 000 GHz 

Fig.5 : The room temperature Sll reflection on the IF line in Figs. 3 & 4 with the mixer 
chip replaced with a chip resistor (51 ohms). A time gate was applied around the co-ax 
to strip transition, the strip itself and the resistor. Marker 1 is at 4 GHz and Marker 2 
is at 8 GHz. 



a- 
■v 



3 
a 



to 

3 

o 
</> 

£" 

3 

o 

a 

CO 

-a 
& 

r> 



3 

a- 

a 

5" 

88 



3- 
-«° 

a. 




E 


E 


E 


E 


E 


E 


95 GHz 


100 GHz 


105 GHz 


110 GHz 


115 GHz 


120 GHz 


H 


H 


H 


H 


H 


H 





3 



e 
3 
o 

3 

tj 
& 

r> 

J? 

& 

3- 

N 

«? 

3- 
3 
O 



Fig.6 : Antenna patterns of the double dipole / ellipsoidal lens combination. The 
patterns were measured with a Bismuth bolometer, and the radial scale is 5 dB per 
division. 



5 

K> 
to 



400 



300 



I 

o 



Q) 
(0 



Z 

m 

CO 
Q 



200 




100 







-I — I — I — I — I — I — I — I — I I I I I I ■ I ■ 



J I l_ 



95 



100 



105 



1 10 



1 15 



120 



125 



3 



3 

sa 
S" 

3 

a 

•—1 

en 

3 

© 

«■•** 

C 

3 
o 

3 
C/> 

a 

ro 
re 



Frequency (GHz) 

Fig.7 : The double sideband noise temperature of the 100 GHz receiver. The data was 
corrected for the transmission of the LO injection beam splitter (95%) . 



3 

o 

3 
O 



£ 



Third International Symposium on Space Terahertz Technology 



Page 231 




Fig.8 : The 400 GHz mixer chip. 




Fig.9 : The 400 GHz antenna chip, detail. 



CARC.1 



CARC. 2 



2 s 




100 



200 



300 



400 



500 



600 GHz 



Fig. 10 : The Fourier transform spectrogram (arbitrary units on the vertical scale). 
"CARC2" marks the frequency range where noise temperature measurements have been 
made. 



3 



3 

S' 



in 

3 
o 

3 
.CO 

5 

r> 
re 



3 

3 
O 

5* 

J5 



a* 



1500 



¥ 1000 



Q) 
CA 



z 
m 

0) 

Q 500 




4.3 GHz IF, T IF =4K 



re 

a 

& 

S° 

3 



tn 



Q 
(n 

5' 
3 

o 
a 

</> 

& 

Cl 

re 
H 

3 



3- 

3 

D 



o L — 

400 



420 



440 



460 



480 



500 



Frequency (GHz) 

Fig.ll : DSB Noise measurement with the 400 GHz receiver. The data was corrected for 
the transmission of the LO injection beam splitter (65%) . 



S 3 

00 
re 



/ 



Page 234 Third International Symposium on Space Terahertz Technology 

S/6)—£ 2--" Slot Antenna SIS Mixers for Submillimeter Wavelengths 

/(p 533 Jonas Zmuidzinas*, H. G. LeDuc**, and J. A. Stern^j Q 3 * ^ ^ ^ 

\ ^ 

s^' We are developing improved versions of a slot antenna SIS mixer which we have 
previously described [1]. The initial work demonstrated a double sideband noise 
temperature of 420 K for a 500 GHz quasi-optical SIS mixer employing a twin-slot antenna 
on a quartz dielectric substrate. A quartz hyperhemispherical lens is used to focus the 
incoming radiation onto the twin-slot antenna. The advantages of a twin-slot antenna 
include a low impedance (35 CI) and a clean, symmetric beam pattern into the dielectric with 
a 70% efficiency. In our original mixer, the radiation was coupled from the two slot 
antennas onto superconducting microstrip lines which fed the SIS junction. By performing 
an impedance transformation using tapered lines and by feeding the radiation from the two 
slots to the junction in parallel, the effective (real) impedance seen by the junction was 
reduced to just 4 Q. This very low impedance allowed a junction area of 2.3 (im 2 to be 
used at 500 GHz, which was manufactured using optical lithography. However, no 
attempt was made to tune out the junction capacitance. We estimate that this capacitance 
reduces the impedance coupling efficiency to T|z = 0.23, for our junction with cdRn C = 5.3 
at 500 GHz. 

The recent development [2] of techniques using electron-beam lithography to 
manufacture junctions with very small areas (= 0.1 pirn 2 ) now allows considerably more 
flexibility in the design of SIS mixer circuits. We have redesigned the slot-antenna mixer 
to take advantage of this possibility. In particular, we have included a novel circuit which 
allows the junction capacitance to be tune out over a broad bandwidth. For instance, 
mixers designed for 800 GHz using NbN/MgO/NbN junctions with realistic parameters 
achieve a 3 dB impedance bandwidth of nearly 400 GHz. Furthermore, our circuit uses 
only short lengths of microstrip and should therefore be less sensitive to RF losses than 
other designs. The improved impedance match should give a large reduction in noise 
temperature as compared to our previous mixer. The new devices are currently under 
fabrication. Further details of the design and any available experimental results will be 
discussed. 



*Jonas Zmuidzinas is with the Downs Laboratory of Physics, California Institute of 
Technology. 

**H. G. LeDuc and J. A. Stern are with the Jet Propulsion Laboratory, California 
Institute of Technology. 

Wj. Zmuidzinas and H. G. LeDuc, "Quasi-optical Slot Antenna SIS Mixers," IEEE 
Trans. Microwave Theory Tech., in press, 1992. 

[2lH. G. Leduc, A. Judas, S. R. Cypher, B. Bumble, B. D. Hunt, and J. A. Stern, 
"Submicron Area NbN/MgO/NbN Tunnel Junctions for SIS mixer Applications," IEEE 
Trans. Magn., MAG-27, 3192, 1991. 



Third International Symposium on Space Terahertz Technology Page 235 

A PLANAR QUASI-OPTICAL SIS RECEIVER ~ 

p 7 

FOR ARRAY APPLICATIONS ' ~ 

Philip A. Stimson, Robert J. Dengler, Peter H. Siegel and Henry G. LeDuc 
Jet Propulsion Laboratory, Pasadena, CA, 91109 



Abstract — A novel planar, quasi-optical SIS receiver operating at 230 GHz is 
described. The receiver consists of a 2x5 array of half wave dipole antennas 
with ten niobium-aluminum oxide— niobium SIS junctions on a quartz dielec- 
tric-filled parabola. The 1.4 GHz intermediate frequency is coupled from the 
mixer via coplanar strip transmission lines and 4:1 balun transformers. The 
receiver is operated at 4.2 K in a liquid helium immersion cryostat. We report 
here accurate measurements of the performance of single receiver elements. 
A mixer noise temperature of 89 K DSB, receiver noise temperature of 156 K 
DSB and conversion loss of 8 dB into a matched load have been obtained. 



INTRODUCTION 

The quasiparticle superconductor-insulator-superconductor (SIS) mixer is the most 
sensitive detector in the millimeter-wave region and forms the basis of most high quality 
receivers for millimeter- wave astronomy [l]. The quantum limit for noise temperature (in 
a SSB mixer) has essentially been reached at 100 GHz [2] [6], but at higher frequencies the 
available performance is poorer, with 10 times the quantum limit being a more realistic 
goal. This figure has recently been reported from the best waveguide mixers around 
200 GHz [3] [4] [5]. The major cause of the performance reduction at high frequency is 
the SIS junction capacitance, which presents a smaller parallel reactance and shunts 
the quasiparticle response. Tuning structures can, in principle, alleviate this limitation 
but are not yet well understood at higher frequencies [8]. The approach most often 
used, and that used here, is to fabricate high current density junctions with exceedingly 
small areas (< 1 /im 2 ) to reduce the capacitance. Another serious problem is control of 
Josephson currents in the junction. Noise temperatures obtained with broadband hot 
and cold loads may be highly inaccurate in the presence of Josephson currents. These 
effects become more important as the frequency and/or bandwidth is increased. Other 



Page 236 



Third International Symposium on Space Terahertz Technology 



problems include losses in conductors and dielectrics, the fabrication difficulties of small 
waveguide components and difficulties in obtaining convenient local oscillators. 

Quasi-optical receivers with planar circuit mixers are an attractive approach for sys- 
tems at frequencies in the neighborhood of 1 THz [7] [8] [9]. They suffer the disadvantage 
of being fixed tuned but provide the advantage of convenient monolithic fabrication. 
Planar configurations are also a desirable approach to realizing array receivers. 

In this paper, we report accurate measurements on a quasi-optical array-type receiver 
at 230 GHz. We have been able to suppress Josephson currents almost completely, and our 
intermediate frequency versus bias voltage curve exhibits the smooth oscillatory behavior 
of the best waveguide mixers [3]. Our configuration is designed to allow an array of mixers 
to be measured during one cool down cycle. We report here the performance of a single 
array element. We will report on complete array performance in a separate paper. The 
SIS junctions used for these experiments were nominally identical to those used in recent 
waveguide receivers [3] [4], with which our results may be compared. 



SIS JUNCTION FABRICATION 

The junction wafer used for this receiver carries a 2x5 array of resonant dipole an- 
tennas with 0.4x0.4 //m niobium-aluminum oxide-niobium SIS junctions at the termi- 
nals. The junctions were fabricated using a self aligned lift-off trilayer process. The 
niobium-aluminum oxide-niobium trilayer was sputtered onto the 0.25 mm thick, 17 mm 
diameter quartz substrate through a photoresist stencil. The trilayer remaining af- 




Fig. 1. The mixer block with the upper half removed. The central dielectric-filled parabola (dark), 
containing 10 antenna and mixer elements, is surrounded by 10 IF baluns (light) and SSMA connectors 
at the edge of the block. 



Third International Symposium on Space Terahertz Technology 



Page 237 



ter lift-off formed half of each antenna and the ten coplanar strip transmission lines 
used for the. IF. The junction mesa was patterned using electron beam lithography on 
1200 A thick PMMA over a 4000 A thick polyimide layer, followed by evaporation of 
500 A of chromium metal and lift-off. The chromium stencil was transferred to the poly- 
imide underlayer by reactive ion etching in an oxygen plasma. The contact regions of the 
trilayer were then protected with a resist stencil and the chromium/polyimide mask was 
used to etch the junction. Thermal SiO was deposited using the same stencil to provide 
electrical isolation of the base electrode and to provide dielectric for two RF blocking 
capacitors located one quarter and three quarter wavelengths away from the junction 
down the coplanar strips. The polyimide was then removed with dichloromethane. The 
second half of the antennas was made by deposition of niobium and reactive ion etching. 



RECEIVER DESIGN 

The mixer block, shown in Figure 1, consists of the junction/antenna wafer, a quartz 
reflector, and IF baluns and connectors mounted in a brass housing. The wafer is held 
on the flat face of a quartz parabolic lens, whose rear surface is metalized. Incoming 
radiation is reflected by the metal surface and focussed onto the antenna elements at 
the center of the wafer. The configuration, called a Dielectric-Filled Parabola (DFP), is 
analogous to a conventional parabolic dish antenna. The IF signals are coupled from the 
wafer via coplanar strip transmission lines. Monolithic IF baluns transform the 200 $7 
characteristic impedance of the coplanar strips to that of 50 fl coaxial transmission line. 
Details of this design, including extensive low frequency modeling, are described by Siegel 
et al. [10] [16]. A superconducting magnetic field coil is mounted on the block to suppress 
Josephson currents in the junctions. 



chopper -i 



mixer block 



12 IF cobles 




bias 
to flange 



20 dB coupler isolator 



5Z 




to flange 



atten 



IF amplifier 
38dB 



© 



Voltage tuned 
oscillator 



variable temp 
IF load 



Fig. 2. Schematic diagram of the IF system of the array receiver. The entire system is immersed in 
liquid helium except for the 77 K load which is bolted to the liquid nitrogen shield of the cryostat. 



Page 238 



Third International Symposium on Space Terahertz Technology 



The IF system shown in Figure 2, consists of ten IF cables routed through two 6- 
position coaxial switches and one 2-position switch to a single amplifier chain. The 
remaining two switch positions are used to connect a short and a variable temperature 
IF load to the amplifier input. The load consists of a resistor terminating a stainless 
steel coax cable on a thermally isolated plate which contains a heater resistor and diode 
thermometer. The structure is enclosed in an indium sealed can. This permits accurate 
calibration of the IF system and very accurate mixer measurements [ll]. An isolator 
is used to reduce the SWR at the amplifier input and a directional coupler with cooled 
attenuators allows signals to be injected into the IF system to measure the mixer reflection 
coefficient. After removal from the cryostat the IF signal is further amplified and passed 
through a variable center frequency 50 MHz wide filter and fed to a power detector. The 
IF system noise temperature is approximately 7K at 1.4 GHz. 

The optical system consists of a chopper mounted directly in front of the mixer, and 
the hot and cold loads. When the chopper blade is closed the input beam is directed 
onto a 4 K (cold) load mounted on the receiver plate; when it is open the beam passes 
through a quartz window to a 77 K (hot) load mounted on the liquid nitrogen shield of the 
cryostat. The loads are pyramidal absorbers manufactured from Eccosorb CR-110, which 
is known to provide high absorption and low reflection at this frequency. Reflection from 
a flat plate of CR-110 has been measured at less than -10 dB in this frequency range [12]. 
The window is exactly five wavelengths thick and passes almost all the incident 230 GHz 
radiation. The theoretical transmittance is 0.999; we measured a transmittance of over 
0.95. Local oscillator radiation is produced by a Gunn diode and Schottky diode doubler 
and is injected through the back of the mixer block. No diplexer is required. 

The entire receiver is immersed in liquid helium which eliminates heat sinking prob- 
lems. The dielectric constant of the helium is 1.048 [15]. The switches, thermometers, 






a -o.oo 

0) 



3 
U 




' ' ' ' I I I— I I— I I — I — L- 




Pumped 



Unpumped 



-2024 

Voltage (mV) 

Fig. 3. Pumped and unpumped IV curves for a typical Nb-A10 r -Nb SIS junction used in the planar 
receiver. 



Third International Symposium on Space Terahertz Technology 



Page 239 



liquid level meter.. .etc., and all data aquisition is controlled by a computer. 



MEASUREMENT TECHNIQUE 

We use a variation of the technique of McGrath et al. [ll], to obtain mixer gain and 
noise temperature. First, the IF system is calibrated by plotting the temperature of 
the IF load as a function of the IF output power. This measures the IF system noise 
temperature Tip. The receiver noise temperature Tr is measured using the hot and cold 
loads (Th and Tc), the ratio of the IF output powers Y = Pifh/Pifc and Equation 1. 



Tr = 



Th-YTc 
Y-l 



T T (Txp + Tsr 2 ) 



Lm V 



(1-P) 

T ifh -TifcA / l ~7 2 
Th-Tc )\l-V\ 



(1) 
(2) 

(3) 



Next, the temperatures of the IF load, Tifh and TjfC) which produce output powers 
Pifh and Pifc are calculated from the calibration, and the effective bath temperature 
Ts determined by measuring the power output from the IF system with a shorted input. 
The IF reflection coefficient of the mixer T (and of the load 7) is measured by injecting 
a signal from a voltage tuned oscillator through the coupler and recording the difference 
in reflection between the mixer and the short. The loss into a matched load and noise 
temperature are then calculated from Equations 2 and 3. 




-2 2 

Voltage (mV) 



Fig. 4. IF output power as a function of bias voltage for hot and cold load inputs. The curve exhibits a 
smooth oscillatory behavior similar to that expected from theory with no sharp spikes or discontinuities 
indicating excellent control of Josephson currents. 



Page 240 



Third International Symposium on Space Terahertz Technology 



RESULTS 

Typical pumped and unpumped IV characteristics are shown in Figure 3, and IF 
output power as a function of bias voltage for hot and cold load inputs is shown in 
Figure 4. A superconducting magnet was used to suppress Josephson currents. The curve 
exhibits a smooth oscillatory behavior similar to that expected from theory [13] [14] with 
no sharp spikes or discontinuities. The IF output power is expected to decline towards 
zero bias; the fact that there is some power output at zero bias indicates some remaining 
Josephson currents which were not fully suppressed. These remain visible on the IF curve 
even though the IV curve appears smooth. Nevertheless, we believe that this is the best 
IF behaviour reported from a planar quasi-optical SIS receiver. 

The most recent experiments performed with this receiver used junctions with an 
area of 0.2 /an 2 . The normal state resistance was 56 fi, the critical current density was 
15kAcm -2 and the ujRC product was approximately 1.3. The mixer and receiver noise 
temperatures and mixer conversion loss are plotted as a function of IF frequency in Fig- 
ure 5. The LO frequency was 230 GHz. The best results are obtained at 1.35 GHz where 
a Tm of 89 K DSB, a Tr of 156 K DSB and conversion losses of 8dB (into a matched 
load) were measured. The IF mismatch is approximately 1 dB across the IF band. Esti- 
mated uncertainties in the noise temperatures are ±5K, and in the loss, ±0.5 dB. These 
values neglect any uncertainty due to RF load reflections or beam spillover. The largest 
Y-factor was obtained on the first quasiparticle step below the energy gap, at a bias 
voltage of approximately 2.3 mV. An inferior Y-factor was noted on the second step. The 



500 



oj 400 

u 

cd 

U 300 



J 1 * 200 

en 
"o 100 




1.1 



1.2 



1.3 



1.4 



1.5 



Frequency (GHz) 



1.6 



1.7 



Fig. 5. Mixer and receiver noise temperatures and mixer loss as a function of IF frequency. The best 
results are obtained at 1.35 GHz where a Tm of 89 K DSB, a Tr of 156 K DSB and conversion losses of 
8 dB were measured. 



Third International Symposium on Space Terahertz Technology Page 241 



mixer noise temperature and conversion loss are seen to be essentially constant across 
the IF band. Mixer noise temperature is referred to the optically coupled loads at the 
system input and includes the effects of all components through to the IF connectors at 
the output of the balun transformers. The receiver noise temperature follows the noise 
behavior of the IF amplifier. 

At each data point on the curves, the change in IF reflection coefficient, and the 
change in bias point, caused by switching between the hot and cold loads was measured. 
This is necessary to ensure that the observed Y-factor is not produced by different LO 
pumping conditions, or change in bias point when observing the hot and cold loads. 
Different pumping would be expected to change the junction output impedance and the 
IV curve shape. The reflection coefficient change was verified to be less than 1 %, and 
the change in bias voltage less than 0.02 mV. This indicates that the observed Y-factor 
has no appreciable component due to these factors. 

Recent results from waveguide mixers at similar frequencies using junctions with 
similar specifications from the same fabrication process [3] [4] give mixer temperatures of 
48 K DSB and 60 K SSB and conversion losses of 2dB. Our noise temperature results, 
although a factor of two higher, are consistent with these values given the lack of tuning 
capability inherent in our planar circuit. 



CONCLUSION 

We have demonstrated a planar quasi-optical SIS mixer and low noise receiver which 
is suitable for array applications. Best performance of an individual element at 230 GHz 
was a mixer noise temperature of 89 K DSB, a receiver temperature of 156 K DSB and a 
conversion loss of 8 dB. The IF output shows a smooth variation with bias, indicating good 
control of Josephson currents. The noise results are consistent with recent measurements 
using similar junctions in waveguide receivers, and are only a factor of two higher. The 
conversion loss is rather large, but consistent with other planar mixer values. We will 
report on array performance in a future publication. 



ACKNOWLEDGEMENT 

We are extremely grateful for the constant assistance and encouragement of Dr. W.R. McGrath, 
without whom this work could not have been completed. We also thank Dr. H.H.S. Javadi and Dr. 
M.A. Frerking of JPL, and Dr. A.R. Kerr and Dr. S.K. Pan of NRAO for useful advice and discussions. 
We acknowledge the support of Mr. B. Bumble, Dr. J. Stern and Mr. S.R. Cypher on junction 
fabrication, Mr. H. Moham for fabricating the array mount, and Mr. R. McMillan for fabricating the 
quartz parabola. This work was carried out at the Jet Propulsion Laboratory, California Institute of 
Technology under contract with the National Aeronautics and Space Administration. 



Page 242 Third International Symposium on Space Terahertz Technology 



REFERENCES 

[I] P.L. Richards and Q. Hu, Proceedings of the IEEE, vol. 77, 8, pp. 1233-1245 (1989). 

[2] C.A. Mears, Q. Hu, P.L Richards, A.H. Worsham, D.E. Prober and A.V Raisanen, IEEE Transactions 
on Magnetics, vol. 27, 2, pp. 3363-3369 (1991). 

[3] W.R. McGrath, H.H.S. Javadi, S.R. Cypher, B. Bumble, B.D. Hunt and H.G. LeDuc, Second Inter- 
national Symposium on Space Terahertz Technology, Pasadena, CA, Feb. 26-28 (1991), pp. 423-428. 

[4] J.W. Kooi, M. Chan, T.G. Phillips, B. Bumble and H.G. LeDuc, Second International Symposium 
on Space Terahertz Technology, Pasadena, CA, Feb. 26-28 (1991), pp. 459-472. 

[5] A.W. Lichtenberger, D.M. Lea, A.C. Hicks, J.D. Prince, R. Densing, D. Petersen and B.S. Deaver, 
Second International Symposium on Space Terahertz Technology, Pasadena, CA, Feb. 26-28 (1991), pp. 
439-458. 

[6] S.K. Pan, A.R. Kerr, M.J; Feldman, A.W. Kleinsasser, J.W. Stasiak, R.L. Sandstrom and W.J. 
Gallagher, IEEE Transactions on Microwave Theory and Techniques, vol. 37, 3, pp. 580-592, (1989). 

[7] J. Zmuidzinas and H.G. LeDuc, Second International Symposium on Space Terahertz Technology, 
Pasadena, CA, Feb. 26-28 (1991), pp. 481-490. 

[8] Q. Hu, C.A. Mears, P.L. Richards and F.L. Lloyd, IEEE Transactions on Magnetics, vol. 25, 2, pp. 
1380-1383, (1989). 

[9] T.H. Biittgenbach, R.E. Miller, M.J. Wengler, D.M. Watson and T.G. Philips, IEEE Transactions on 
Microwave Theory and Techniques, vol. 36, 12, pp. 1720-1725 (1988). 

[10] P.H. Siegel and R.J. Dengler, IEEE Transactions on Antennas and Propagation, vol. 39, 1, pp. 
40-47 (1991). 

[II] W.R. McGrath, A.V. Raisanen and P.L. Richards, International Journal of Infrared and Millimeter 
Waves, vol. 7, 4, pp. 543-553 (1986). 

[12] J.B. Peterson and P.L. Richards, International Journal of Infrared and Millimeter Waves, vol. 5, p. 
1507, (1984). 

[13] J.R. Tucker and M.J. Feldman, Rev. Modern Physics, vol. 57, pp. 1055-1113 (1985). 

[14] J.R. Tucker, IEEE Journal of Quantum Electronics, vol. 15, 1234-1258 (1979). 

[15] "Handbook of Chemistry and Physics", CRC Press, 56'th ed. (1976), p. E-55. 

[16] P.H. Siegel, First International Symposium on Space Terahertz Technology, Ann Arbor, MI, Mar. 
5-6, (1990) pp. 218-227. 



Third International Symposium on Space Terahertz Technology Page 243 

N93-27747 

A Fixed Tuned Broadband Matching Structure for ~>£^/'~3^b 



Submillimeter SIS Receivers 

* 




Thomas H. BiittgenbadT, Henry G. LeDuc**. /fuS, ON J 

Paul D. Maker**, and T. G. Phillips* / £ £3£T 

A / 

We have designed, fabricated and tested a quasi optical spiral antenna mixer with a 
Nb/A10x/Nb tunnel junction. This design incorporates a hybrid antenna fed by a planar 
logarithmic spiral antenna to couple to the radiation field, as previously done with Pb based 
devices, as well as a newly designed matching circuit. This matching circuit is a relatively 
complex structure requiring several layers of photolithographic processing on top of the 
actual tunneling device. Computer modeling of the device predicted the measured 
bandwidth to within 8%, making scale model measurements unnecessary. We have 
obtained a good match from 210 GHz to 460 GHz between the antenna and a relatively 
large area (1.25 by 1.25 Jim^) tunnel junction with co Rn C ~ 2 - 4.4. This compares to 
simple inductive stubs that attain only a few percent of total bandwidth in the submillimeter 
band or inductively tuned SIS arrays with an upper limit of operating frequencies well 
below the submillimeter band. Noise temperatures were measured at 345 GHz, 426 GHz 
and 492 GHz yielding double sideband noise temperatures at 200 K, 220 K and 500 K, 
respectively. 



Thomas H. Biittgenbach and T. G. Phillips are with the Division of Physics, 
Mathematics and Astronomy, California Institute of Technology. 

Henry G. LeDuc and Paul D. Maker are with the Jet Propulsion Laboratory, 
California Institute of Technology. 



Page 244 Third International Symposium on Space Terahertz Technology 

N93 . 27748 



/6o£3<o 



[A 



Modelling and Performance of Nb SIS Mixers 
in the 1.3mm and 0.8mm Bands 

Karpov, A., Carter, M., Lazareff, B., Billon-Pierron, D., 

Gundlach, K.H. 

Institut de Radioastronomie Millimetrique (IRAM) 

300, Rue de la Piscine 
38406 ST MARTIN D'HERES Cedex (FRANCE) 



Abstract 

We describe the modelling and subsequent improvements of SIS waveguide 
mixers for the 200-270 and 330-370 GHz bands (Blundell, Carter, and Gund- 
lach 1988, Carter et ad 1991). These mixers are constructed for use in re- 
ceivers on IRAM radiotelescopes on Pico Veleta (Spain, Sierra Nevada) and 
Plateau de Bure (French Alps), and must meet specific requirements. 

The standard reduced height waveguide structure with suspended stripline is 
first analyzed and a model is validated through comparison with scale model 
and working scale measurements. In the first step, the intrinsic limitations 
of the standard mixer structure are identified, and the parameters are op- 
timized bearing in mind the radioastronomical applications. In the second 
step, inductive tuning of the junctions is introduced and optimized for min- 
imum noise and maximum bandwidth. In the 1.3mm band, a DSB receiver 
temperature of less than 110K (minimum 80K) is measured from 180 through 
260 GHz. In the 0.8mm band, a DSB receiver temperature of less than 250K 
(minimum 175K) is obtained between 325 and 355 GHz. All these results 
are obtained with room-temperature optics and a 4 GHz IF chain having a 
500 MHz bandwidth and a noise temperature of 14K. 

Design goals 

A receiver for radioastronomical use should meet specific design goals besides 
low noise at a particular frequency such as : reliability, ease of tuning, wide 
tuning range, capability for SSB tuning (increasingly important when the 



Third International Symposium on Space Terahertz Technology p age 245 



atmospheric radiation is a significant contribution to the system noise), good 
coupling to the antenna, wide IF bandwidth (especially for extragalactic 
work): 

Junctions 

The mixers described in the this report use Nb/A^C^/Nb junctions fabri- 
cated in the IRAM facility (Lehnert et al 1991) Two junction series arrays 
are fabricated with an integrated IF filter. Trilayers are deposited into resist 
stencils, followed by lift-off. This technique is thought to reduce the mechan- 
ical sress in the trilayer (Yuda, Kuroda, and Nakano 1987). The substrate 
is fused quartz 100/im thick. The base electrode is 130nm, the counter elec- 
trode 30nm, and the wiring layer 240nm thick. The junctions are isolated by 
anodisation, up to 10V, and a sputtered Si02 layer 200nm thick. 

Individual junction areas used in this work are about 2/im 2 , but progress in 
photolithography should allow use of smaller areas. The results presented 
here are obtained with 2-junction arrays having a total normal-state resis- 
tance Rn near 50ft. Some 4-junction arrays were also fabricated. 

The normal resistance of the junctions can be adjusted after fabrication by 
controlled thermal annealing (Lehnert et al 1992). 

Mixers currently in use on IRAM telescopes 

A simple equivalent circuit was used for the suspended-stripline, reduced- 
height waveguide mixer mount (Karpov et ad 1992). The values of the circuit 
elements were derived from electromagnetic theory. They were then vali- 
dated by measuring the embedding impedance of the pure SIS junction as 
measured via a coaxial probe on a scale model, and comparing the results 
with model predictions. Figure 1 shows the good agreement between mod- 
elled and measured values for the return loss. 

Figure 2 shows the measured DSB receiver noise. The degradation of re- 
ceiver noise at the band edges and in the vicinity of 220 GHz was caused by 
degradations of coupling efficiency at corresponding frequencies, which are 
intrinsic to the basic mixer structure. Because 220 GHz is an astronomically 
important frequency : 13 CO(2 — 1), it was decided to adjust one parameter of 
the mixer — the length of the last section of the suspended stripline filter — 
to shift this problem to a lower frequency; see fig. 3. 

Good agreement is found between predicted and measured values for several 
parameters : optimum backshort position (fig. 4), upper sideband rejection 



Page 246 



Third International Symposium on Space Terahertz Technology 



DSSll! 




i — r 

1.7S 



-3.000 - 



-14.00. 



0O.00 



i\nHi 



\T 



100K0' F7EJ-GHZ 120.0 



■ SU 



a ooo 




-e.000 



mm. a mix/ 



XbcaiiS mm 2.0 



-U00 



wnir 



BO. 00 



100.0 FSEO-GnZ 



120.0 



Figure 1. Comparison of measured and computed values 
for the return loss between the embedding circuit and a 50fi 
junction. Frequencies have been scaled to the 3mm range. 



200 



150 



100 - 



50 




IF=(3.7-4.3)GHz T ampl=14 K 



200 220 240 

LOCAL OSCILLATOR FREQUENCY (CHj) 



260 



o SO 




220 230 2<0 

LOCAL OSCIUAIOR mtOUENCY (GHi) 



Figure 2. DSB receiver noise ver- 
sus LO frequency for the original 
1.3mm mixer. 



Figure 3. Same as Fig. 2 for mod- 
ified mixer. 



Third International Symposium on Space Terahertz Technology 



Page 247 



versus backshort position (fig. 5), DSB receiver noise versus backshort posi- 
tion for fixed LO frequency (fig. 6). Note that at 230GHz, an SSB receiver 
temperature of 135K results when the mixer is tuned for 10 dB rejection of 
the USB, versus 215K SSB when tuned DSB. 



0.5 - 





_^_ 


1 ' ' *^" " 

MEASURED 


""^sC^*,^ 


1 ' ' F—l 1 ' *" 


- 




CALCULATED 




^-.^ 


- 


--v 


_ 










1 . 




~~^..„^ 



210 



220 



230 
FREQUENCY 



240 
(GHz ) 



250 



Figure 4. Backshort positions for 
optimum performance at each fre- 
quency. Comparison of predicted 
and measured values. 



-20 




F=231 GHz 



0.J 0.4 0.5 

BACKSHORT POSITION (MM) 

Figure 5. USB rejection versus 
backshort position. Comparison of 
predicted and measured values. 



o.s 



o 



J00 



200 - 



100 



0.2 



— . — ' — • — • — i — ' — • — • — > — r — ' — '" • 
MEASURED 




MODEL 


i ' - 


"•-'■-^J/ 


. 


F=231 GHZ 



0.3 0.4 0.5 

8ACKSH0RT POSITION (MM) 



0.6 



Figure 6. DSB receiver tem- 
perature versus backshort position. 
Comparison of predicted and mea- 
sured values. 



Next generation receivers in the laboratory. 



The large capacitance of SIS junctions causes a large mismatch, especially at 
the higher frequencies. A reactive tuning structure with a high transforma- 
tion ratio is needed. A backshort can in principle accomplish this, but only 



Page 248 



Third International Symposium on Space Terahertz Technology 



over a limited frequency range (unless one is ready to accept the complication 
of a two-backshort structure), and the performance is critically sensitive to 
the backshort losses. 

Local tuning with superconducting circuit elements overcomes these limita- 
tions. With the junction capacitance tuned out at least approximately over 
the frequency range, the demand on backshort reactive tuning is considerably 
diminished, and better performance can be obtained. We have designed op- 
timized tuning structures for the 1.3mm and 0.8mm bands. Figure 7 shows 
the predicted mismatch loss and the measured laboratory performance for 
the 1.3mm receiver. Figure 8 shows the same quantities for the 0.8mm re- 
ceiver. Both receivers were measured with a Mach-Zender diplexer for LO 
injection, a room- temperature lens producing a beam matched to the f/10 
Nasmyth focus of the IRAM 30-M telescope, and a 4 GHz IF chain having a 
14 K input noise temp-erature. We plan to improve these receivers by using 
cold optics. 



200 



150 



100 



MEASURED 
CALCULATED 



-*^*w\--- 




// 
/ 



lF=(3.7-4.3)GHz Tampl=14K 



400 



300 



200 




IF=(3.7-4.3)GHz, T ampl=14 K 



ISO 200 220 2«0 260 280 
LOCAL OSCILLATOR FRECUENCY (GHz) 



320 330 .340 350 ' 360 
LOCAL OSCILLATOR FREQUENCY (CHz) 



Figure 7. DSB receiver noise ver- 
sus LO frequency for 1.3mm mixer 
with inductive compensation. Com- 
parison of predicted and measured 
values. 



Figure 8. Same as Fig. 7, for ihe 
0.8mm receiver. 



Figure 9 summarizes the performance of receivers now operating on the 
IRAM telescopes, and of improved receivers being developed in the labora- 
tory. Figure 10 illustrates the discovery of Aluminum fluoride in the evolved 
star IRC-f-10216, made at the IRAm 30-M telescope with one of the two 
1.3mm SIS receivers. 

Conclusion. 



A relatively simple equivalent circuit can be used successfully to model and 



Third International Symposium on Space Terahertz Technology 



Page 249 



400 



§ 

Q. 

3 

UJ 



O 



m 

Q 



UJ 

o 



a: 



300 



200 



£ 100 - 



-" 1 r " 






L-^- 



100 




1111 




200 300 

LOCAL OSCILLATOR FREQUENCY (GHz) 



Figure 9. DSB noise versus frequency for IRAM SIS re- 
ceivers. Dotted lines : receivers, now on the telescopes; con- 
tinuous lines : receivers in the laboratory. 



0.4 



0.3 



0.2 



0.1 



i — i r 



t — i — i — r 



t — i — i r 



-0.1 



II. I 



in 



CO (J-2-1) 



A1F (J=7-8) 



r\^^v^^^ 



>A 



>V^M^W^J^ 



I ! I I 



J L 



I 



I I I I I I I » I 



I 



230500 



230600 230700 230800 

Rest Frequency (MHz) 



230900 



Figure 10. Detection of aluminum fluoride with one of the 
two 1.3mm SIS receivers at the IRAM Pico Veleta radiote- 
lescope (Cernicharo and Guelin 1987). 



Page 250 Third International Symposium on Space Terahertz Technology 



improve the suspended stripline mixer mount. The performance of such 
a mixer using an SIS junction can be significantly improved by employing 
inductive tuning of the junction capacitance. It is also noteworthy that 
focusing the design effort on mismatch losses, and leaving aside intrinsic 
conversion loss, we can show good correlation between computed mismatch 
losses and measured receiver temperature, and that such modelling can serve 
as an effective guide to improving significantly the receiver performance. 

References 

Blundell, R. Carter, M., and Gundlach, K.H. 1988 Int. J. of Infrared and 
Millimeter Waves, 8, 361 

Carter, M.C., Navarro, S., Karpov, A., Billon-Pierron, D., Lehnert, T., 
Gundlach, K.H. 1991 Proceedings 16 th Int. Conf. Infrared and Millimeter 
Waves. 

Cernicharo, J., and Guelin, M., 1987 Astronomy and Astrophysics 183, L10. 

Karpov, A., Blondel, J., Mattiocco, F., and Lazareff, B., 1992 Journees 
d'Etudes Micro-Ondes et Espace, Toulouse CNES/CNET. 

Lehnert, T., Grassl, C, Gundlach, K.H., and Blondel, J. 1991 Supercond. 
Sci. Technol. 4, 419 

Yuda, M., Kuroda, K., and Nakano, J. 1987 Japan. J. Appl. Phys. 26, 166 

Lehnert, T., Billon-Pierron, D., Grassl, C, and Gundlach, K.H. 1992 Iram 
Working Report 210 



Third International Symposium on Space Terahertz Technology Page 251 

Comparison of Measured and Predicted Performance of a SIS Waveguide Mixer at 
345GHz fl9 3«§5L£^9 

CE.Honingh*, G.de Lange', M.M.T.M.Dierichs*, H.H.A. Schaeffer*, J.Wezelman*, 
J.v.d.Kuur', Th.de Graauw # , and T.M.Klapwijk* ^ t ^ 

# Space Research Organization of the Netherlands (S.R.O.N.), Landleven 12, 9747 
AD Groningen, The Netherlands 

* Dept. Applied Physics and Materials Science Centre, University of Groningen, 
Nijenborg 4, 9747 AG Groningen, 

The Netherlands 

Abstract 

The measured gain and noise of a SIS waveguide mixer at 345 GHz have been 
compared with theoretical values, calculated from the quantum mixer theory using a 
three port model. As mixing element we use a series array of two Nb-Al 2 3 -Nb SIS 
junctions. The area of each junction is 0.8 \xvtf- and the normal state resistance is 52 
n. The embedding impedance of the mixer has been determined from the pumped 
DC-IV curves of the junction and is compared to results from scale model measure- 
ments (105 x). Good agreement was obtained. The measured mixer gain however is a 
factor of 0.45 ± 0.5 lower than the theoretical predicted gain. The measured mixer 
noise temperature is a factor of 4 - 5 higher than the calculated one. These discrepan- _ 



Page 252 Third International Symposium on Space Terahertz Technology 

ties are independent on pump power and are valid for a broad range of tuning 
conditions. C <\ \ 



Introduction and measurement set up. 

This study is done as part of an ESA research contract to investigate the feasibility of 
SIS-mixers as space qualified THz-mixers. Predictions of the mixer performance are 
mainly based on the quantum mixer theory, by Tucker, reviewed in l . At lower 
frequencies the validity of the theory has been investigated thoroughly 2 , and quan- 
tum limited noise behaviour has been measured in very few cases 3 . 
Our main purpose for this study is to identify sources of noise in the receiver and to 
asses the quality of the tuning of the mixer. Receiver noise temperatures measured 
with the Y-factor method are shown in Fig.l. An overview of the route that we follow 
to obtain all information using only noise measurements is outlined in Fig.2. 
Measurements were done with two different mixerblocks. One mixer block (TT) a 
backshort and an E-plane tuner 4 , and another similar mixerblock (ST), without the 
E-plane tuner. We use non-contacting backshorts with two quarter lambda high/low 
impedance sections covered with an insulating Si0 2 layer of 200 run. 
As mixing element we use an array of two Nb-Al 2 3 -Nb junctions in series, each with 
an area of .8 Mm 2 and a normal state resistance of 52 n. The toRC product of the 
array is approximately 5 at 350 GHz. All measurements have been done with a 
magnetic field of two fluxquanta in the junctions and over an IF bandwidth (B) of 80 



Third International Symposium on Space Terahertz Technology Page 253 

MHz around 1.4 GHz. 



Measured mixer data 

The mixer gain (GMM) is calculated from the subtraction of the IF-output power in 
response to a 300 K and a 77 K input load. GMM={Pout(300)-Pout(77)}/{Gif.Gf.- 
APin}, where Gif is the gain of the IF-chain, Gf is the gain of the IR-filter at 77 K, 
and APin is the difference in input power between a 300K and a 77K load on the 77 
K radiation screen in the dewar. 

To achieve the highest accuracy Gif is determined in situ by using the unpumped 
mixer junctions as a calibrated noise source as a function of bias voltage 5 . The total 
IF output power as a function of bias voltage is given by 

Pif^iV) = G„ [leBG, J de (lOcot:h(-gl) (*£& ( V) *G X ) ' 2 +kB ( T iaol \T ir ( V) \ 2 * T jr ) } 



and is fitted to the measured power. V is the biasvoltage, and I DC (V) is the unpum- 
ped IV-curve. e is the electron charge, k is Boltzmanns' constant and T is the physical 
temperature of the junction, taken to be 4.5 K. G, is the input impedance of the IF- 
chain. 

Gif is obtained with an accuracy of 5% from the slope of measured IF-power as a 
function of bias voltage above aprroximately two times the gap voltage. The noise 
temperature of the IF-stage is T IF + | r(V) 1 2 T isol , where T IF is the noise temperature 
of the HEMT-amplifier (Berkshire Technologies) and T^, assembles the noise contri- 



Page 254 Third International Symposium on Space Terahertz Technology 

butions from the bath temperature, and possible contributions of imperfect isolation 
between amplifier and mixer. r(V) is the reflection due to the impedance mismatch 
between the IF-chain and the junction. Since Tif= 3 ±0.5 K and Tisol=5.5±0.5 K are 
obtained from the fitting, the second term, which is essentially depedend of the 
dynamical conductance of the junction array, can have a significant contribution. 
APin is calculated from Plancks' law. The gain of the dewar window (Gw), the 
beamsplitter (Gbs) and the IR-filter (Gf) have been measured separately with a 
Michelson interferometer. Gbs = 0.89 ±1%, Gw=95±2% and Gf=95±l% for the fre- 
quency of interest. In the calculation of the input power on the mixer it is assumed 
that the window is at 300 K. 
GMM is given in Fig. 3 as a function of bias voltage for both mixers. 

Determination of the embedding impedance 

Knowledge of the embedding impedance is crucial to the theoretical calculation of 
the mixer performance of an SIS junction. For design purposes we used a 105 x scale 
model of the mixer mount. The impedance measured on the final structure as a 
function of backshort position and at optimum E-plane tuner position, is given by the 
larger circle in Fig. 4.. The estimated geometrical capacitance of the junction array (22 
fF) has been added in parallel to the impedance measured in the scale model. 
The embedding impedance in the real mixer has been determined from the pumped 
IV-curves. We regard the series array of two junctions as one equivalent junction. The 
measured and calculated pumped curves are compared using the voltage match 
method 6 , where both the embedding impedance and the pump power are adapted to 



Third International Symposium on Space Terahertz Technology Page 255 

give a best fit. A typical example of a measured and a fitted curve is given in Fig. 5. 
The correspondence between the two curves was always very good execpt for a small 
region at the quasi particle step above the gap voltage. 

The embedding impedance has been determined for various backshort positions at 
one (optimum) E-plane tuner position. The expected circle in the Smith chart is fitted 
through the points in Fig. 4. The given points are lying in a very small part of one 
half lambda cycle of the backhort. The pattern is repeated for the next half lamba 
cycle, without a measureable increase in loss. To make that more clear the data of 
two cycles are given as a function of backshort position in Fig 4. The data as predic- 
ted by the scale model and a direct measurement of the coupled power (the pump 
step height) are also given as a function of backshort position. The DC-current at a 
biaspoint on the quasiparticle step has been normalized to one. 



Comparison between measured and calculated mixer performance 

The embedding impedances determined from the pumped IV-curves (and checked by 
the scale model measurement) have been used to calculate the gain and the noise 
behaviour of the mixer. We used the three port model in the low IF approximation, 
justified by the &>RC-product of the junctions and the IF frequency of 1.4 GHz. The 
terminations on the LO-port and at both side band ports were each determined 
separately. They differed considerably as can be seen in Fig.6, giving the pumped IV- 
curves at a single tuner setting for three different frequencies. 



Page 256 Third International Symposium on Space Terahertz Technology 

1GAIN 

The calculated mixer gain (GMC) 

GMC(V) = 4G L (G ush |Z 01 (V)| 2 + G Js „|Z . 1 (V)| 2 ) 

is given as a function of bias voltage in Fig.3. Z 0l and Zq , are the relevant elements 
of the 3x3 conversion matrix 1 and Gusb and Glsb are the real parts of the terminating 
impedances at both side band frequencies, as determined from the pumped IV-curve. 
This gain is directly compared to the gain (GMM) determined from the measure- 
ments in Fig.3. 

The discrepancy between GMM and GMC is independent of LO-power and also 
within a 15% error independent of the tuning conditions. It must be noted this has 
only been checked for the points given in the Smith Chart of Fig. 4. Around those 
points the fitting of the embedding impedances is the most accurate. For the most 
inductive tuning points the discrepancy in the gain is larger. At those points the bias 
supply seems to skip over the regions with negative differential resistance, deteriating 
the DC-curve and IF-output. For points more to the edge of the Smith Chart the 
amount of pump power necessary to get a well developed pump step is larger and the 
gap of the superconductor decreases, making accurate fits more elaborate. 

2 NOISE 

To obtain a measure for the noise contribution of the mixer we compared the 



Third International Symposium on Space Terahertz Technology Page 257 

measured and the calculated total noise output of the receiver in an IF-bandwidth of 
80 MHz. 

The noise contribution of the mixer is calculated from the DC-IV curve and the 
embedding impedances using the current correlation matrix 1 . As in the unpumped 
case the junctions array is regarded as one equivalent junction obtained by dividing 
both the measured current as the measured voltage by the number of junctions. The 
mixer gain used in the calculation is the gain determined from the measurements . 
This means that we attribute the discrepancy between GMC and GMM fully to the 
loss/coupling efficieny of the lens/horn/waveguide at 4K in front of the mixe. The 
calculated and the measured IF-output power as a function of bias voltage are given 
in Fig. 7a. However to get the correspondence at the first pump step as shown in Fig. 
7a , an extra input noise power kBT ex , with T ex = 80 ± 20 K, had to be added in the 
calculation at both side bands in addition to the shotnoise and temperature noise 
contribution . This value for T ex is again independent of pump power and tuning 
conditions within the same restrictions to the tuning range as mentioned in the 
calculation of the gain. The calculated and measured noise contributions of the 
various parts of the receiver are given in terms of noise temperature in Fig. 7b. 
The results in Fig. 7 are for the ST-mixerblock but a similar performance is found in 
the TT-block. Though still within the error margin the deviation in the gain has a 
tendency to be less in the ST-mixer compared to the TT-mixer, probably as a result of 
the improved fabrication and the use of an integrated horn. 

We verified that the extra noise contribution was not a real extra input signal due to 
LO-signal at the side band frequencies by filtering the LO with a high Q Fabry-Perot 
filter. 



Page 258 Third International Symposium on Space Terahertz Technology 

Discussion and Conclusions 

We compared the performance of two types of waveguide SIS-mixers with the three 
port quantum mixer theory. We have obtained good agreement between the scale 
model measurements and impedances determined from pumped IV-curves. The 
quality of the fittings is very high in the sensitive tuning region of the mixer. 
However we observed a reproducible difference between the measured and the 
calculated gain of both mixers. The difference can be explained partly by losses in the 
lens and horn. 

The performance of the backshort seems to be quite lossless regarding the good 
agreement between the scale model measurements and the impedances fitted to the 
pumped IV-curves. 

The noise values are more than a factor of four higher than expected from theory. 
This seems to be a general feature of mixers using a series array of junctions. Up to 
now we did not yet have single junction mixerrs available. 



We acknowledge the financial support of the European Space Agency for this work 
under contract 7898/88/NL/PB(SC) and Herman v.d. Stadt for careful reading of this 
summary and Anders Skalare in general. 



Third International Symposium on Space Terahertz Technology Page 259 

References 

1 J.R.Tucker, and M J.Feldman, Rev.Mod.Phys. £7, 1055 (1985) 

2.W.R.McGrath, P.L.Richards, D.W.Face, D.E.Prober, and F.L.LIoyd, J.Appl.Phys. £3_, 
2479 (1988) 

3.C.A.Mears, Qing Hu, P.L.Richards, A.H.Worsham, D.E.Prober, and A.V.Raisanen, 
IEEE Trans.Magn. MAG-27, 3363 (1991) 

4. B.N.Ellison, P.L.Schaffer, W.Schaal, D.Vail, and R.E.Miller, IntJ. of IR and MM- 
wavesiQ, 937 (1989) 

5. J.R. Tucker , IEEE J. Quantum Electron. QE-15, 1243 (1979) 

6. A.Skalare, Int. J. of IR and MM waves K), 1339 (1989) 



G 



./ 



Page 260 Third International Symposium on Space Terahertz Technology 

Captions 

Fig. 1 Receiver noise temperature tor the two types of waveguide mixers measured 

with the Y-factor method, corrected for the beamsplitter loss. 

Fig. 2 Overview of the different input an output parameters in the process of compa- 
ring the measured and calculated performance of the mixers. 

The measurements yield Tree as result of a Y-factor (H/C) measurement. The gain 
and noise of the IF(Gif,Tif) and of the mixer(Gm,Tm) are obtained from the absolute 
IF-output power at different input loads, knowing the loss and the physical temper- 
ture of the input window(Gw.Tw). 

The embedding impedance of the junctions is determined either with use of a scale 
model or by fitting the pumped IV-curves to the theory. When a scale model is used 
the geometrical (and parasitic) capacitance of the junction has to be estimated se- 
parately. 

When the embedding impedance is known, the mixer performance is calculated as a 
function of bias supply at different LO-power levels. 

Fig. 3 Measured ( + ) and calculated ( • ) coupled gain for both waveguide mixers. The 
TT-mixer has a 500-50 n transformer at the IF-port to enhance the gain. 

Fig. 4 Embedding impedance as a function of backshort position, as calculated from 
the scale model (-+-) and as determined from the pumped IV-curves (M,A). As a 
direct measure of the coupled power the pumped step height at the optimum 
biaspoint (-o-) is also given as a function of backshort position 



Third International Symposium on Space Terahertz Technology Page 261 

Fig. 5 Measured (-■-) pumped IV-curve at 351 GHz and calculated(-) curve using the 

given fitting parameters for the embedding circuit. The admittances are normalized 

to the 104 n. 

Fig. 6 Detailed view of DC-IV curve of the series array of junctions, pumped at three 

different frequencies. The tuning conditions and the pump power are identical at all 

frequencies. 

Fig. 7A Total measured ( + ,o) and calculated (-) IF-output power in a bandwidth(B) 

of 80 MHz at two different input signals, as a function of bias voltage. For the 

calculated IF-power an extra noise power of 80kB has been added to the input of the 

mixer. The contributions of the shot noise and temperature noise of the junctions 

(dPjunctie) and of the IF-stage (dPif) are given also. 

Fig. 7B The total measured ( + ) and calculated ( • ) reciever noise temperature as a 
function of bias voltage. For reference the contribution to the calculated reciever 
noise temperature of the IF-stage (dTif), the junctions (dTm) and of the input losses 
(dTw) are also given. 



Page 262 



Third International Symposium on Space Terahertz Technology 



300 



ffl 

09 

Q 200 



-+- 1 tuner -A_ 2 tuner 



o 
o 

u 

h 



100 



A— A y 



\ 



+-+- 



^<^~* 






-K 



./ 



/ 



340 348 356 364 372 380 
frequency (GHz) 



Fig.1 



GUIDE LINE 



H/C 



Receiver 
(dewar) 



Phot 



Pcold 



Tree 

fif(V) 
LQpm.Mxer, jp — f Gif(V) 




V 




eg 



Scale- 
model 



DC- 
IV 



Tm(V) 

Qm(W 



Analysis 
Pumped 
rV-curves 



Ylo, 
Yusb,Ylsb 



Plo 



Three-port 
mixer model, 
Tucker theory 



-* Ti 



m 



Fig. 2 



"Vto/V««0 



H 



■t 
& 
o 



C/> 



ST-mlxer 
Mitx*!- ■■In 



TT-mixor 
Kflm*r aala 



1 .2 
















CKlo. 


so o 








Meat. 


I** 


1 -O 










o.« 










o.e 










O.-* 






y\ 




0.2 

a.o ' 


■^ 


Mi 


A 





i .a 



> .o 



o.a 



©.« 



<>.■♦ 



o.a 



o.o 




SOO O 
IP 



Q 
t/i 

£' 
3 

o 
a 

» 
o 

H 

a- 
55 



3 

n 
a- 
a 
o 

I 



«n* 



Fig. 3 






Page 264 



Third International Symposium on Space Terahertz Technology 




ZO-110 Q 



Coupled power as a function 
of backshort position 



I COLD uODl 



■» fi {Yemb from fit 



Wc 
In 




1500 Bacfakpos. (>im> 2300 



Fig. 4 



G.emb=1.60 












2 










B.emb=-0.33 




■ meas. 
— calc. 








Io>=1.41 


lb 
Io 


• 


i 
\ 

• 













^■i — _ 








( 


> 


VtyVgap 


2 





Fig. 5 



Third International Symposium on Space Terahertz Technology 



Page 265 




2.0 3.0 4.0 
Vb (mV) 



3.0 6.0 



Fig. 6 



ST-mixer 

2 nx. trriy (Nb-SIS) 




ST-mixer 

NoiM contribution* (II) 



o 
o 

u 

H 



300 



250 



200- 



150 



100 



50 




i i 
i < 
i i 

i i 
\ i 
ti 
u 



1 

Vb/Vjtp 



B 



<JTm 

— — <JTif 

— ♦— Ttot 

* Tree 

dTw 



Fig 7. 



Page 266 Third International Symposium on Space Terahertz Technology 

jho&B N93-2? 750 



A Low-Noise 492 GHz SIS Waveguide Receiver 



C. K. Walker'.t, J. w. Kooi' . M. Chan', H.G. LeDuc* , P.L. Schaffer' , J.E. 

Carlstrom' , and T.G. Phillips 1 

1 California Institute of Technology, Pasadena CA 

2 Jet Propulsion Laboratory, Pasadena C A 
f Presently at the University of Arizona, Tucson AZ 



Abstract 



In this paper we discuss the design and performance of an SIS waveguide 
receiver which provides low noise performance from 375 to 510 GHz. At its design 
frequency of 492 GHz the receiver has a double sideband noise temperature of -172 
K. By using embedded magnetic field concentrators, we are able to effectively 
suppress Josephson pair tunneling. Techniques for improving receiver performance 
are discussed. 

Introduction 

Over the last decade SIS receivers have been replacing Schottky diode based 
systems at millimeter and submillimeter wavelengths. SIS junctions have a lower 
shot noise and a more pronounced D.C. nonlinearity than Schottky diodes, with the 
result being that mixers constructed with them are more sensitive and require less 
local oscillator power than their Schottky diode counterparts. 

SIS mixers can be constructed using waveguide or an open structure geometry. 
To date, at all frequencies where they have been built, SIS waveguide mixers provide 
superior performance. The main advantage waveguide mixers have over open 
structure designs is that adjustable backshorts can be readily incorporated into the 
mixer block. Backshorts are usually needed to match the complex impedance, 
although recently structures employing lithographically produced matching networks 
for waveguide mounted junctions has proved highly successful (Kerr et al. 1987). 
Open structure mixers typically utilize a combination of lenses and planar antenna 
structures to couple the incoming radiation to the junction. With this combination of 
components it is not easy to incorporate an adjustable backshort. Fixed tuned reactive 
stubs can be fabricated along with the SIS device to tune out the junction's 
capacitance. A significant advantage of waveguide designs is that well characterized, 
efficient feedhoms can be used to couple waveguide modes to free space modes. 



Third International Symposium on Space Terahertz Technology 



Page 267 



Until recently SIS waveguide receivers have been constructed with center 
frequencies only as high as 345 GHz. In this paper we discuss the construction and 
performance of an SIS waveguide receiver with a center frequency of 492 GHz. It is 
now permanently installed on the Caltech Submillimeter Observatory and has been 
used for astronomical observations since September 1991. 

Receiver Construction 



Optics 

A block diagram of the 500 GHz receiver is shown in Figure 1 . The receiver's 
optics are designed to provide a -10 db taper on the edge of the telescope's secondary 
mirror. The beam from the secondary is reflected from an offset parabola, a flat, and 
a final offset parabola before entering the cryostat. At the entrance to the cryostat a 
thin (- 0.2 mil) mylar beamsplitter is mounted at 45* to the signal and local oscillator 
beams. With this thickness of mylar only 0.5% of the signal and local oscillator beam 
is reflected. Therefore only a tiny fraction of the signal is lost, while almost all the 
local oscillator power is terminated in an absorbing load. 

, par coked 
MKxn.D«moo«» 




IF. LNA 

m '.0- 20O.HI 



PGOUTEUPAMR 
HITS)) 



IGOft TEW* AMP. CSERTQO 



10TALPCWER 
OUTRjr 



L 



| Cl.'SCXi 

I «.i.0 5aw 4 

I I^.^Tt.1 R^T— I 



SO LAWOETECTCn 



4f.- i.o a* 



TO OSSPECTROUEIEfl 



Figure 1 

The cryostat vacuum window is a 1.0 mil mylar sheet. The windows on the 12 
K and 77K shields are made from disks of fluorogold laminated to a quarter 
wavelength thickness of black polyethylene. The polyethylene reduces reflection 
losses. The overall thickness of the fluorogold and black polyethylene disks is 17 
mils. The disks serve as near infrared blocking filters for the system. 



The last optical component before the mixer block is a small, low density 



Page 268 



Third International Symposium on Space Terahertz Technology 



polyethylene lens mounted in front of the feedhom. The lens is designed using the 
equations of Silver (1966) and is located so that it provides frequency independent 
illumination of the secondary (Goldsmith 1982). 



E PONE T & WAVEGUIDE 
TRANSFORMER SECTION 




IF OUTPUT 

IF MATCHING CIRCUIT 



CORRUGATED 
FEEOHORN SECTION 



..aACKSHOflr TUNER 



Diagram of Mixer Block 
Figure 2 



Mixer Block 



A schematic representation of the mixer block is shown in Figure 2. An 
exploded view of the completed block is shown in Figure 3. The basic design of the 
waveguide portions of the block follows that of Ellison et al. (1987). The block is 
divided into five sections along the longitudinal axis of the block. The first section 
consists of a corrugated feedhom (Thomas 1978) which terminates in circular 
waveguide. The beamwidth of the horn, at its -10 db points is 13.4*. In the second 
section, the signal passes through a three step, circular to full-height rectangular, 
quarterwave transformer. This section also contains the waveguide for the E-plane 
tuner. The center of the E-plane tuner is located - Xg/2 in front of the SIS junction. 
The third section contains the SIS device, the IF impedance matching circuit, 
magnetic field concentrators, and the waveguide for the backshort. A face-on view of 
this section is shown in Figure 4. The location of the main circuit elements are 
indicated. The SIS junction substrate resides in a long 4.5 by 4 mil channel milled 
across the center of the waveguide. When the junction substrate is placed in the 
channel, it is oriented so that the junction itself is in the center of the waveguide 
facing the oncoming radiation. A RF choke is fabricated on the substrate at the same 
time the junction is made. The choke, shown in Figure 5, is made from a series of 
high and low impedance sections of microstrip line. Each section is a Ay 4 in length at 
the RF center frequency. The ground side of the junction is held in place with silver 
paint. The "hot" side of the junction is connected to the IF matching circuit via a 
short 1 .0 mil gold wire. The wire is silver painted to the last section of the RF choke 
located on the junction substrate. To keep the silver paint from inadvertently shorting 
the hot side of the junction substrate to the block, the block is designed so that the last 
8 mils of the junction substrate is suspended in free space. 



Third International Symposium on Space Terahertz Technology 



Page 269 



Matching and Bias. 
Circuit 



Corrugated 



Lens Holder 





E-flane 
Backshort 
Drive 



Waveguide Transformer 
and E-Ptane Tuner 



Exploded View of the 490 Bloc k 
Figure 3 



IF Matchi 
Circuit 



Signal 
Waveguide 

Field 
Concentrator 




F Connector 



as Lines 



lunction 
Channel 



Field 
Concentrator 



Junction Block 
Figure 4 

A detailed discussion of the integrated IF matching circuit has been given by 
Kooi et al. (1992). It utilizes a 5-pole Chebyshev low pass filter and transformer to 
match the IF impedance of the SIS junction to the input impedance of the IF amplifier 
(- 50 £1) over a 1 to 2 GHz frequency range. The IF impedance of the junction is 
roughly 2.5 times the junction's normal state resistance. The matching circuit is 
designed for an SIS IF impedance of - 160 CI. 

The magnetic field concentrators and the core of the external electromagnet to 
which they are connected are made out of Cryoperm, a material which retains a high 
magnetic permeability even at 4 K. To reduce heat loading on the 4 K cold plate, the 
magnet's coil is made from niobium wire. The backshorts are non-contacting and 
were designed using the techniques discussed by Brewer and Rasianen (1982). Non- 



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Third International Symposium on Space Terahertz Technology Page 271 



Junction Fabrication 

The Nb/ALO x /Nb tunnel junctions used in this receiver are planar, submicron 

area devices defined by electron beam lithography. The junctions are fabricated using 
a variant of the self-aligned-liftoff trilayer process (Shoji et al. 1983) with 
modifications for electron beam lithographic patterning of junction area. The 
Nb/ALO x /Nb trilayer is deposited in a high vacuum sputter deposition system (base 

pressure 1.3 X 10 Torr). The Nb/Al layers are dc/rf -magnetron sputtered from 75 
mm diameter targets. The A10 x tunnel barrier on the trilayer is formed by an in situ 

oxidation in an oxygen/argon mixture. A gold layer (» 30 nm) is deposited on the 
trilayer to act as a contact layer. The junctions are patterned by forming submicron 
holes in PMMA by electron beam lithography (JEOL JBX5) followed by the 
deposition and lift-off of chromium metal. The chromium pattern is transferred to an 
underlying polyamide film using oxygen reactive ion etching (RIE). Junctions are 
formed by RIE using a gas mixture containing CCI2F2 (chosen for its highly 

anisotropic etch characteristics) and electrically isolated using thermally evaporated 
silicon monoxide. Contact wiring is deposited and patterned using RIE to complete 

the device. Tunnel junctions with areas of 0.25 um^ and 0. 13 urn 2 were fabricated on 
on the same wafer 

Receiver Performance 

Figure 6a is a plot of the I-V (current versus voltage) curve of the SIS junction 
used in the receiver. The solid line is the I-V curve with no local oscillator power 
applied. The curve has a sharp knee at the voltage (- 2.78 mV) corresponding to the 
band gap energy of the niobium junction. At the knee, the leakage current through the 
junction is about 5 uA. The normal state resistance of the junction is about 90 ft. 
The dotted line is the I-V curve with local oscillator applied. With local oscillator 
power applied a single wide photon step is observed below the knee of the curve. A 
single wide photon step is observed below the knee of the curve. The width of this 
step corresponds to the voltage {(to/*) of a 492 GHz photon (- 2 mV). Figure 6b is a 
plot of IF power versus SIS bias voltage. The top curve is the IF power obtained with 
the receiver looking into a room temperature load (Tpj - 280 K). The lower curve is 

the IF power obtained with the receiver looking into a cold load (Tq - 80 K). As 

expected, the IF conversion peak occurs at a bias voltage corresponding to the middle 
of the first photon step below the gap voltage. The ratio of the IF power obtained with 
the receiver looking into the room temperature load to the IF power obtained with it 
looking into the cold load is a measure of the receiver's sensitivity and is often 
referred to as the Y-factor. The receiver noise temperature is derived from the Y- 
factor using the following relation. 

To = T " • rcy 

R Y- 1 



Page 272 



Third International Symposium on Space Terahertz Technology 



At 492 GHz the highest Y-factor obtained with this receiver was 1 .84, which 
corresponds to a double sideband receiver noise temperature of - 172 K. This value is 
a true receiver noise temperature, no corrections have been made for signal losses in 
the beamsplitter or input optics, losses resulting from impedance mismatches, or from 
IF amplifier noise. The bias voltage and current used during this measurement were 
2.3 mV and 12 uA. At this bias voltage the receiver noise temperature increased 
when the LO power was reduced tp a level where the junction current became less 
than - 9 uA. Similarly, if the LO power was increased such that the junction current 
reached a value greater than 16 urn, the receiver sensitivity decreased. 




V(mV) 



SIS RECEIVER I/V CURVE 
FIGURE 6a 




T T 

V(mV) 10 

SIS RECEIVER IF POWER vs. BIAS VOLTAGE 
FIGURE 6b 



Third International Symposium on Space Terahertz Technology 



Page 273 



During these measurements the magnetic field strength was adjusted so as to 
minimize the manifestation on the I-V characteristics of the Josephson pair tunneling 
current. As the magnetic field strength was increased, the Josephson super-current 
went through several minima. A minimum in the super-current occurs when one or 
more magnetic flux quanta are present across the junction. Since more than one 
minimum was observed, we conclude the magnetic circuit used in this design is 
capable of placing several flux quanta across the junction. Without the magnetic field 
the smooth IF power curves of Figure 6b become jagged. In Figure 7 we present IF 
power curves made at 420 GHz. Figure 7a shows the smooth IF power curves that 
can be obtained with an optimized magnetic field. In Figure 7b the same IF power 
curves are plotted, but with the applied magnetic field less than optimum. In 7a and 
7b we also plot the corresponding junction I-V curve. The small dips in the IF power 
curve of Figure 7a become more prominent in Figure 7b, and occur at voltages where 
Josephson steps are seen in the pumped I-V curve. The association between the dips 
and the Josephson steps indicates that these structures are the result of Josephson 
effect tunneling. These results show that, even at high frequencies, a magnetic field 
can be used to effectively suppress Josephson pair currents in small area junctions. 




i — r 

V(mV) 1.0 

JUNCTION I/V & IF POWER CURVES WITH MAGNETIC FIELD 
FIGURE 7a 



Page 274 



Third International Symposium on Space Terahertz Technology 




V(mV) 1.0 2.0 3.0 4.0 5.0 

JUNCTION I/V & IF POWER CURVES WITH REDUCED MAGNETIC FIELD 

FIGURE 7b 



In Figure 8 we show spectrum analyzer measurements of the receiver bandpass 
from 0.9 GHz to 2.1 GHz. The lower curve in the figure is the bandpass with the 
receiver looking into a room temperature (280 K) load. The bandpass is fairly flat, 
with a total power variation of only - 3.5 db from 1 to 2 GHz. The upper curve is a 
plot of the receiver's Y-factor across the IF band. This curve is essentially flat, 
indicating the receiver's sensitivity is constant across the band. 

In Figure 9 we present a double sideband, 500 MHz wide spectrum taken with 
the receiver on the Caltech Submillimeter Observatory toward the young stellar 
source Orion IRC2. The center frequency of the spectrum is 492.16 GHz. An 
acousto-optical spectrometer was used to produce the spectrum. An atomic line and 
several molecular lines were observed. They are identified in Figure 9. The ordinate 
is in units of antenna temperature (K) and the abscissa is in units of frequency (GHz). 
The total on source integration time was - 1.3 minutes. At this frequency, the system 
noise is dominated by the atmosphere. During the time this spectrum was taken the 
single sideband noise temperature on the sky, which includes the noise contributions 
of the atmosphere, telescope, and receiver, was - 5000 K. 



Third International Symposium on Space Terahertz Technology 



Page 275 



ATTEN 2MB 
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♦ 94.7 



494.8 



Upper SidcSud Frequency (GHz) 
494.9 495 495. 1 



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492.4 



Page 276 Third International Symposium on Space Terahertz Technology 



The receiver noise temperature, *R, is determined by a number of factors. 
These include the mixer noise temperature, ( *M ), the conversion loss in the mixer 
(Closs ), the noise temperature of the first IF amplifier (TlF), and the coupling 
efficiency between the IF port of the junction and the input port of the first IF 

amplifier ("HlF). 

Receiver Performance Comparison 



Parameter 


230Nb 


345Pb 


492Nb 


Rn(fl) 


82 


54 


85 


TR(K) 


48 


159 


176 


Tmix (K) 


34 


129 


123 


C. Loss (db) 


3.1 


8.3 


8.9 


Tmix Cor 


26 


91 


95 


TIF(K) 


7.0 


4.2 


6.8 


IFConnib. 


14 


30 


53 



Table 1 

In Table 1 we list the values of these parameters for the receiver described in 
this paper and for the two other SIS waveguide receivers presently in use at the 

Caltech Submillimeter Observatory. The values of 'M, TV, Closs, and "-IF were 
calculated using the shot noise technique described by Tucker and Feldman (1985). 
The other two receivers in the table were designed with center frequencies of 230 and 
345 GHz. All three receivers use the same basic double stub tuner design. Like the 
492 GHz receiver, the 230 GHz system employs a niobium junction and the IF 
matching circuit described by Kooi el al. (1992). The 345 GHz receiver uses a lead 
based SIS junction fabricated by Ron Miller at AT&T Bell Laboratories. This 

receiver uses an older, less efficient IF matching circuit. In the table °N refers to the 

-rcor 
normal state resistance of the SIS junction used in each receiver. l M refers to the 

mixer temperature corrected for the different beam splitter thicknesses used in each 

receiver. The value of Tr for the 230 GHz system is a factor of 3 to 4 less than the 

Tr achieved with the 492 GHz system. The decrease in system performance at 492 

GHz is due to a factor of - 3 increase in Closs and Tm. The performance of the 345 

and 492 GHz receivers are comparable. This similarity in performance is most likely 

due to the niobium junction in the 492 GHz receiver having a better high frequency 

response than the lead junction used in the 345 GHz receiver. 



Third International Symposium on Space Terahertz Technology 



Page 277 



Broadband Heterodyne Recaverg 



1000 
a? 

CO 

in 

>^ 

$ 

100 



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[ A SIS Waveguide 

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1000 



Frequency [GHz] 
FIGURE 10 



In Figure 10 we compare the performance of the 492 GHz receiver to other 
receivers operating at millimeter and submillimeter wavelengths. At each frequency 
where they have been built, SIS waveguide receivers have proven to be the most 
sensitive. These receivers typically operate with noise temperatures 10 to 14 times the 
quantum noise limit (fv/IQ. The receiver reported in this paper continues this trend. 
We have also tested the 492 GHz receiver at 376 GHz and 420 GHz. We obtained Tr 
values of 238 and 212 K respectively. (These measurements were taken at sea level 
where the physical temperature of the mixer was - 0.62 K higher than at the altitude 
where the 172 K, 492 GHz noise temperature measurements were made.) These noise 
temperatures include all the losses in the system. If we simply compensate for the 
difference in the thickness of beamsplitter used in the measurements, then we infer a 
noise temperature of 200 K at 376 GHz and 168 K at 420 GHz. These results suggest 
the receiver will have its optimum performance between 420 and 492 GHz. 

Improving Receiver Performance 

Lower values of Tr would be achieved if we could reduce 'IF, increase ^H\ r 
decrease either C-Loss and/or Tm. In a properly designed amplifier, the value of iff 
is a function of the quality of the HEMT devices used and the desired IF bandwidth. 

In the receiver described here the value of ^IF is almost unity, so there is not much to 
be gained in improving the design of the IF matching circuit. Significantly lower 



Page 278 Third International Symposium on Space Terahertz Technology 



noise temperatures could be achieved if the conversion loss of the mixer were 
reduced. Lower conversion losses can be obtained by improving the impedance 
match between the junction and waveguide embedding impedence. In the current 
design this match is achieved by adjusting the positions of the two backshorts. The 
ultimate quality of the impedance match depends on the performance of the 
backshorts. Since this receiver will be tuned several times a day, non-contacting 
backshorts were used to reduce wear. The backshorts are insulated from the block by 
a thin (- 16 urn) layer of mylar tape. The performance of the backshorts could be 
improved (perhaps at the sacrifice of durability) by using a thinner insulating material. 
Ellison et al. (1991) have recently demonstrated that a 1 to 2 urn layer of silicon 
dioxide evaporated onto metallic backshorts serves as an effective insulating layer and 
improves backshort performance. An improvement in the rf match could also be 
achieved by reducing the capacitance of the SIS junction. The most direct way of 
doing this is by reducing the size of the junction. If the normal state resistance of the 
junction is to remain the same, the thickness of the insulating layer must be decreased 
simultaneously. These last two statements are equivalent to saying we need a high 
current density junction with a low coRC. The limiting factor in obtaining small, high 
current density junctions is the junction fabrication process itself. However, over the 
past few years great strides have been made in improving processing techniques. 
There is every indication that this trend will continue. The effective capacitance that 
the backshorts need to tune out can also be reduced by fabricating broadband, 
inductive stubs on the SIS junction itself. This technique has worked well at 
frequencies below 300 GHz (Kerr et al. (1987)). However, as one goes up in 
frequency, the dimensional tolerances on these matching circuits becomes more 
severe, making fabrication difficult. Even so, recent work by Buttgenbach et al.. 
(1992), and Jacobs K. et al. (1992) suggests that if a broadband rf matching network 
is included, this technique can be used effectively even at submillimeter wavelengths. 
Indeed, we plan to test a mixer block like the one discussed here with a broadband, 
two section, RF transformer matched SIS junction in the near future. Another way to 
reduce the effective capacitance is to use a series array of junctions. Two difficulties 
with this technique are that the array elements must be similar (a fabrication 
challenge) and the required LO power increases as the square of the number of array 
elements. At submillimeter wavelengths, where it can be difficult to generate LO 
power, the extra power needed to drive a series array can become prohibitive. 

If there were no Josephson pair tunneling, one might expect that an SIS bias voltage 
corresponding to the center of the first photon step below the gap would provide the 
best receiver noise temperatures. However, we find that even with an optimized 
magnetic field, the best receiver noise temperatures are found at bias voltages close to 
the gap voltage. To some degree this result is due to the matching requirements of the 
waveguide and IF matching network. However, this dependence of the receiver noise 
temperature on bias voltage suggests that, while significantly suppressed by the 
magnetic field, Josephson pair tunneling does add noise to the system. To avoid this 
problem, SIS junctions with larger bandgap energies are needed. 



Third International Symposium on Space Terahertz Technology Page 279 



Summary 

We have constructed an SIS waveguide receiver which provides low noise 
performance from 375 GHz to 510 GHz. The receiver is a facility instrument at the 
Caltech Submillimeter Observatory, where it has been in use since September 1991. 

The SIS junction used in the receiver is a 0.25 \urr niobium trilayer device with a 

current density of - 10 A cm . At its design frequency of 492 GHz, the receiver has 
a double sideband noise temperature of - 172 K. By embedding magnetic field 
concentrators in the mixer block, we are able to put several quanta of magnetic flux 
across the SIS junction. By adjusting the strength of the magnetic field we are able to 
effectively suppress Josephson pair tunneling. The success of this receiver suggests 
that SIS waveguide receivers can provide low noise performance at even shorter 
wavelengths. 

This work was supported by NSF contract AST 9015755 to the CSO and a gift from 
AT&T to purchase the spectrum analyzer used to perform some of the measurements. 



References 

Brewer, M. K. and Raisanen, A. V. 1982, EEE. Trans. Microwave Theory and 

Techniques, 30, 708. 
Buttgenbach, T. H. 1992, private communication. 
Ellison, B. N., Little, L. T., Mann, C. M., and Matheson, D. N. 1991, Electronics 

Letters, 27, 139. 
Ellison, B. N. and Miller, R. E. 1987, Int. J. Infrared and Millimeter Waves, 6. 697. 
Goldsmith, P.F. 1982, in. Infrared and Millimeter Waves 
Kerr, A. R., Pan, S. K., and Feldman M. J. 1987, Int. J. Infrared and Millimeter 

Waves, 9, 203. 
Kooi, J. W., Chan, M., Phillips, T. G., Bumble, B., and LeDuc, H. G. 1992, preprint. 
Shoji, A., Kosaka, F., Shinoki, M., Aoyagi, M. and Hayakawa, H. 1983, IEEE Trans. 

Magnetism, 19, 827. 
Silver, S. 1966, 1EE Electromagnetic Waves Series, 19, 95. 
Thomas, B. M. 1978, IEEE Trans. Antennas and Propagation, 26, 367. 
Jacobs K., Kotthaus U„ andVowinkel B., International Journal of Infrared and 
Millimeter Waves, Vol. 13, No. 1, 1992 



Page 280 Third International Symposium on Space Terahertz Technology 

Sur&l'' N93-27751 

/bo^ Slot-line end-fire antennas 

^\ for THz frequencies 

by 

H. EkstrSm, S. Gearhart*. P. R. Acharya, H. Dave**, 
G. Rebeiz*. S. Jacobsson, E. Kollberg, G. Chin** 

Department of Applied Electron Physics 

Chalmers University of Technology 

S-412 96 Gfiteborg. Sweden 

•NASA/Center for Space Terahertz Technology 
Electrical Engineering and Computer Science Department 
University of Michigan, Ann Arbor. MI 48109-2122, USA 

••Planetary System Branch 

NASA/Goddard Space Flight Center 

Greenbelt. MD 20771. USA 



/ 



ABSTRACT 
Tapered slot-line endfire antennas, of BLTSA type, have been fabricated on 1.7 \im 
thin S102/S13N4 (£ r = 4.5) dielectric membranes. The antenna patterns, in the E-, H-, 
D- and D-cross planes, were measured at 270. 348, 370 and 802 GHz using bismuth 
micro bolometer detectors. The antennas have approximately 12 dB directivity, and 
the -10 dB beam widths are 50° and 55° in the E- and H-planes at 348 GHz. 
respectively. The measurements at millimeter/ submillimeter wavelengths compare 
well with scale measurements at 45 GHz as well as with theoretical predictions. The 
overall results are encouraging and show that slot-line antennas can be fabricated for 
use at THz frequencies. Furthermore, it is shown that the very thin SIO2/SI3N4 
membranes are strong enough to be used In practical applications. 



Third International Symposium on Space Terahertz Technology 



Page 281 



INTRODUCTION 
Tapered slot-line antennas are often considered for integration in planar millime- 
ter/submillimeter wave circuits: e.g. In quasi optical mixers. These antennas can be 
operated over a wide bandwidth and radiate wide or narrow beams. Various types of 
endfire slot-line antennas can be found in the literature; e.g. the linearly tapered slot- 
line antenna (LTSA) 11], the exponentially tapered slot-line antenna "Vivaldi" [2] and 
the constant width slot-line antenna (CWSA) [3]. A review of antennas suitable for 
integration in circuits for millimeter and terahertz frequencies has been written by G. 
Rebeiz[4l. 

In this work we have studied yet another member belonging to the family of endfire 
slot line antennas; namely the BLTSA (Broken Linearly Tapered Slotline Antenna) 
15], Fig. 1. The BLTSA has the advantage, among the slot-line antennas, to require the 
small substrate area. In addition, the BLTSA has been extensively studied by the 
Chalmers group. 

Antennas of endfire slot-line type require a certain optimum substrate thickness 
t = 0.03 X(^Tf -iy l [31 to avoid pattern degradation and power loss due to surface 
modes. In the millimeter wave range the thickness should be only a couple of mi- 
crometers, hence the antenna must be fabricated on a thin dielectric membrane, a fact 
which introduces delicate manufacturing problems. A further complication is intro- 
duced by the fact that the membranes must be left unsupported in the endfire direc- 
tion. Fig. 2. 



00 







3.75 



*±5_» 



1.5 



Fig. 1. Dimensions of the BLTSA. All dimensions are normalised to the 
vacuum wavelength. 



Page 282 



Third International Symposium on Space Terahertz Technology 




Fig. 2. The endfire slot line antenna deposited on BLTSA on 1.7 ^m thin 
Si02/Sl3N 4 (£r = 4.5) membrane. The Si frame supporting the membrane 
has a thickness of 385 pm. Note that the membrane is unsupported in the 
end-fire direction. 

FABRICATION 
The 1.7 nm thin dielectric membrane supporting the antenna consists of three layers; 
thermally grown Si02. LPCVD deposited Si3N5 and SiC>2. With compressive oxide 
and tensile nitride, the relative thickness of the layers could be selected to form a 
slightly tensile, and consequently flat and rigid membrane. The membrane layers 
were deposited on both sides of 385 (im thick silicon wafers. To form the membrane 
region for the antennas, the silicon was etched in EDP from the backside of the wafer, 
with the backside nitride and oxide layers patterned with the membrane layout and 
used as etch mask. The nitride and oxide layers on the front side acted as etch stops 
for EDP. A considerably manufacturing problem is due to the fact that the antenna 
requires that the membrane is not supported by silicon in the endfire direction. In 
order to simplify the photo lithography were the antennas fabricated on fully sup- 
ported membranes. After the antennas were fabricated was the silicon support in the 
endfire direction removed. The fragility of the membrane limits the area to roughly 
3 x9 mm 2 . Thus, the maximum available membrane size limits this particular an- 
tenna design to frequencies above 300 GHz. 



MEASUREMENTS 
Several scaled versions of BLTSA's were fabricated on 25.4 |im thick Kapton foil and 
measured at 45 GHz. The best antenna design was then scaled to 348 and 802 GHz 
(dimensions normalized to vacuum wavelength are given in Fig. 1). These antennas 
were fabricated on the 1.7 urn thin dielectric membranes. Fig. 2. 



Third International Symposium on Space Terahertz Technology 



Page 283 



Bismuth micro bolometers were used as detectors. A Gunn oscillator with a 
tripler/quadrupler was use as signal source at 270 and 348 GHz, whereas an optically 
pumped far infrared laser was used to generate the 802 GHz signal. The dynamic range 
in the antenna pattern measurements was approximately 20 dB. 

The E-, H-, D- and Dx- planes of the 348 GHz antenna were measured at four frequen- 
cies: 270, 348. 370 and 802 GHz, respectively, whereas the 802 GHz design was only 
measured at 802 GHz. Measurements of the 348 GHz design at 270 and 802 GHz, Figs. 
3a, b and c, show the wide bandwidth of this type of antenna. At the design frequency 
the -10 dB beam width varies between 43 and 55° in the measured planes. The D-plane 
crosspol level is as high as -6 dB, which is a typical value for endfire slot-line anten- 
nas. The directivity was calculated to approximately 12 dB. In the calculations the 
lobes outside the measured range and the back lobes were set to -14 dB and -20 dB re- 
spectively. 



-H 



D Dx 




-25.0 



i i i i i — i— i — i— i — i i i i i i i i i i 



-60 -40 -20 20 40 60 
Angle [deg] 

Fig. 3a. Antenna patterns for the BLTSA designed for 348 GHz but 
measured at 270 GHz 



Page 284 



Third International Symposium on Space Terahertz Technology 




25.0 I 



-I I > — ' — u 



-60 -40 -20 20 40 60 
Angle (deg] 
Fig. 3b. Antenna patterns for the BLTSA designed for the 348 GHz design 
measured at 348 GHz 



H 



D Dx 



O.Oi •■ ■ ■ ■ 




-20 
Angle (deg] 

Fig. 3c. Antenna'patterns for the BLTSA designed for the 348 GHz design 
measured at 802 GHz 



At 802 GHz the antenna beam is narrower and more symmetric (39 °- 43° at -10 dB 
level and the directivity is approximately 13 dB) with slightly lower sidelobes than 
the pattern at 348 GHz, also the D-plane cross-pol level is lower (-8 dB). Fig. 4a. b. The 
improved pattern at 802 GHz was expected since the membrane is relatively thicker 



Third International Symposium on Space Terahertz Technology 



Page 285 



(expressed in wavelengths) and closer to the optimum thickness. In fact, the optimum 
frequency for these membranes Is approximately 4 THz Thus, at THz frequencies. It is 
expected that the antenna pattern would be even more symmetric and the D-plane 
crosspol level could be as low as -10 to -15 dB below the co-polarized level. 



0.0 



E, 348GHz 

H, 348GHz 
E, 802GHz 
H. 802GHz 




-25.0 

-60 -40 -20 20 40 60 

Angle (degl 

Fig. 4a. Measured E- and H plane of the 802 GHz design at 802 GHz. and the 
348 GHz design at 348 GHz 
0.0 




-25.0 



D, 348GHz v 

Dx, 348GHz 



_i — , — , — i — u 



D, 802GHz : 
Dx, 802GHz 



-60 -40 -20 20 40 60 
Angle [degj 

Fig. 4b. Measured D- and Dx plane of the 802 GHz design at 802 GHz, and 
the 348 GHz design at 348 GHz 



Page 286 



Third International Symposium on Space Terahertz Technology 



The performance of the BLTSA end-fire antennas on 1.7 |jm thin membranes is sum- 
marized in the table below. 



Frequency GHz 


270 


348 


370 


802 


E- plane -10 dB beam width 

side lobe level dB 


59° 
-12 


50° 
-19 


49° 
-19 


43° 
-20 


H-plane -10 dB beam width 

side lobe level dB 


64° 
-8 


55° 
-11 


53 

-11 


44° 
-11 


D-plane -10 dB beam width 

side lobe level dB 


50° 
-7 


43° 
-10 


42° 
-10 


39° 
-12 


D-cross lobe level dB 


-4 


-6 


-6.5 


-8 


Directivity dB 


11 


12 


12 


13 



Table 1 . Compiled measured antenna data. All antennas were fabricated 
on a 1.7 urn thick SiC«2/Si3N4 membrane. The thick frames around 348 
and 802 GHz indicate the design frequencies. The 348 GHz antenna was 
also measured at 270. 370 and 802 GHz. The 802 GHz antenna was only 
measured at 802 GHz, 



Scale measurements at 45 GHz show that the E and H-plane patterns agree reasonably 
well with the patterns at the higher frequencies. Fig. 5. These antennas were made on 
25.4 pi thick Kapton™ foil (£^3.5). Note that, this substrate thickness is compara- 
tively thicker (measured in wavelengths) at the scale frequency than the thickness of 

the dielectric in the 270 - 802 GHz measurements. 




— H-plane 
E-plane 



CQ 

u 

> 
OS 



-10 




-20 20 
Angle [deg] 

Fig. 5. E and H-plane patterns of a scale model of the BLTSA. The 
measurements are performed at 45 GHz. 



Third International Symposium on Space Terahertz Technology 



Page 287 



THEORY 
The end-fire slot-line antennas have been analyzed by using method previously 
described by e.g. Janasawamy [6]. In this method, the antenna tapering is 
approximated by a "stalrcase'-function in a number of steps of different widths but 
equal lengths, as shown in figure 6. Thus, the slot-line antenna can now be treated as 
a linear array of apertures each fed with different phases and amplitudes. Fur- 
thermore, power conversion is applied to relate the amplitudes and fields in 
neighbouring apertures. The characteristic impedance and the wavelength of each 
slot is calculated by using a spectral domain technique [7]. The far field pattern of the 
antenna is calculated by applying an appropriate Green's function and adding the 
field contributions from all apertures. In these calculations we ignore the reflections 
from the discontinuities in the antenna taper (which has been confirmed in computer 
simulations). 

The theoretical patterns agree reasonably well with measured patterns. Fig. 7, 8, con- 
sidering the approximations in the calculations. The calculated and measured beam 
width correspond very well, and it is possible to predict level and position of the side- 
lobes within 2 dB and 5°. The biggest difference between the calculated and measured 
patterns is in the crosspolarized D-plane. where the theory predicts a level 2-3 dB 
higher than the measured level. 




Fig. 6. Approximation of the slot-line antenna by an array of slots. 



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Third International Symposium on Space Terahertz Technology 



0.0 




-25.0 



-60 \/-40 -20 20 401/ 60 
Angle [deg] 

O.Or 



-5.0 



CQ 

? -10.0 

1 

§2 -15.0 

J2 

OS 

-20.0 



-25.0 




-60 -40 -20 20 40 60 
Angle [deg] 



0.0 



- - - D (exp) 
V - -Dx (expj) 




-25.0 



-60 -40 -20 20 40 60 
Angle [degl 

Fig. 7. Calculated (th) and measured (exp) antenna patterns at 348 GHz 



Third International Symposium on Space Terahertz Technology 



Page 289 



0.0 




-60 -40 -20 20 40 60 
Angle (deg] 
0.0 



£ -20.0 



-25.0 



-1 — i — | — i — I — r— r 



1 — I — I — I — I — I — I — I— 

D (exp) 
i Dx (exp i 




-60 -40 -20 20 40 60 
Angle (deg] 

Fig. 8. Calculated (th) and measured (exp) patterns at 802 GHz 



Page 290 Third Interna lional Symposium on Space Terahertz Technology 



REFERENCES 

[1] S.N. Prasad, and S. Mahapatra. A novel MIC slot-line aerial Proc. 9th Eur. 

Microwave Conf.. pp. 120-124. Brighton. UK, 1979. 
[2] P.J. Gibson, The Vivaldi aerial 9th Eur. Microwave Conf.. pp. 101-105. Brighton. 

UK, 1979. 
(3) K.S. Yngvesson. D.H. Schaubert. T.L. Korzeniowski, E.L. Kollberg, T. Thungren. 

and J. Johansson. Endjire tapered slot-line antennas on dielectric substrates, 

IEEE Trans. Antennas. Propagat.. vol. AP-33. N.12. ppl392-1400. 1985. 
[41 G. Rebeiz. Millimeter-wave and terahertz tntegrated-clrcuit antennas, preprint, 

to appear in the October issue on Space terahertz Technology in the IEEE- 
Proceedings. 
[51 P.R. Acharya. J. Johansson, and E.L. Kollberg. Slotltne antennas for millimetre 

and sub millimetre wavelengths. Proc. 20th Eur. Microwave Conf.. pp. 353-358. 

Budapest. Hungary. Sept 1990. 
[61 R. Janasawamy, D. Schaubert. Analysis of Tapered Slotline Antenna, IEEE 

Trans. Antennas Propgat.. vol. AP-35. no.9. pp. 1058-1065. Sept. 1987. 
[7] G. Johansson, P.R. Acharya. J. Johansson. Determination of Slot Line 

Characteristics, Technical Report No. 97L. 1991. Chalmers University of 

Technology. 



f 

Third International Symposium on Space Terahertz technology Page 291 




93-27752 

QUASI-OPTICAL ANTENNA-MIXER- ARRAY DESIGN 

FOR TERAHERTZ FREQUENCIES 

Yong Guo 

Department of Electrical and Computer Engineering 

College of Engineering, Clemson University 

Clemson, SC 29634 

Kent A. Potter, David B. Rutledge 
Division of Engineering and Applied Science 
California Institute of Technology- 
Pasadena, CA 91125 

Abstract 

A new quasi-optical antenna-mixer-array design for terahertz frequencies is presented 
in this paper. In the design, antenna and mixer are combined into an entity, based on 
the technology in which millimeter-wave horn antenna arrays have been fabricated in 
silicon wafers. It consists of a set of forward- and backward-looking horns made with 
a set of silicon wafers. The front side is used to receive incoming signal, and the back 
side is used to feed local oscillator signal. Intermediate frequency is led out from the 
side of the array. Signal received by the horn array is picked up by antenna probes 
suspended on thin silicon-oxynitride membranes inside the horns. Mixer diodes will be 
located on the membranes inside the horns. Modeling of such an antenna-mixer-array 
design is done on a scaled model at microwave frequencies. The impedance matching, 
-RF and LO isolation, and patterns of the array have been tested and analyzed. 

I. Introduction 

In submillimeter-wave and terahertz frequency systems, because of their much shorter 
wavelengths compared with microwave systems, waveguide circuits become much 
smaller, which makes them very difficult and expensive to build. However, quasi- 
optical components provide a solution to this problem. Quasi-optical antenna-mixer- 
array combines antennas and mixer circuits into a single entity. The design is based 
on an existing technology by which dipole excited integrated-circuit horn antennas are 
made in silicon [1]. The horn antennas consist of probes suspended on a thin oxyni- 
tride membrane inside pyramidal horns which are chemically etched in silicon. The 
antennas are free of dielectric losses and have plenty of space for electronic intercon- 
nections between the probes. The aperture efficiency of these etched horn antennas 
has been improved to 72% [2]. Recent research shows that the experimental results 
agree well with the theoretical analysis, including radiation patterns and resonant 
dipole impedances [3]. Various antenna probes inside the pyramidal horns have also 
been studied [4]. 

This integration of antenna and mixer eliminates the need for RF and LO circuit- 
fashion connections. Such construction offers a device with potential of smaller size, 



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Third International Symposium on Space Terahertz Technology 



lighter weight, more ruggedness and less cost, as compared to conventional methods. 
Moreover, this kind of design potentially can be mass produced by standard integrated- 
circuit technology. The applications of the integrated-circuit antenna-mixer array 
include imaging systems, radars, and satellite communications. 

II. Horn Structure and Mixer Circuitry Design 

In order to avoid the difficulty to supply an LO power at a frequency close to RF, 
a subharmonically-pumped antenna-mixer array is designed which is pumped by an 
LO at only half of the RF frequency. Since the RF and the LO frequencies differ 
by approximately a factor of two, in principle, it is easier to realize the isolation 
between the RF and the LO. Furthermore, spurious responses associated with the 
odd harmonics of the LO can be rejected by using an antiparallel diode pair. The 
subharmonically-pumped horn-antenna-mixer array is shown in Figures 1 and 2. It 
consists of a set of forward- and backward-looking horns facing back to each other 
made silicon wafers. Every four RF horns are provided with one LO horn which is a 
rectangular-shaped horn. The spacing between RF horns is 1A. By using the sub-array 
concept, every four RF horns can be taken as a sub-array. Four RF horns, together 



Trough 



Horn 



Membrane 




Monopole probe 




Coplanar strips Loaded dipole probe 
(a) 



(b) 



Figure 1 The horn structure of the subharmonically-pumped mixer design, one LO horn 
corresponding to four RF horns; (a) LO horns, the trough made of two silicon wafers is put 
on the top of the horns, (b) RF horns, monopoles are used for the RF reception. 



Third International Symposium on Space Terahertz Technology 



Page 293 



Dipole probe — - 



pr? — Silicon oxynitride 
r membrane 




P=1.0Xrf 
Li=0.26\rf 
L=0.40Xrf 
D=0.11Xlo 
d: varies 



Membrane wafer 



Monopole probe 




Mixer diode 
position 



(a) 



(b) 



Figure 2 (a) Cross-sectional view of the antenna-mixer array, (b) The mixer circuit design 
for a unit cell; four monopoles for the RF reception and one dipole for the LO reception; 
both RF and LO are detected by the mixer diodes located in the center of the unit cell; IF 
is led out from the ends of the LO horn through a coplanar-strip transmission line. 

with one LO horn, form a unit cell. This design will keep the best symmetry, and the 
beam patterns of the sub-array will be improved by a factor of 4 compared with that 
of the single RF horn. Since the size of the LO horns should be twice that of the RF 
horns, half of the area on the LO side in each unit cell would be left unused. This 
would cause strong reflection from the flat surface, namely, a 3 dB reflection loss. In 
order to eliminate this 3 dB reflection loss, a structure is designed to be placed on the 
top of the LO horns. The structure has a long trough on each row of the LO horns 
and will fill up the space between the LO horns to converge the incoming energy into 
the LO horns. Figure 1 shows the antenna-mixer array with 2x2 LO horns looking 
from the LO side and 4x4 RF horns looking from the RF side. The cross-sectional 
view is shown in Figure 2(a). The mixer circuit design is shown in Figure 2(b). Every 
monopole from each of the four RF horns will couple the RF signal down to the LO 
horn through a coplanar-strip transmission line. A dipole probe is employed to receive 
the LO. The mixer diodes are located in the center of the LO membrane, and the IF 
is led out from the ends of the LO horn. The dipole probe is loaded on the ends near 
the sidewalls with a short stub, which, as a result, could compensate for the capacitive 
characteristic impedance of the short dipole probe. 



Page 294 Third International Symposium on Space Terahertz Technology 



III. Modeling 

The mixer array design has been tested on a scaled model sub-array, which has four 
RF horns and one LO horn. The gain of such a sub-array will be increased by a factor 
of 4 compared with a single horn. Based on this scaled model sub-array, antenna 
impedances and receiving patterns were measured. Antenna probes are required not 
only to couple the free-space wave energy to the mixer circuit but also to provide a 
suitable impedance, the embedding impedance, to the mixer diodes. This impedance 
over a wide frequency range is also important for mixer performance because various 
frequency components exist in the mixer circuits. In order to achieve good isolation 
between the RF and the LO, as well as to match the impedances of the RF and the 
LO to the diode impedance, various mixer circuits have been tested. Trade-off has 
been made among the impedance match and the isolation between the RF and the 
LO so as to minimize losses. 

The actual model consists of two square RF horns and a half rectangular LO horn, 
a half of the designed unit-cell, sitting on a big copper-clad circuit board which was 
used as an image plane. The monopole built in each of the RF horns will couple 
the incoming signals to the loaded dipole in the LO horn through the coplanar- strip 
transmission line. A small channel in the middle of the horns will let the monopole 
probes go through between the LO and RF horns. The design frequency for the RF 
is 10 GHz and the LO is 5 GHz, corresponding to the wavelength of 3 cm and 6 cm, 
respectively. The opening of RF horns is 1 Ar F square, while the height of the LO 
horn is Alo/2, and the LO horn width is 1 Alo- Mixer diodes are to be placed in the 
center of the dipole probe in the LO horn. An SMA bulkhead feed-through connector is 
soldered from the back of the circuit board to the place where the diodes are supposed 
to be. The inner conductor of the connector is soldered to the dipole probe, and the 
outside conductor is soldered to the circuit board used as a ground plane. 

Measurements were done on an HP 8510 Network Analyzer and data were collected by 
a PC. Full two^-port calibration was made in order to measure not only the reflection 
coefficients but also the receiving properties and the isolation between the RF port 
and LO port by measuring the transmission coefficient Si2- For this purpose, a broad- 
band horn antenna is used as a transmitting horn which has a working frequency 
range from 2 GHz to 18 GHz. The measured impedances are marked on the Smith 
chart in Figure 3(a). Although the impedances were measured on only a half unit-cell, 
consisting of two RF horns and a half LO horn, the impedances in a full unit-cell 
can be easily obtained by doubling those measured impedances in the half unit-cell. 
The impedances in Figure 3(a) are plotted by using Puff, a software CAD program [5]. 
Both the RF and the LO impedance should be matched to 50 Q. because each beam- 
lead diode in the antiparallel diode pair has a resistance of about 100 fi. In the 
graph, when the loading stub on the dipole decreases in length, the LO impedance at 
5 GHz changes from the inductive to the capacitive impedance, passing the resonant 
resistance at about 50 ft, which is a very good matching impedance for the diode pair. 
This LO impedance of the circuit can be regarded as the LO dipole-probe impedance 



Third International Symposium on Space Terahertz Technology 



Page 295 



parallel with the impedance of the coplanar transmission line plus the RF probes. 
At 5 GHz, the impedance of the coplanar transmission line plus the RF probes is 
very high as is illustrated by the LO frequency mark "5" when the entire LO dipole 
probe is taken away. Hence, the resonant LO impedance is mainly determined by 
the loaded dipole probe and is relatively independent of rest of the circuits. On the 
other hand, the RF impedances at 10 GHz are pretty high and independent of the 
loading-stub length changes. The average value of those RF impedances at 10 GHz is 
about 84 + ;82 ft. 



The normal-incident power receptions by the RF and the LO horns were tested over 
a wide frequency range, from 2.0 GHz to 12.0 GHz. This was done by putting a wide- 
band transmitting horn in front side of the RF horns or the LO horn. Figure 3(b) 
shows the measured power received by the RF horns when the transmitting horn 
is on that side (solid line) and by the LO horn when the transmitting horn is that 
side (dashed line). At the LO frequency of 5 GHz, the difference between the LO and 
the RF power is defined as the LO-RF isolation. The higher isolation, the lower the 
coupling loss, under the condition that other parameters stay the same. Similarly, 
the RF-LO isolation is the power difference between RF and LO at the RF frequency 



■ LO at 5 GHz 
• RF at 10 GHz 



1:d=0.12ALO 
2:d=0.08XLO 
3:d=0.06XLO 
4:d=0.02\LO 
5: d=0.00 Xlo 
D=0.00 Xlo 



-10 




-0.2 



-60 



-70, 



H I I | I I I | I I I | I I I | I I T" 

LO-RF isolation 13dB 




_i i i i i i_ 



i i i 



j i_ 



RF side 
LO side 



i I i 



6 8 

Frequency, GHz 



10 



12 



(b) 

Figure 3 (a) The circuit impedances is indicated on the Smith chart with respect to the 
different loading-stub length d\ measured at 10 GHz for the RF and 5 GHz for the LO. 
(b) Measured normal-incident power, received by the RF horns (solid line) and by the LO 
horn (dashed line) with equal distance. 



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Third International Symposium on Space Terahertz Technology 



Loss component 


Simple-probe design 


Split-LO-probe design 


RF mismatch, dB 


1.7 


4.9 


LO mismatch, dB 


0.0 


0.2 


RF-LO coupling, dB 


1.3 


0.0 


LO-RF coupling, dB 


0.2 


0.2 



Figure 4 Comparison of the impedance- mismatch losses and the coupling losses between the 
two different mixer circuit designs. 

of 10 GHz. The measured RF-LO isolation is 6 dB or a 25 % loss. If the LO probe 
is split into two, by tuning the spacing between two LO probes, minimum RF-LO 
isolation of 20 dB was achieved. As a comparison, of the simple-probe design and 
the split-LO-probe design, the impedance-mismatch losses and the coupling losses of 
the two designs are listed in the table in Figure 4. For both designs, the biggest loss 
comes from the RF-impedance mismatch, 1.7 dB for the simple probe and 4.9 dB for 
the split-LO-probe. 

IV. Conclusion 

A new antenna-mixer array design has been presented, which potentially can be made 
and used at submillimeter and terahertz frequencies. Modeling work shows that, for 
this design, compromises have to be made between the RF-impedance mismatch, the 
LO-impedance mismatch and the RF-LO coupling losses. In some applications, if cer- 
tain LO loss is tolerable, then lower RF loss can be achieved (which means lower con- 
version loss) by sacrificing some LO impedance-mismatch losses and LO-RF coupling 
losses. The mixer elements could be either Schottky diodes [6] or superconducting SIS 
mixers [7]. 

V. Acknowledgements 

We appreciate the support of Aerojet Elect roSystems Co., Azusa, CA. and the Army 
Research Office through the Jet Propulsion Laboratory. 

VI. References 

[1] G. M. Rebeiz, D. P. Kasilingam, Y. Guo, P. A. Stimson, D. B. Rutledge, "Mono- 
lithic Millimeter- Wave Two-Dimensional Horn Imaging Arrays," IEEE Transactions 
on Antennas and Propagation, September, 1990. 

[2] Y. Guo, K. Lee, P. Stimson, K. Potter, and D. Rutledge, "Aperture Efficiency 
of Integrated- Circuit Horn Antennas," Microwave and Optical Technology Letters, 
January, 1991. 

[3] G. V. Eleftheriades, W. Y. Ali-Ahmad, L. P. Katehi, G. M. Rebeiz, " Millimeter- 
Wave Integrated-Horn Antennas: Part I-Theory, Part II-Experiment " IEEE Trans- 
actions on Antennas and Propagation, November, 1991. 



Third International Symposium on Space Terahertz Technology Page 297 



[4] Y. Guo, J.C. Chiao, K.A. Potter, D.B. Rutledge, "Probe Modeling for Millimeter- 
Wave Integrated- Circuit Horn Antennas," submitted to the IEEE AP-S International 
Symposium, July 18-25, 1992, Chicago, Illinois. 

[5] R.C. Compton, S.W. Wedge, D.B. Rutledge, "PuiE Computer Aided Design for 
Microwave Integrated Circuits," Caltech Press, January, 1990. 

[6] T.W. Crowe, W.C.B. Peatman, "GaAs Schottky Diodes for Mixing applications 
Beyond ITHz," Second International Symposium on Space Terahertz Technology, 
JPL, Pasadena, CA, February 26-28, 1991. 

[7] M.J. Wengler, N. Dubash, G Pance, R.E. Miller, "High Gain and Noise in SIS 
Mixers at the Submillimeter Wavelengths," Second International Symposium on Space 
Terahertz Technology, JPL, Pasadena, CA, February 26-28, 1991. 



Page 298 Third International Symposium on Space Terahertz Technology 



5*7-32- N93-2? 753 

ANALYSIS OF A NOVEL NON-CONTACTING 
WAVEGUIDE BACKSHORT 

T. M. Weller and L. P. B. Katehi, 
University of Michigan NASA Center for Space Terahertz Technology 

W. R. McGrath, 
Jet Propulsion Laboratory Center for Space Microelectronics Technology 



M^ 



ABSTRACT A new non-contacting waveguide backshort has been developed for mil- 
limeter and submillimeter wave frequencies. The design consists of a metal bar with rect- 
angular or circular holes cut into it, which is covered with a dielectric (mylar) layer to form 
a snug fit with the walls of a waveguide. Hole geometries are adjusted to obtain a periodic 
variation of the guide impedance on the correct length scale, in order to produce efficient 
reflection of rf power. It is a mechanically rugged design which can be easily fabricated for 
frequencies from 1 to 1000 GHz and is thus a sound alternative to the miniaturization of 
conventional non-contacting shorts. To aid in high-frequency design, a rigorous full-wave 
analysis has been completed which will allow variations of the size, number and spacing of 
the holes to be easily analyzed. This paper will review the backshort design and the method 
developed for theoretical characterization, followed by a comparison of the experimental and 
numerical results. Low frequency models operating from 4-6 GHz are shown to demonstrate 
return loss of > —0.2 dB over a 33% bandwidth. The theory is in good agreement with 
measured data. 



Third International Symposium on Space Terahertz Technology Page 299 

INTRODUCTION 

Waveguides are used in a wide variety of applications covering a frequency range from 
1 GHz to over 600 GHz. These applications include radar, communications systems, mi- 
crowave test equipment, and remote-sensing radiometers for atmospheric and astrophysical 
studies. Components made from waveguides include transmission lines, directional couplers, 
phase shifters, antennas, and heterodyne mixers, to name a few. In addition to the many 
commercial applications of waveguides, NASA needs such components in radiometers oper- 
ating up to 1200 GHz for future space missions, and the Department of Defense is interested 
in submillimeter wave communications systems for frequencies near 1000 GHz. 

One of the most frequent uses of waveguide is as a variable length transmission line. 
These lines are used as tuning elements in more complex circuits. Such a line is formed by a 
movable short circuit, or backshort, in the waveguide. A conventional approach is to use a 
contacting backshort where a springy metallic material, such as beryllium copper, makes DC 
contact with the broadwalls of the waveguide. The contacting area is critical, however, and 
must be maximized to produce an acceptable short circuit. These backshorts are excellent in 
that they provide a good short circuit over the entire waveguide band. The contacting areas 
can degrade, however, due to wear from sliding friction. It is also extremely difficult to get 
a uniform contact at frequencies above 300 GHz, where the waveguide dimensions become 
0.5 mm x 0.25 mm for the 300-600 GHz band. 

An alternative approach is the non-contacting backshort shown in Figure 1. A thin 
dielectric layer (such as mylar) prevents contact and allows the backshort to slide smoothly. 
In order to produce an rf short and, hence, a large reflection, this backshort has a series 
of high- and low-impedance sections which are approximately -f in length, where \ g is the 
guide wavelength. The rf impedance of this design is given approximately by [1] 

Zrf = (|^-r Z low (1) 



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Third International Symposium on Space Terahertz Technology 



WAVEGUIDE 



vN 




\ 



MYLAR 
INSULATOR 



wmM%#M;M&WffimMWg%%%\ 



(FRONT) 



v\ 








M?Mmm,*-m 



mmmmmmmm\ 



\S HTfiH >J 



LOW V fflGH ^ WAVEGUIDE 

IMPEDANCE IMPEDANCE OPENING 

SECTIONS SECTIONS 

Figure 1: Cross sectional view of a conventional non-contacting backshort. 

where Zi ow is the impedance of the thick (low-impedance) sections, Zhigh is the impedance 
of the thin (high-impedance) sections, and n is the number of sections. Beginning near 
100 GHz, the thin high- impedance sections become difficult to fabricate, and fabrication 
may not even be feasible beyond 300 GHz. It would also be difficult to have the short slide 
snugly inside the waveguide at these high frequencies, as the thin sections would be very 
weak. To circumvent these problems, a novel non-contacting backshort design has recently 
been developed [2, 3] which is suitable for millimeter and submillimeter wave operation. 
It is a mechanically rugged design which can be easily fabricated for frequencies from 1 
to 1000 GHz, and is thus a sound alternative to the miniaturization of conventional non- 
contacting shorts. Previously, however, the new backshort was optimized empirically using 
low-frequency models. This paper will discuss the new design and outline a new method 
developed for theoretical characterization. The formulation is a rigorous full-wave analysis 
which involves both mode-matching techniques and a coupled set of space domain integral 
equations. A description of the experimental setup is included, followed by a comparison of 
experimental and theoretical results. The new theoretical formulation fits these results well. 



Third International Symposium on Space Terahertz Technology 



Page 301 



DIELECTRIC COVER 




HOLES 



(FRONT) 



2«S 






3^ 




DIELECTRIC COVER 



METAL SHORT 



(into waveguide) 

Figure 2: The new non-contacting backshort design, shown with three rectangular holes. The size, 
shape, and spacing of these holes are important in determining the rf properties of the short. S 
is the spacing, L\ is the length, and Li is the width of each hole. The front of the backshort is 
inserted into the waveguide opening. 

NOVEL NON-CONTACTING BACKSHORT DESIGN 



The novel non-contacting backshort has the merits of easy fabrication up to Thz frequen- 
cies, flexibility of design, and very good performance over relatively broad bandwidths. The 
important features are briefly reviewed here. In order to obtain a large reflection, a non- 
contacting backshort must provide a periodic variation of guide impedance on the correct 
length scale. This is accomplished in the new design by either rectangular or circular holes, 
with the proper dimensions and spacing, cut into a metallic bar. A representative design is 
shown in Figure 2. This bar is sized to fill the waveguide cross-section and slide smoothly 
with a dielectric (mylar) insulator along the broadwalls. The holes replace the thin-metal, 
high-impedance sections in the conventional design shown in Figure 1. Since the holes ex- 
tend completely through the bar, this yields a higher impedance than the corresponding 
sections in the conventional design. Thus, the high-to-low impedance ratio is larger in the 
new design. In addition, the electromagnetic fields are concentrated near the central axis of 



Page 302 Third International Symposium on Space Terahertz Technology 

calculated return loss for a backshort with no holes, inserted about 4.5 inches into the end 
of the waveguide, and covered with mylar (e r = 3.35). The waveguide opening is assumed to 
present a Cl impedance (i.e., it is covered with a metallic plate). Although the gap height 
is only 2.5% of the waveguide height, roughly 65% of the incident power is lost at resonance 
due to finite conductor and dielectric loss. The utility of the holes, then, is to minimize or 
eliminate these dropouts. 

THEORETICAL CHARACTERIZATION 

The theoretical characterization of the JPL backshort design is performed using a com- 
bination of two well known full-wave analysis methods, namely mode-matching and the 
application of equivalent magnetic currents in a space domain integral equation. In what 
follows, the approach will be outlined and the major governing equations presented. It is 
noted here that the symmetry of the backshort about the x-z plane (parallel to the plane of 
the waveguide broadwalls) has been utilized to reduce the number of unknown parameters. 
Furthermore, only rectangular holes (not round) have been considered in order to simplify 
the analysis. Neither of these points, however, are necessary restrictions in the formulation. 

A discussion of the analysis is aided by the schematic in Figure 4, which represents the 
cross-sectional view of a backshort with two holes, inserted a distance d into the end of a 
rectangular waveguide. The structure is symmetric about the x-z plane, with equal dielectric 
regions (which are the dielectric covers shown in Figure 2) above and below the metal short. 
The problem of interest is to determine the reflection coefficient for the dominant waveguide 
mode, travelling in the +z direction, at the front of the backshort (z = 0). 

The formulation is based on the decomposition of the problem into two primary compo- 
nents. In the first part, we wish to compute the scattering matrix [S] at z = 0, as depicted in 
Figure 4. As [S] represents simply the scattering at a waveguide discontinuity, the presence 
of the holes may be neglected and thus becomes decoupled from the problem at hand. The 



Third International Symposium on Space Terahertz Technology 



Page 303 






.3 

6 

3 
o 




-5.0 



4.0 4.4 4.8 5.2 5.6 6.0 

Frequency, GHz 

Figure 3: Calculated return loss versus frequency for a backshort with no holes, where the gap 
height is 2.5% of the total guide height. 

the waveguide, such that the holes are effective in producing large correlated reflections, and 
thus acting as an efficient rf short. The new design is also easy to fabricate and can be used 
at any frequency between 1 GHz and 1000 GHz. For very high frequencies, above 300 GHz, 
the metallic bar is a piece of shim stock polished to the correct thickness. The holes can 
be formed by drilling, punching, or laser machining, or they can be etched using common 
lithography techniques. 

It is important to note that the holes are a critical factor in obtaining efficient reflection 
from the non-contacting short. With the backshort inserted in the waveguide, a cavity forms 
between the metal bar and the broadwall of the waveguide, in the region occupied by the 
dielectric insulator. This cavity is terminated by the large discontinuities at the front of 
the short and at the waveguide opening. (This is more clearly illustrated in Figure 1 for 
the conventional design.) Deep dropouts in the return loss will occur at frequencies for 
which this cavity resonates, even though the height of each gap may be only a small fraction 
of the total waveguide height. The effect is well illustrated in Figure 3, which shows the 



Page 304 



Third International Symposium on Space Terahertz Technology 



REGION 
I 



REGION (into waveguide) 



-#- 



WAVEGUIDE 
FLANGE 



[S] 



M 



upper 



M 



«pper 



M 



lower 



HOLE1 



M 



lower 



HOLE 2 



(metal) 



PLANE OF 



(dielectric-filled gap) 



Z=0 



WAVEGUIDE 
BROADWALL 



7? 



-ff- 



SYMMETRY 



k 



Z=d 



Figure 4: Cross- sectional schematic diagram (not to scale) of a two-hole non-contacting backshort, 
inserted a distance d into the end of a waveguide. The waveguide broadwalls are on the top and 
bottom in the figure. 

well documented mode-matching method, which has been used to solve a variety of wave- 
guide problems [4, 5, 6] is applied to determine [S], With this method, the fields at each 
side of the reference plane (2 = 0) are expanded in infinite series of orthogonal mode pairs 
(e.g. TE-to-z and TM-to-z), and continuity of the tangential electric and magnetic fields 
is enforced to determine the scattered field amplitudes. This results in the following set of 
generalized equations, 

00 00 00 00 

£(«.' + O^f '' + I>m + &)W = IK' + tf W + IK' + %)W (2) 



n,m 



n,m 



E(«i - «)*? •' + IK - &)$«>' = £ -k" - 6J')*?." + E -(«" - tf)^" 0) 



n,m 



n,tn 



n,m 



where (2) satisfies continuity of tangential E and (3) satisfies continuity of tangential H. In 
the above, a and b represent the coefficients for waves travelling toward and away from the 
reference plane, respectively. The subscripts e and m are for TE-to-z and TM-to-z, while 



Third International Symposium on Space Terahertz Technology Page 305 

the superscripts denote the field type (electric or magnetic) as well as the region to which 
they pertain (to the left or right of the reference plane). The vector functions $ contain 
the appropriate constants and x- and y-dependencies for the transverse components of the 
respective fields. At this point inner-products are formed using $f ,J and $^ ,J with (2), 
and $^- /7 and $™' 7/ with (3). As these inner-products involve integration over the guide 
cross-section, a system is linear equations results due to the orthogonality of the modal 
components. This system is assembled into a matrix representation and, after inversion, the 
solution is expressed as 

{b} = {a} T [S] (4) 

With [S] determined, the unknown scattered- field amplitudes {b} are found from (4) given 
the known incident-field amplitudes, {a}. It is noted that the presence of a termination at 
z = d (see Figure 4) is easily accounted for by assigning 

b 1 = a I Sn+a I S 1 2(I-r L S 22 )- 1 T L S 21 , (5) 

where I is the identity matrix and [Tl] is a matrix which accounts for the reflection at 
z = d. As shown in the figure, we assume that the waveguide opening is terminated in a 
complex load Zl for simplification. (This approximation is necessary because the conditions 
outside the short are difficult to control experimentally and, likewise, difficult to accurately 
characterize analytically. This will be discussed further in the section on results.) The matrix 
[Ti] is thus a diagonal matrix of elements 

< r ^ - ferf °- Hd < 6 > 

In (6), Z x g and Y z are the guide impedance and propagation constant, respectively, for the 
i th TE/TM mode. Conductor and dielectric loss may be included in the factor Y z - 

The second principle step in the formulation is to apply the space domain integral equa- 
tion technique to solve the boundary value problem at the aperture of each of the holes. The 



Page 306 Third International Symposium on Space Terahertz Technology 

introduction of the equivalent magnetic currents, M upper and M loweT (see Figure 4), allows 
the hole openings to be closed by an imaginary metallic surface, provided that no natural 
boundary conditions are violated. This is a crucial step in that it transforms the backshort 
structure into a combination of a simple rectangular waveguide, which is the dielectric-filled 
gap region, and a series of isolated metallic cavities, which are the holes. These unknown 
magnetic currents radiate electromagnetic fields in the dielectric region, and a modified form 
of (5) therefore results when treating a backshort with holes. The new expression is 

b = a Sn + a S\2(I — TlS22)~ ^lS 2 \ + 

{(F>T L + F<)S 22 (I - T L S 22 )- 1 T L + F>T L + F<}S 21 . (7) 

Note that the only unknown variables in this equation are F K and F > , as the components 
of the matrix [S] and [Tl] have previously been determined. These unknown components 
are functions of the imposed equivalent magnetic currents. 

The solution for the unknown surface currents is uniquely determined by enforcing con- 
tinuity of the total tangential fields across the hole apertures. This insures that the natural 
boundary conditions of the original problem are preserved. Continuity of the tangential 
electric field is satisfied immediately by setting M upper = -M lower = M. Assuming a 
backshort with N holes, continuity of the magnetic field at the k th hole leads to the following 
space domain integral equation (SDIE) in the unknown M: 

— n x ti = n x [Ji + 

N I r r 

d S '(^-(k 2 I+VV) ■G B )'Mn) + 



n=l 




5„ we// 



f f <fa'(^L_(*'/ + w) • G c ) ■ M k ) (8) 

J Js k uje ft 

In the above, H ,nc represents the known incident magnetic field, which results from scattering 
of the incoming wave at the waveguide step discontinuity (the reference plane). It is expressed 



Third International Symposium on Space Terahertz Technology Page 307 

in terms of TE and TM modes, the coefficients being given by 

a 11 = a I S 12 (I-T L S 22 )- 1 T L 

b n = a n S 22 (9) 

for +z and —z travelling waves, respectively. Gb and Gc represent the dyadic Green's func- 
tions for an infinite rectangular waveguide and a metallic cavity, respectively. Closed-form 
expressions for these functions can be derived using well established boundary value for- 
mulations [7]. The use of an infinite- waveguide potential in the dielectric-filled gap region, 
which does not account for the actual finite length of uniform guide, is possible by consid- 
ering the fields to be a superposition of primary and scattered components. The primary 
components satisfy boundary conditions at the source, and radiate away from M in the 
presence of matched conditions in either direction. These components are precisely those of 
the second term on the right hand side of (8). The scattered components are required to 
satisfy the boundary conditions away from the source, at the discontinuities at z = 0, d and 
are also functions of M. Expressions for these fields, which are represented by H scat in (8), 
are similar in form to the primary components but also include factors from the scattering 
matrix [S] and the matrix [r^]. 

The final step in the formulation is to solve the coupled set of integral equations which 
results from enforcing (8) over all N holes. This set may be reduced to a system of linear 
equations by applying the method of moments (Galerkin's method) [8]. This approach has 
been proven to yield excellent results and the convergence characteristics have been well 
documented [9, 10, 11]. Using this procedure, the aperture of each hole is first divided into 
discrete subsections using a rectangular grid. The unknown currents are then expanded in 
terms of overlapping subsectional rooftop basis functions of the form, 

M = Y.{^tM x ')<f>^') + ^t j <j> j {x')f i {z')) (10) 



Page 308 



Third International Symposium on Space Terahertz Technology 



sin[k (u>'-u> n .i)) . r , <w ' <w 



<t> n (w') = < 



1 if u; n _i <w'< w n +i 

else 

where Mfj and M£ are constant coefficients, / n is the length of the n th subsection in the 
^-direction, and k is the wave number in the medium. This expansion is inserted into the 
integral equation, and inner-products are then formed using weighting functions which are 
identical to the basis functions. The coupled equations are thereby reduced to the following 
matrix form: 



(Y xx ) (Y X2 ) 



\ 



t 



{M x } 



\ 






(11) 



(Y«) (Y„) J \ {M*} ) 
where (Y^(C, £ — x, z)) represents blocks of an admittance matrix. The unknown current co- 
efficient vectors {Mfj} and {M£} are then determined by solving (11). With M determined, 
all elements of (7) may be computed and the solution is complete. 

MEASUREMENT TECHNIQUES 

The backshort design was optimized by testing the performance in WR-187 band wave- 
guide (3.16 GHz - 6.32 GHz), for which the dimensions are 47.5 mm x 22.1 mm. The 
dielectric layer around the metal short was formed by stacking sheets of mylar tape. The 
magnitude and phase of the reflection coefficient were measured with an HP 8510B Vector 
Network Analyzer. A commercially available coaxial-to-waveguide transition connected the 
waveguide to the network analyzer. This measurement system was calibrated using two off- 
set contacting shorts set at -^ and -g 4 , and a sliding waveguide load. Subsequent verification 
using a contacting short indicated a measurement error of about ±0.2 dB in the magnitude 
measurement. 



Third International Symposium on Space Terahertz Technology Page 309 

RESULTS AND DISCUSSION 

This section presents examples of measured data and analytical calculations. It will also 
address some conclusions drawn regarding the theoretical characterization and performance 
of the new design. Regarding the numerical aspects, the code developed to calculate the 
scattering matrix [S] at the waveguide discontinuity agreed very well with results found in 
[12]. In particular, results were compared for the reflection coefficient from asymmetric (i.e., 
single-step) E-plane and H-plane waveguide junctions. The validation of the remainder of 
the theoretical formulation and the associated software was completed by comparison with 
measured data. Part of this validation included a study of convergence as a function of 
the hole (aperture) mesh size and the number of modes used in the dyadic Green's function 
expansions. It was found that using subsections which are approximately ^f on a side, where 
\ 9 is the guide wavelength, yields a good compromise between accuracy and the requirements 
on storage and computation time. The number of modes for the Green's functions is kept 
> 600. 

The measurements performed to investigate the new design involved many variations on 
the size, shape, number, and spacing of the holes cut into the metal bar [2]. An additional 
test variable was the number of stacked mylar sheets used to form the dielectric layer. In 
many cases, the height of the backshort was such that a relatively large space was left between 
either side of the metal bar and the waveguide broadwalls. This large gap, combined with 
the variations in the mylar thickness, are used to. help understand the effect that typical 
machining tolerances will have for operation at 200-300 GHz and above. 

Results which are typical of the best performance to date are given in Figure 5b. This 
data is for a backshort with three rectangular holes, each with dimensions L\ = 19.3 mm, 
L? = 28.4 mm and spacing S = 8.7 mm. The width and height of the bar are 47.5 mm 
and 19.7 mm, respectively, leaving a gap of 1.2 mm between the bar and the waveguide 
broadwalls. The measured results in Figure 5b were obtained using a total mylar thickness 



Page 310 



Third International Symposium on Space Terahertz Technology 



CO 



O 

O 

Hi 

_l 

LL 
UJ 
GC 



CD 

;o 

Z 

O 

O 

UJ -1 

U- 
LU 



OQ 

z 

o 

O 

UJ 

_l 

LL 
HI 
X 









■1 - 



-2 

4 



(c) 



4.5 



5.5 



FREQUENCY [GHz] 



Figure 5: a) Reflected power measured from a solid bar without holes. This does not make a 
good backshort due to the several large dropouts across the frequency band, b) Reflected power 
measured from a backshort with three rectangular holes. The mylar is 0.89 mm thick. Excellent 
performance is obtained over a broad bandwidth, c) Same backshort as in (b), but mylar thickness 
has been reduced to 0.64 mm. 



Third International Symposium on Space Terahertz Technology Page 311 

of 0.89 mm. The reflection coefficient in this case is greater than —0.2 dB (0.95 reflected 
power) over a 33% bandwidth centered around 4.8 GHz. For comparison, the measured 
results for the same backshort without holes are shown in Figure 5a. This data clearly 
illustrates the improvement from the holes. The complex structure of this response, relative 
to that shown in Figure 3, is caused by asymmetrical positioning of the bar inside the 
waveguide. Other measurements made with the gap completely filled by dielectric, which 
forced a near-symmetric positioning of the bar, agreed very well with our theory and were of 
the form shown in Figure 3. The effect of reducing the mylar thickness is seen in Figure 5c, 
which gives measured data using 0.64 mm of mylar. The large dropout near 5.8 GHz has 
been shifted out of band, due to the decrease in the. effective dielectric constant. This 
response is comparable to that obtained for the conventional type of backshort shown in 
Figure 1. As expected, increasing the mylar thickness (and thus increasing the effective 
dielectric constant) moved the large dropouts lower in frequency. 

Performance similar to that with rectangular holes could be obtained using circular holes. 
Results obtained with 3 circular holes and a mylar thickness of 0.89 mm demonstrated greater 
than 95% reflected power over a 32% bandwidth centered around 4.75 GHz. This is encour- 
aging since round holes are easier to fabricate than rectangular holes for high frequencies. 

Many other variations of the backshort parameters were tested. Also, the small dips 
around 4.5 GHz in Figure 5b, and those seen in Figure 5c, are currently being investigated. 
As noted for the plain metal bar, these dips may result from asymmetrical positioning of the 
backshort inside the waveguide [13]. A more extensive discussion of the systematic parameter 
variations, measurement" observations, and comparisons with theory will be given at a later 
date. Some millimeter wave tests have also been performed and are discussed elsewhere [2]. 

In order to theoretically model the backshort performance, appropriate values were re- 
quired for an effective dielectric constant, e r , and the terminating load impedance for the 
waveguide opening (Zl in Figure 4). The problem of the dielectric constant arises because 



Page 312 Third International Symposium on Space Terahertz Technology 

the gap above and below the metal bar is only partially, and non-uniformly, filled by the 
mylar sheets. The transverse resonance technique [4] may be used to approximate e r by 
solving the exact inhomogeneous problem, and then assuming the entire guide is filled with 
some "average" material. (For the inhomogeneous case, a two-layer guide is assumed, where 
one layer is air-filled and the other is mylar-filled and of a thickness equal to the total mylar 
thickness.) A simpler approach, which yields higher values for the dielectric constant, is to 
merely compute e r based on the percentage of mylar relative to the total gap height. By 
numerical experimentation, it was found that the best approximation lies nearly midway 
between the two values. It is noted that obtaining an exact solution for a layered waveguide 
is not justified due to the unpredictable spacing of the various mylar sheets. 

The other issue was determining the correct value for the load impedance, Zl, to use 
in the calculations. Although an exact analysis is a formidable task, an approximate load 
impedance can be obtained to adequately model the waveguide discontinuity. This is done 
by first considering the exact impedance for a very thin aperture opening onto an infinite 
ground plane. For typical gap dimensions used here, this is a large, capacitive value, with 
real and imaginary parts which are both 3-5 times the dominant waveguide mode impedance. 
The extension of the backshort beyond the waveguide opening, however, will provide a better 
transition to free-space and effectively lower this impedance toward a better match. In many 
test cases, use of a normalized load impedance of Zl « 2.5 ± .5 + j\ in equation (6) 
yielded good agreement with the measured results. An important point, however, is that the 
predicted performance is nearly independent of Zl in precisely the frequency bands where 
the backshort works well. This is as expected, as most of the incident power is reflected and 
never reaches the end of the guide. It follows that changes in Zl, do modify the dropout 
regions in the return loss, as these dropouts result from out-of-band power leaking past the 
backshort. (The performance of the bar without holes is likewise strongly dependent on Zl- 
This fact can be used in determining an appropriate impedance for a given geometry, by 



Third International Symposium on Space Terahertz Technology 



Page 313 



1.0 r 






2 



0.0 



-1.0 - 



1 ■■■ r 



■ ■ i ■ • 



■ 1 1 1 1 ■ ■ 



Measured 

--•a.--. Calculated 




O A * . . ..I. ........ I...... . . . I . . . . f . ...I i . . 



4.0 4.4 4.8 5.2 5.6 6.0 

Frequency, GHz 

Figure 6: Measured data and calculated performance for the backshort with three rectangular holes. 
The mylar thickness is 0.89 mm. 

comparing measured and theoretical results for various load values.) 

A comparison between measured data and calculated performance is given in Figure 6. 
These results are for the backshort with three rectangular holes, using a mylar thickness of 
0.89 mm. Very good agreement has been obtained. The broadening of the dropouts in the 
measured data, relative to the calculated results, is believed to be due Zl and to loss in 
the measurement system which is not accounted for by the theory. The bandwidth is very 
accurately predicted, however, such that it should now be possible to design and analyze 
these backshorts for specific applications. 

CONCLUSIONS 

In summary, we have developed a theoretical analysis to predict the rf performance of 
a new non-contacting waveguide backshort. This backshort consists of a metallic bar with 
rectangular or circular holes which enhance the reflections of rf power. The simplicity of this 
design allows it to be easily scaled to millimeter wave and submillimeter wave frequencies. 
The new theoretical development is a rigorous full-wave analysis which employs a coupled 
set of space-domain integral equations and mode-matching techniques. Comparison between 



Page 314 Third International Symposium on Space Terahertz Technology 

theory and experiment on model backshorts optimized for best performance at 4-6 GHz show 
very good agreement. 

ACKNOWLED GEMENTS 

This work was supported in part by the Jet Propulsion Laboratory, California Institute 
of Technology, under contract with the National Aeronautics and Space Administration, and 
the Innovative Science and Technology Office of the Strategic Defense Initiative Organization, 
and by the University of Michigan NASA Center for Space Terahertz Technology. 

References 

[1] Collin, R. E. Foundations for Microwave Engineering, New York: McGraw-Hill, 1966, 
pp. 259-262. 

[2] McGrath, W. R. "A Novel Non-Contacting Waveguide Backshort for Millimeter and 
Submillimeter Wave Frequencies," Conference Proceedings of the Second National Tech- 
nology Transfer Conference, NASA Conference Publication 3136, Vol. 1, pp. 161-168, 
December 1991. 

[3] McGrath, W. R. "Non-contacting Waveguide Backshort," U.S. Patent pending. 

[4] Itoh, Tatsuo (editor). Numerical Techniques for Microwave and Millimeter- Wave Pas- 
sive Structures, John Wiley & Sons, 19S9. 

[5] Eleftheriades, G. V., Ali-Ahmad, W. Y., Katehi, P. B., and Rebeiz, G. M. "Millimeter- 
Wave Integrated- Horn Antennas: Part I - Theory", IEEE Trans. A. P. , vol. 39 , No. 
11, November 1991, pp. 1575-1581. 

[6] Masterman, P. H. and Clarricoats, P. J. B. "Computer field-matching solution of wave- 
guide transverse discontinuities", Proc. IEE, vol. 118 , No. 1, January 1971, pp. 51-63. 



Third International Symposium on Space Terahertz Technology Page 315 

[7] Collin, R. E. Field Theory of Guided Waves, Piscataway, NJ: IEEE Press, 1991, pp. 
78-86. 

[8] Harrington, R. F. Field Computations by Moment Methods, New York: Macmillan, 
1968. 

[9] Dib, N. I., Katehi, P. B., Ponchak, G. E., and Simons, R. N. "Theoretical and Experi- 
mental Characterization of Coplanar Waveguide Discontinuities for Filter Applications," 
IEEE Trans. MTT , vol. 39 , No. 5, May 1991, pp. 873-882. 

[10] Dib, N. I., Katehi, P. B. "Modeling of Shielded CPW Discontinuities Using the Space 
Domain Integral Equation (SDIE)," Journal of Electromagnetic Waves and Applica- 
tions, vol. 5 , No. 4/5, 1991, pp. 503-523. 

[11] Dunleavy, L. P. "Discontinuity characterization in shielded microstrip: A theoretical 
and experimental study," Ph.D. Thesis, Radiation Laboratory, University of Michigan, 
Ann Arbor, 1988. 

[12] Marcuvitz, N. Waveguide Handbook, vol. 10 of MIT Rad. Lab. Series, New York: 
McGraw-Hill, 1948. 

[13] Kerr, A. R. "An Adjustable Short-Circuit for Millimeter Waveguides," Electronics Divi- 
. sion Internal Report No. 280, National Radio Astronomy Observatory, Charlottesville, 
Virginia, July, 198S. 




Page 316 Third International Symposium on Space Terahertz Technology 

9 3 -. 2? •? 5 4 

SILICON MICROM ACHINED WAVEGUIDES FOR 
MILLIMETER AND SUBMILLIMETER WAVELENGTHS 

MarkusYap, 1 Yu-Chong Tai, 1 William R. McGrath, 2 Christopher Walker 3 



1. Department of Electrical Engineering, California Institute of Technology, 
Pasadena, CA91125 

2. Center for Space Microelectronics Technology, Jet Propulsion Laboratory, 
California Institute of Technology, Pasadena, CA 91109 

3. Department of Astronomy, University of Arizona, Tucson, AZ 85726 



Abstract ~ The majority of radio receivers, transmitters, and components operating 
at millimeter and submillimeter wavelengths utilize rectangular waveguides in some 
form. However, conventional machining techniques for waveguides operating above a 
few hundred GHz are complicated and costly. This paper reports on the development 
of silicon micromachining techniques to create silicon-based waveguide circuits 
which can operate at millimeter and submillimeter wavelengths. As a first step, 
rectangular WR-10 waveguide structures have been fabricated from (110) silicon 
wafers using micromachining techniques. The waveguide is split along the broad wall. 
Each half is formed by first etching a channel completely through a wafer. Potassium 
hydroxide is used to etch smooth mirror-like vertical walls and LPCVD silicon nitride 
is used as a masking layer. This wafer is then bonded to another flat wafer using a 
polyimide bonding technique and diced into the U-shaped half waveguides. Finally a 
gold layer is applied to the waveguide walls. Insertion loss measurements show 
losses comparable to those of standard metal waveguides. It is suggested that 
active devices and planar circuits can be integrated with the waveguides, solving the 
traditional mounting problems. Potential applications in Terahertz instrumentation 
technology are further discussed. 



Third International Symposium on Space Terahertz Technology 



Page 317 



I. Introduction 

Rectangular waveguide is a well characterized transmission medium which is 
used in a variety of complex rf components and circuits. Many sophisticated 
applications including radar, communications systems, test instruments, and 
heterodyne radiometers use waveguide components up to millimeter wave 
frequencies. The long history of development of waveguide components provides a 
broad base of knowledge to synthesize and evaluate new designs for higher 
frequencies. 

Waveguide is typically fabricated from metals such as brass and copper using 
conventional machining techniques. However, at frequencies above a few hundred 
GHz, waveguide becomes so small (less than 0.3 mm x 0.1S mm for 500 GHz - 
1000 GHz waveguide) that fabrication utilizing these conventional techniques is time 
consuming, costly and difficult In addition, mounting active and passive devices such 
as mixer diodes, filters and planar probes on these waveguides is difficult. 

A substantial research effort in recent years has been devoted to fabricating 
micromechanical structures in silicon using micromachining techniques. Moveable 
structures such as slider, gears, and spiral springs in the dimensional scale of 50-200 
|im have been fabricated [1, 2]. We have taken a new approach in developing and 
adapting silicon micromachining techniques to create silicon-based waveguide 
circuits which can operate up to millimeter and submillimeter wavelengths. 

As a first step we have started fabricating rectangular waveguides for frequencies 
between 100 GHz and 1000 GHz. Here we only emphasize WR-10 waveguides 
(operating at 75 GHz - 115 GHz) because it is compatible with our existing 
measurement equipment. Conventional WR-10 waveguide is a rectangular channel 
with inner wall dimensions of 0.1 x 0.05 inches. Our waveguide, however, is made of 
two half sections split along the broadwall as shown in Fig. 1. The reason for 
splitting the waveguide is to simplify the fabrication process and to facilitate 
integration of planar circuits and devices, which is further discussed in Section IV. 




Waveguide Channel 

Top Si Wafer 

onding Layer 
Bottom Si Wafer 



Fig. 1. a) The waveguide is split into 2 half sections, 
b) One half section of a waveguide. 



Page 318 Third International Symposium on Space Terahertz Technology 



n. Fabrication Process 

The fabrication process for the half sections with emphasis on the cross 
section is shown in Fig. 2. A thick (0.05 inches) double-side polished silicon wafer 
with (110) surface orientation is used. The major flat has a normal in the [111] 
direction within 0.4°. After a standard piranha-bath cleaning, a 1000 A Low 
Pressure Chemical Vapor Deposited (LPCVD) silicon nitride layer is deposited on 
both sides of the wafer. Photolithographic techniques are then utilized to pattern 
the waveguide etching windows. Photoresist is used as a masking layer for etching 
the silicon nitride windows with an SFg plasma. The silicon nitride is used as an 

etching mask to define b, the waveguide height, shown in Fig. lb. After removal of 
the photoresist using acetone solvent as shown in Fig. 2a, the wafer is put in a 
reflux system and etched in a water based solution of 40 % KOH at 80 °C. Figure 
2b shows the wafer after it has been etched completely through to form half of the 
waveguide. The etching rate of (110) silicon in this KOH solution is 2 jim/min and 
the etching ratio of (110):(111) planes is 170:1. At this rate 0.05 inches (1270 p.m) 
of silicon is etched thru in -11 hours. Following removal of the nitride mask using 
hot hydrophosporic acid at 150 °C, a polyimide bonding technique is used to glue 
these etched grooves to a smooth silicon wafer with an identical thickness (0.05 
inches) as shown in Fig. 2c. The wafer is then diced into pieces of half waveguides. 
Such a half waveguide is shown in Fig. 2d. Metalization is done by first depositing a 
thin (200 A) chrome layer followed by a thicker (5000 A) gold layer on the waveguide 
walls using vacuum evaporation. Further metalization is done by electroplating gold 
to a thickness of ~3 \im to reduce rf conduction losses. 



HI. Experimental Results 

In order to perform insertion loss measurement, we designed a pair of brass 
mounting blocks. The two waveguide . half-sections are put on the brass mounting 
blocks and mated together. This allows the silicon waveguide to be connected to 
microwave test equipment using conventional waveguide flanges. The silicon 
waveguides are rugged and can be firmly clamped to metallic flanges. The insertion 
loss of the WR-10 waveguide is measured over a frequency range of 75 GHz to 110 
GHz. The measurement system is shown in Fig. 3. The source is a BWO which 
produces several milliwatts. A reference sweep is first taken without the waveguide. 
This is compared to a sweep with the waveguide inserted between the source and 
detector. The insertion loss for a 2.5 cm long section of waveguide is shown in Fig. 4 
(the small wiggles in these curves are noise and do not reflect any resonances in the 
waveguide components). The measured loss is about 0.05 dB per wavelehgth (at 100 
GHz) across most of the band. This is very good performance and is comparable to 
the result for commercially available waveguide which shows a loss of about 0.024 
dB per wavelength. The small difference of 0.026 dB per wavelength is most probably 
due to differences in the quality of the gold plated surfaces. Our evaporated gold 
showed small pits which were still present in the plated layer. Also there was no 
gold on the ends of the silicon waveguide where contact was made to the metallic 
flanges of the test equipment. We expect improvements in the gold surface to be 
directly reflected in improvements in the rf losses. 



Third International Symposium on Space Terahertz Technology 



Page 329 



(110) silicon (0.05 inches) 



polyimide bonding layer 



KOH etch 




(a) 



(b) 



(c) 



silicon nitride (1000 A) 





(100) silicon (0.05 inches) 




^-chrome (250 A) 
pr gold (5000 A) 



(d) 



Fig. 2. A cross section view of the fabrication process. 



BWO 
75 GHz to 110 GHz 





\ 



WAVEGUIDE UNDER 
TEST 

\ 



DETECTOR 



2 



r~> 



10 dB 



10 dB 



SWEEPER 



SCALAR 
NETWORK 
ANALYZER 



Fig. 3. Block diagram of millimeter wave insertion loss test system. 



Page 320 



Third International Symposium on Space Terahertz Technology 




90 95 

FREQUENCY [GHz J 



110 




90 95 

FREQUENCY [GHz] 



110 



Fig. 4. (a) Measured loss of a 2.5 cm long section of Si-based WR-10 waveguide. 
The surface of the silicon was metallized with approximately 3 ujn of gold to 
reduce rf losses, (b) Measured loss of a 2.5 cm long section of conventional 
metallic waveguide. 



IV. Discussion 

Waveguide circuits are preferable at frequencies above 100 GHz since 
waveguide has the advantage of adjustable rf tuning. This solves the difficulties of 
accurately designing fixed-tuned planar microwave integrated circuits. 
Unfortunately, millimeter and submillimeter waveguide components are hard to 
manufacture by conventional machining techniques. We have shown here the 
feasibility of making silicon waveguides. It is also possible to use silicon 
micromachining techniques to fabricate other components such as: directional 
couplers, waveguide transformers, waveguide-to-planar circuit transitions, 
low-loss filters, rectangular and conical feedhorns, and dichroic plates. This wide 
variety of waveguide components will become the building blocks for complicated 
circuits. For example, complex mixer and frequency multiplier embedding circuits 
can be built. These are important for ground-based and space-based radar, 
communications, and remote-sensing applications. 



Third International Symposium on Space Terahertz Technology Page 321 



Silicon micromachined waveguide components have several important 
advantages: 1) These structures are produced by projecting the desired 
pattern onto silicon with photolithographic techniques. Therefore waveguides with 
dimensions suitable for use above 100 GHz can be easily fabricated. 2) 
Dimensional accuracy is in the order of a few microns, which is essential for the 
fabrication of high-Q components. 3) The waveguide walls would be 
atomically smooth, thereby minimizing rf losses [3]. 4) Several versions of a 
single component (with variations of a critical parameter) can be produced at the 
same time on a single wafer. This would allow for rapid optimization and reduced 
cost compared to conventional machining techniques where only one variation at a 
time is produced. 5) Most importantly, active and passive devices can be 
integrated with the waveguide. For example, a thin (~ l^im) rf transparent silicon 
nitride membrane can be fabricated across the end of the waveguide or parallel to 
its length in the E-field direction. Active devices such as Schottky diodes and SIS 
tunnel junctions as well as micromechanical rf tuning elements[l, 4] can then be 
fabricated directly on the membrane as shown schematically in Fig. 5. This would 
eliminate the long-standing problem of mounting the devices and would represent a 
significant advance for waveguide technology. 




Active Device 
1(i SiN Membrane 

, Etched 
Waveguide 



Silicon wafer with several \ ^x^LlsXVs. ^ RF Filter 

etched waveguides and 

devices 




RFin 



Fig.5 A schematic view of an integrated waveguide circuit. Several waveguide 
components can be produced on a single wafer. Active devices and planar 
circuits can be integrated directly on thin membranes spanning the 
waveguide. Micromechanical rf tuning elements can also be included in the 
waveguide. 



Page 322 Third International Symposium on Space Terahertz Technology 



Currently, we are fabricating WR-10 waveguides with SiN membranes in 
between the two half pieces of the waveguide. Metalization of the half sections of 
these waveguides will require selective plating of the silicon walls without plating 
the silicon nitride membranes. Tungsten substitution of silicon in an LPCVD 
environment [5] is proposed to meet this need. In this process, tungsten hexafluoride 
(WFg) gas attacks silicon surface and a thin layer of tungsten atoms substitute for 

silicon atoms on the surface. Further metalization can be done by electroplating the 
tungsten surface with gold. 



V . Summary 

We have demonstrated a new approach in fabricating waveguide' circuits using 
silicon micromachining technology. In particular, we have fabricated a 100 GHz silicon 
rectangular waveguide. The insertion loss of 0.05 dB/X is comparable to a 
commercially available metal waveguide. As we improve our plated gold quality, we 
expect to improve the insertion loss. We have also proposed a new approach of 
integrating active/passive devices and micromechanical rf tuning elements with 
waveguide. 



This work is supported in part by California Institute of Technology President's Fund 
under grant PF-347 and the Jet Propulsion Laboratory, California Institute of 
Technology, under contract with the National Aeronautics and Space Administration 
and the Innovative Science and Technology Office of the Strategic Defense Initiative 
Organization. 



Third International Symposium on Space Terahertz Technology Page 323 

References 



[1]. L.-S. Fan, Y.-C. Tai, and R. S. Muller, "Integrated Movable MicromechanicaJ 
Structures for Sensors and Actuators," IEEE Trans, on Electron Devices, vol. 
35, pp. 724-730, June 1988. 

[2]. M. Mehregany, K. J. Gabriel, and W. S. N. Trimmer, "Integrated Fabrication of 
Polysilicon Mechanisms," IEEE Trans, on Electron Devices, vol. 35, pp. 
719-730, June 1988. 

[3]. F. J. Tischer, "Experimental Attenuation of Rectangular Waveguides at 
Millimeter Wavelengths," IEEE Trans. Microwave Theory and Tech., vol. 
MTT-27, pp. 31-37, January 1979. 

[4]. V. M. Lubecke, W. R. McGrath, and D. B. Rutledge, " Sliding Backshorts for 
Planar Circuits," International Journal of Infrared and Millimeter Waves, vol. 
12, pp. 1387-1397, December 1991. 

[5]. N. Kobayashi, M. Suzuki, and M. Saitou, "Tungsten Plug Technology Using 
Substitution of W for Si," IEEE Trans, on Electron Devices, vol. 37, pp. 
577-582, March 1990. 



Page 324 Third International Symposium on Space Terahertz Technology 

This is intended as a review paper only and it summarizes work which has been submitted for 

publication in the IEEE Transactions on MTT . 

-Z&^Z^ N93-27755 

\\ PROGRESS IN INTEGRATED- CIRCUIT HORN 

ANTENNAS FOR RECEIVER APPLICATIONS 

PART 1: Antenna Design 

George V. Eleftheriades, Walid Y. Ali- Ahmad, and Gabriel M. Rebeiz 

NASA/Center for Space Terahertz Technology 

Electrical Engineering and Computer Science Department 

University of Michigan 

Ann Arbor, MI 48109-2122 



ABSTRACT 

The purpose of this work is to present a systematic method for the design of multimode quasi- 
integrated horn antennas. The design methodology is based on the Gaussian beam approach and 
the structures are optimized for achieving maximum fundamental Gaussian coupling efficiency. 
For this purpose, a hybrid technique is employed in which the integrated part of the antennas 
is treated using full-wave analysis, whereas the machined part is treated using an approximate 
method. This results in a simple and efficient design process. The developed design procedure 
has been applied for the design of a 20dB, a 23dB and a 25dB quasi-integrated horn antennas, 
all with a Gaussian coupling efficieny exceeding 97%. The designed antennas have been tested 
and characterized using both full-wave analysis and 90GHz/370GHz measurements. 



Third International Symposium on Space Terahertz Technology Page 325 

I. QUASI-INTEGRATED HORN ANTENNA DESIGN : INTRODUCTION 

The integrated-circuit horn antenna was introduced in [1] and analyzed using a full-wave analysis 
technique in [2]. It consists of a dipole (or monopole) feed evaporated on a thin dielectric 
membrane which is suspended in a pyramidal cavity etched in silicon or GaAs. Recently, this 
antenna has been used in several millimeter and submillimeter-wave applications including a 
double-polarized antenna design at 93GHz [4], a 256 element imaging array at 802GHz [5], and 
a monopulse tracking system at 94GHz [6]. However, the wide flare-angle of the integrated- 
circuit horn antenna, which is dictated by the anisotropic etching involved in its fabrication 
(70° in silicon), limits its useful aperture size to 1.6A and its gain to 13dB. To this end the 
quasi-integrated horn antenna was introduced [3], which consists of a machined small flare- 
angle pyramidal section attached to the integrated portion (fig.l). The resulting structure is a 
simple multimode pyramidal horn with circularly symmetric patterns, high gain, and low cross- 
polarization, which is particularly attractive for submillimeter quasi-optical receiver applications. 
The minimum machined dimension involved in its fabrication is around 1.5 A which enables its 
fabrication to frequencies up to 2THz. The purpose of this paper is to describe a systematic 
approach towards the design of these horn antennas, and to provide a full range of practical 
quasi-integrated horn antenna designs along with their detailed radiation characteristics. Since a 
very desirable property of antennas intended for use in quasi-optical systems is the high Gaussian 
content of their radiated fields [7], the developed design methodology is based on the optimization 
of the quasi-integrated horns for achieving maximum fundamental Gaussian coupling efficiency. 
The Gaussian coupling efficiency is particularly important in quasi-optical receiver applications 
because it directly influences the total system performance with a pronounced effect on the 
receiver noise temperature [8]. 

II. MULTIMODE APERTURE ANALYSIS FOR MAXIMUM FUNDAMENTAL 

COUPLING EFFICIENCY 



Page 326 



Third International Symposium on Space Terahertz Technology 




Integrated on Si section 



Machined gain and phasing section 



Fig.l The general configuration of the quasi-integrated multimode horn antenna. 





1.1 




1.0 




0.9 


{=■ 


0.8 


o 


0.7 


c 




.2- 


0.6 


t£ 




4) 


0.3 


00 

c 


0.4 










3 


0.3 


O 




U 


0.2 




0.1 



0.0 



iitijiiffftrtiittiiiiifft r't i 'i i i t t r i i i t t I i 



TTTTTTT 




0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 



w^a 



Fig. 2 The maximum Gaussian coupling efficiency as a function of the 
w / a ratio for various aperture modes available for beamshaping (up 
to TE m ,n/TM m ,n , m = 1, 3 . . . M, n = 0, 2 . . . N) . 



Third International Symposium on Space Terahertz Technology Page 327 

Consider a square aperture of side a in a ground-plane which is radiating in the half-space 
z > 0. The transverse electric field of the aperture at z = can be expanded in terms of the 
eigenfunctions of a square waveguide of the same side a : 

M,N 

Et„(x,y)= £{A mn e™(x,y) + C mn e™(x,y)} . (1) 

m,n 

In (1) it is assumed that only modes with indices (m = 1, 3, 5 ... A/ and n = 0, 2, 4, 6 ... N) are 
present as is the case of a pyramidal horn which is either fed by a centered Hertzian dipole or by 
a waveguide which supports only the dominant TEio mode [2]. We now proceed to determine the 
modal coefficients A mn ,B mn so that the coupling between the aperture field and a fundamental 
beam is maximized. If the copolarized and cross-polarized components of the aperture field are 
defined to be the E x , ap and the E yap components respectively, then the transverse electric field 
can be rewritten in the form : 

E y ,a V {x, y) = £ <£„¥£„(*, y) , £ r , op (*, y) = £ «#„*;*(*, y) (2) 

171,71 171,71 

where the orthonormalized copolarized and cross-polarized hybrid modes ^f^m ^mn are: 

▼ /.« / v V^^n , v m + n — 1 TY17T X > , TT7T1J . ,, *— i i .. *«. ,«.* 

*mn(^y) = ^—^(-l ^- cos( )cos(— £ , \x\ < a/2, \y\ < a/2 (3) 

a a a 

**" n (x, y) = 2L_i _i)-V^ S in ) sin(— y -) , |x| < a/2, |y| < a/2 (4) 

a a a 

In (3) and (4) the origin of the Cartesian coordinates is located at the geometrical center of 
the aperture and e n = 2 — S n0 is the Neumann number. The corresponding copolarized and 
cross-polarized modal coefficients of (3-4) are related to the modal coefficients of (1) through: 

,co _ n ^mn ~ rnAmn p _ TlA mn -\- TnL> mn 

Vm 2 + n 2 V"i + n l 

Now the coupling efficiency r](w ) of the aperture to a fundamental Gaussian beam of waist 
radius w , which has its waist on the aperture is given by [11] : 



r)(w ) = 



M,N 



£ d- n I mn (w ) 



m,n 



2 2 M,N 

/(^E(fcr+K p ni 2 )) (6) 



Page 328 



Third International Symposium on Space Terahertz Technology 
here, I mn (w ) = II ^ n (x,y)exp(-{x 2 + y 2 )/w 2 ) dx dy. 

J Japert. 



(7) 



We wish at this point to determine the modal coefficients <f" n and d% n so that the coupling 
efficiency r](w ) is maximized. For this purpose, the application of Schwarz's inequality to (6) 
immediately implies that the maximum coupling efficiency r) max (w ) occurs in the absence of 
cross-polarization and is obtained from : 



M,N 



with the corresponding co-polarization modal coefficients determined by 

d mn I Imn(Wo) = Constant. 



(8) 



(9) 



The condition for vanishing cross-polarization is (see 5) : nA mn = —rnC m n ■ 



(10) 



Therefore, for maximum fundamental Gaussian coupling efficiency the aperture modes should 
add in phase and their relative magnitudes should satisfy conditions (9) and (10). The maximum 
coupling efficiency r) max (w ) of equation (9) still depends on the waist radius w and it is shown in 
figure 2 as a function of the ratio w /a for various indices (M,N). In table 1 we show the relative 
magnitudes between the modes at the optimum w , opt /a ratio, for some practically encountered 
aperture sets of modes. 



Available modes (M,N) 


(1,0) 


(1,2) 


(1,2)+TE 30 


(3,2) 


w ,o P t/a 


0.43 


0.34 


0.32 


0.29 


cpl. efficiency : r) max 


84% 


98.5% 


99.2% 


99.7% 


^12/^10 


- 


0.51 


0.56 


0.64 


dio/diQ 


- 


- 


0.11 


0.17 


df 2 /d\Q 


- 


- 


- 


-0.11 



Table 1: Optimum parameters for maximum fundamental Gaussian coupling efficiency for certain 
practically encountered aperture modes available for beamshaping. 



Third International Symposium on Space Terahertz Technology Page 329 

III. APPROXIMATE ANALYSIS OF THE MACHINED SECTION AND 
DESCRIPTION OF THE DESIGN PROCESS 

Consider the gradually-flared pyramidal machined section of axial length Lm and of half flare- 
angle O (see fig. 1) which is assumed excited at its throat by M x N locally propagating modes. 
Since the machined section is gradually flared and the incident modes propagating, reflections at 
the throat are considered negligible and the corresponding transverse electric field is given by : 

E t M*,v) = £ Ml^ £ (*, v) + Ce M (x,y)} • (ID 

To a first order approximation we can assume that each mode preserves its carried power upon 
propagating from the throat to the aperture. Also, each mode acquires a phase shift computed 
by: 

*m» = I " mn {z)dz (12) 

JO 

where (3 mn (z) is the local propagation constant of the mn'^-mode. The above phase shift has 
been used extensively for the design of multimode horns [9-10] and it can be rigorously justified 
through a coupled-mode analysis of gradually flared tapers [13]. The aperture field is assumed to 
be modulated by a quadratic phase factor Ql t (x, y) of curvature Lt = a/(2 tan O ) with Lt being 
the total virtual length of the taper. Under the above assumptions and neglecting reflections , 
the aperture field is simply given by : 

M,N 

E t , ap (x,y) = Q LT (x,y)E{K P n ^ TE (^y) + C a m P n e a rn P n TM (^y)} (13) 

with the quadratically modulated aperture modal coefficients related to the throat modal coef- 
ficients through : 



4?n = A%jY£ TE /Y eM-J*mn) , <fr B = CJ* Jy£™ /Y exp(-;$ mn ) (14) 

where Y^ n is the throat admittance for the mn th mode and Y is the free-space intrinsic admit- 
tance which has been assigned to the aperture modes. Based on the above simplified analysis for 



Page 330 Third International Symposium on Space Terahertz Technology 

the machined section and on a full-wave analysis of the integrated portion a three-stage design 
process has been established and is summarized below : 

1. The integrated 70° flare-angle section of the antenna structure of figure 1 (including the 
step discontinuity) is selected and analyzed independently of the machined section. For 
this purpose, the dipole-fed integrated portion is assumed to be terminated by an infinite 
square waveguide of side (a 3 + 2s) and is analyzed using the full-wave analysis technique 
of [2] to obtain the throat modal coefficients A^ n , Cj£ n . The junction cross-section a s and 
the step size s (see fig. 1) are selected so that the magnitudes of the radiating aperture 
modal coefficients, as predicted by equations 5 and 14, satisfy the optimal conditions (9) 
and (10) as closely as possible. 

2. The infinite waveguide is now replaced by the gradually flared machined section and the 
assumption is made that the modal coefficients at the throat of the machined section retain 
their computed values of stage 1. This is a good approximation since the actual excited 
modal coefficients are determined by the difference between the integrated portion flare- 
angle and the machined section flare-angle and this difference is always dominated by the 
large 70° flare-angle of the integrated portion [10]. The length Lm and the flare- angle 8 of 
the machined section are then selected iteratively (using 12) so that the modal coefficients 
dmn appear in phase on the radiating aperture. The shortest possible length is chosen in 
order to achieve the maximum bandwidth. 

3. Finally, the length and the flare-angle of the machined section are "fine-tuned" using the 
full-wave analysis of [2] for the entire quasi-integrated horn antenna and again for achieving 
maximum Gaussian coupling efficiency. 

In table 2 we quantify several practical geometries of integrated portions which have resulted 
from the first stage of the design process. 



Third International Symposium on Space Terahertz Technology Page 331 

IV. NUMERICAL AND EXPERIMENTAL RESULTS FOR SPECIFIC 
QUASI-INTEGRATED HORN ANTENNA DESIGNS. 

The algorithm of section III has been employed for the design of a 20dB, a 23dB and a 25dB 
quasi-integrated horn antenna, all with a fundamental Gaussian coupling efficiency exceeding 
97% and with a full-null beam efficiency around 99%. Although, in the design process the 
analysis of the machined section is performed using the approximate method of section III, the 
computation of the radiation characteristics of the finally designed horns is carried out using the 
full-wave analysis technique of [2]. Furthermore, using this full-wave analysis along with 6GHz 
scale-model measurements it was verified that the input impedance of the feeding strip-dipole in 
the integrated portion of the horn is not affected by the attachment of the machined section [3]. 
This is due to the fact that the input impedance of the feeding strip is mainly determined by 
its local geometrical environment which remains unaffected by the attachment of the machined 
section. The input impedance for the integrated-circuit horn antennas has already been analyzed 
theoretically and characterized experimentally in [2] where it was shown that by adjusting the 
dipole position inside the horn, the input impedance can be matched to either Schottky or SIS 
diodes. Therefore, the results of [2] are directly applicable to the case of the quasi-integrated 
horn antennas as well. 

A. 20dB quasi-integrated horn antenna. 

The geometrical parameters for the 20dB realization are calculated to be (a, = 1.35A,s = 
0.0, Lm = 7A, 9 = 9°, dp = 0.39A) and the numerically computed patterns from the third stage 
of the design process along with the corresponding 90GHz measurements have been reported 
in [3]. In fig. 3 the principal patterns are compared to the patterns obtained by analyzing 
the machined section using the approximate method of section III. As shown, the approximate 
model agrees well with both the full-wave analysis and the measurements thus verifying the 
approximations used in the design process. The main radiation characteristics of this horn at 



Page 332 Third International Symposium on Space Terahertz Technology 

the design frequency and at the edges of the ±5% bandwidth are summarized in table 3. The 
indicated 10-dB beamwidth fluctuation corresponds to the variation of the beamwidth in an 
azimuthal far-field cut. The Gaussian-beam rolloff was calculated at the edges of the ±5% 
bandwidth using the Gaussian-beam parameters which were calculated at the design frequency 
f . The calculated phase center was found to be located at a distance of 1.5A from the horn 
aperture for the E-plane and at 1.4A for the H-plane. 

B. 23dB quasi-integrated horn antenna. 

The optimized design parameters for a 23dB quasi-integrated horn are found to be (a, = 
1.52A.5 = 0.17A, Lm = 13A,0 O = 8.5°, dp = 0.39A) and the computed principal patterns from 
both the full- wave analysis of the entire antenna and from the approximate model of section III 
are compared in figure 4 to corresponding 370GHz measurements. In figure 5 we include also 
the computed from the full-wave analysis and the measured patterns for the 45°-plane. The 
radiation characteristics of this horn are being summarized in table 4. For the 23dB horn the 
phase center was calculated to be at 3.7A inside the horn for the E-plane and at 3.5A for the 
H-plane. 

C. 25dB Quasi-integrated horn antenna. 

In order to evaluate the efficiency of the design process and to provide a full range of practical 
designs, a 25dB quasi-integrated horn has also been designed and the computed geometrical 
parameters are found to be: (a, = 1.52A,3 = 0.0A, Lm = 19.5A,0 O = 10°, dp = 0.39A). The 
radiation patterns, as calculated from the full-wave analysis and shown in figure 6 still exhibit 
excellent circular symmetry, low cross-polarization and suppressed sidelobes. The location of the ' 
phase center for this horn was computed to be at a distance of 13A from the aperture for the 
E-plane and at 11 A for the H-plane. The rest of the main radiation characteristics of this horn 
antenna are being tabulated in table 5. 



Third International Symposium on Space Terahertz Technology 



Page 333 





Optimum 


a s = 1.35A 

5 = 0.0 


a s = 1.52A 

5 = 0.0 


a s = 1.35A 
s = 0.17A 


a s = 1.57A 
s = 0.0 


Mi°2 ap l/Ko p l 


0.56* 


0.52 


0.50 


0.55 


0.51 


i«i/ra 


0.114 


- 


0.11 


0.117 


0.146 


arg^/A'S) 


180° 


200° 


183° 


182° 


179° 


\C%\/\A a & 


2 


4.5 


4.4 


5.1 


4.3 



Table 2: Comparison between the optimum aperture modal coefficients and the modal coefficients 
launched at the aperture by four practical integrated portion sections. The exciting dipole is 
positioned at a distance of 0.39A from the apex of the horn. * The optimum ratio |^n' ap |/l^iol 
is 0.51 for the a„ = 1.35A geometry which only triggers the TEi ,TEi 2 /TMi 2 modes. 





0.95/ o 


/• 


1.05/o 


Gain 


19.4dB 


20dB 


20.6dB 


Aperture efficiency 


60.6% 


62.8% 


65.4% 


lOdB Beamwidth 


37° ± 1° 


34° ±1.2° 


32° ±1.8° 


Sidelobe-level (E-plane) 


-23dB 


-27dB 


-26.3dB 


Cross-pol.(45°) 


-22.5dB 


-22.7dB 


-23dB 


Beam-efficiency (to -lOdB) 


85% 


86% 


86.5% 


Gaussian Coupling 


96.4% 


97.3% 


96.9% 


Gaussian Coupling rolloff 


95.5% 


97.3% 


96.5% 



Table 3: The main radiation characteristics of the 20dB quasi-integrated horn antenna (see text). 





0.965/ o 


/• 


1.035/„ 


Gain 


22.2dB 


22.8dB 


23.6dB 


Aperture efficiency 


48.5% 


52% 


58.4% 


lOdB Beamwidth 


27.6 ±0.2° 


25° ±1.1° 


22.5° ±1.3° 


Sidelobe-level (E-plane) 


-28dB 


-33dB 


-29.8dB 


Cross-pol.(45°) 


-20.5dB 


-21dB 


-22dB 


Beam-efficiency (to -lOdB) 


86.6% 


86% 


86.6% 


Gaussian Coupling 


97.2% 


97.3% 


96.8% 


Gaussian Coupling rolloff 


96.3% 


97.3% 


96.0% 



Table 4: The main radiation characteristics of the 23dB quasi-integrated horn antenna (see text). 



Page 334 



Third International Symposium on Space Terahertz Technology 









-5 




-10 


ffl 




T3 


-15 


.s 




cd 




04) 


-20 


<U 




> 




•a 




«j 


-?,5 


<D 




od 





-30 
-35 
-40 



vnm i ii I iiht i ii i m i n i ii 



l 



FULL-WAVE 
90GHz-MEAS. 
APPR. MODL. 



'■■'■' 



-90 



-60 



-30 



T-rT-m-r- 



urii r-r r r 








30 



60 



Elavation angle (deg) 



90 



Fig.3 The E (right) and H-plane (left) patterns of the 20-dB quasi- 
integrated horn. The 90GHz measured patterns are compared to the 
full-wave analysis and the approximate analysis patterns. Detailed pat- 
terns including cross-polarization are shown in [3]. 






o 
u 






« 

a 
o 

S 

3 

in 
o 
a. 

5 



•2 



T3 



00 

V 

> 

■a 

OS 



-5 - 



■10 



•15 



-20 - 



-25 



-30 



-35 





'"• ' 7 


^ ' ' 


. 




\ 


' 


H-PLANE f 


\ E-PLANE 


. 




I 




T- II « - 1 / 




■ 


a 370GHzmeasur. P 


l 




Approx. model / 


| 


- 




\ - 


- 


P 


1 - 


. 




I 


*~ 




j 


- 




I 


? 

1 

1 




1 n'll 




;A,.fl.q.^i 



-90 



-60 



-30 



30 



60 



Elavation angle (deg) 



90 



Fig.4 The E (right) and H-plane (left) patterns of the 
23-dB quasi-integrated horn. The 370GHz measured pat- 
terns are compared to the full-wave analysis and the ap- 
proximate analysis patterns. 






PQ 



CO 
fcfl 

cu 

> 



cu 



-5 



10 



■15 



E-plane(exp.) 

- H-j>lane(exp.) 

1_ '_*_«_*_" 45 -plancfexp. ; i 



2 -20 



-25 



-30 



-35 



itfi i i i 1 1 1 m i rr 1 1 1 1 1 1 1 1 ii 1 1 i /m 1 1 ii 1 1 1 1 1 1 1 i 1 1 1 1 1 1 1 1 1 1 1 1 m 



1 1 1 1 1 1 i 1 1 1 Ai i i/i i 



I 




_ 45°-plane(exp.). 
_ 45°-plane(th.) 



j 1 1 1 i/i 1 1 1 1 1 1 1 1 1 1 1 1 1 1 ii i 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 



-90 -60 -30 30 60 90 

Elevation angle (degrees) 



Fig.5 The measured at 370GHz E/H and 45°-plane pat- 
terns vs. the full- wave patterns of the 23-dB quasi- 
integrated horn. 



Page 336 



Third International Symposium on Space Terahertz Technology 





0.965/ o 


fo 


1.035/ o 


Gain 


24.7dB 


25.5dB 


26.2dB 


Aperture efficiency 


36% 


40% 


44% 


lOdB Beamwidth 


21.6 ±0.8° 


19.2° ±0.7° 


17.5° ±0.5° 


Sidelobe-level (E-plane) 


-28.7dB 


-30.8dB 


-30.SdB 


Cross-pol.(45°) 


-22.6dB 


-24dB 


-24.7dB 


Beam-efficiency (to -lOdB) 


84.5% 


85% 


85% 


Gaussian Coupling 


97.1% 


97.5% 


97.4% 


Gaussian Coupling rolloff 


96.5% 


97.5% 


97.1% 



Table 5: The main radiation characteristics of the 25dB quasi-integrated horn antenna (see text). 



CQ 

c 
'« 

4) 

> 










I T T T' f tllfffrT 1 ^! lltfTfllllll II I JW I 



-10 



-15 



^ -20 



-25 - 



-30 



-35 



«*«**» ■ ■ * I ■ ■ 'i 'i ■ '■ ' • ' ■ ' ' ■ '* • • • ' * ■ 




-90 



-60 



-30 







30 



60 



90 



Elevation angle (degrees) 



Figure 6: The calculated from the full- wave analysis patterns of the 25-dB quasi-integrated horn. 



Third International Symposium on Space Terahertz Technology 



Page 337 



References 



[1] G.M Rebeiz, D.P. Kasilingam, P.A. Stimson, Y. Guo and D.B. Rutledge, "Monolithic 
millimeter- wave two-dimensional horn imaging arrays," IEEE Trans. Antennas Propagat.. 
vol. AP-28, pp. 1473-1482, Sept 1990. 

[2] G.V. Eleftheriades, W.Y. Ali-Ahmad, L.P.B. Katehi, and G.M. Rebeiz, "Millimeter-wave 
integrated-horn antennas Part I-Theory, and Part II- Experiment," IEEE Trans. Antennas 
Propagat, vol. AP-39, pp. 1575-1586, Nov. 1991. 

[3] G.V. Eleftheriades, W.Y. Ali-Ahmad, and G.M. Rebeiz, "A 20-dB Quasi-integrated horn 
antenna," IEEE Microwave and Guided Wave Letters, vol. 2, pp. 73-75, Feb. 1992. 

[4] W.Y. Ali-Ahmad and G.M. Rebeiz, "92GHz dual-polarized integrated horn antennas," IEEE 
Trans. Antennas Propagat., vol. AP-39, pp. 820-825, June 1991. 

[5] W.Y. Ali-Ahmad, Gabriel M. Rebeiz, Hermant Dave, and Gordon Chin "802 GHz Integrated 
horn antennas imaging array," International Journal of Infrared and Millimeter Waves, vol. 
12, No. 5.1991. 

[6] C.C Ling and G.M Rebeiz, "94GHz Integrated monopulse antenna," IEEE AP-S Interna- 
tional Symposium , Ontario, Canada, June 1991. 

[7] P.F. Goldsmith, "Quasi-optical techniques at millimeter and submillimeter wavelengths," in 
Infrared and Millimeter Waves, vol. 6, New York : Academic, 1982, pp. 277-243. 

[8] E. N. Grossman, "The coupling of submillimeter corner-cube antennas to Gaussian beams," 
Infrared Phys., vol. 29, pp. 875-885, 1989. 

[9] P.D. Potter, "A new horn antenna with suppressed sidelobes and equal beamwidths," Mi- 
crowave J., vol. VI, pp. 71-78, June 1963. 

[10] S.B. Cohn, "Flare-angle changes in a horn as a means of pattern control," Microwave Jour- 
nal., vol. 13, pp. 41-46, Oct. 1970. 

[11] G.V. Eleftheriades, and G.M. Rebeiz, "High-gain step-profiled integrated diagonal horn- 
antennas," To appear in IEEE Trans. Microwave Theory Tech., mini special issue on Space 
Terahertz Technology, May 1992. 

[12] C. E. Profera, "Complex radiation patterns of dual mode pyramidal horns," IEEE Trans. 
Antennas Propagat, vol. AP-25, pp. 436-438, May 1977. 

[13] L. Solymar, "Spurious mode generation in nonuniform waveguide," IRE Trans. Microwave 
Theory Tech., vol. MTT-7, pp. 379-383, 1959. 



Page 338 Third International Symposium on Space Terahertz Technology 

N93-2775 6 




/ 



PROGRESS IN INTEGRATED- CIRCUIT HORN 
ANTENNAS FOR RECEIVER APPLICATIONS 

Part II: A 90 GHz Quasi-Integrated Horn Antenna Receiver 



Walid Y. Ali- Ahmad, George V. Eleftheriades and Gabriel M. Rebeiz 

NASA/Center for Space Terahertz Technology 
Electrical Engineering and Computer Science Department 

University of Michigan 
Ann Arbor, MI 48109-2122 



ABSTRACT 

A receiver belonging to the family of integrated planar receivers has been developed at 
90 GHz. It consists of a planar Schottky-diode placed at the feed of a dipole-probe suspended 
inside an integrated horn antenna. The measured planar mixer single-sideband conversion 
loss at 91.2 GHz (LO) with a 200 MHz IF frequency is S.3dB±0.3dB. The low cost of 
fabrication and simplicity of this design makes it ideal for millimeter and submillimeter- 
wave receivers. 



\ 



Third International Symposium on Space Terahertz Technology Page 339 



INTRODUCTION 

Fundamental mixers are currently the front-ends components for all millimeter- wave receivers 
above 100 GHz. The mixers use a Schottky-diode suspended in a machined waveguide with 
an appropriate RF matching network. These components are expensive to manufacture 
especially above 200 GHz where waveguide tolerances become severe. A low noise planar 
receiver consisting of a planar Schottky diode integrated with an efficient planar antenna 
is a needed alternative at millimeter- wave frequencies. Recent advances in planar Schottky 
diodes resulted in excellence performance at 94 GHz with measured diode temperatures 
competitive with whisker-contacted diodes [1]. In this work, a planar diode is combined 
with an integrated horn antenna [2,3] to yield a 90 GHz receiver. The antenna feed-dipole 
impedance can be designed to conjugate match the RF diode impedance [4]. This eliminates 
the need for an RF matching network and thereby simplifies the mixer design. A machined 
section is attached to the front of the integrated horn antenna to yield a multi-mode horn [5]. 
The planar configuration results in an inexpensive quasi-monolithic receiver with an expected 
performance as good as the best waveguide receiver at 100 GHz. 

MIXER DESIGN AND THEORETICAL PERFORMANCE 

The length of the feed-dipole and its position inside the integrated horn antenna are designed 
so that its impedance conjugate matches the RF diode impedance [4]. As a result, the planar 
diode is epoxied right at the dipole apex. An RF choke is obtained by using two integrated 
lumped capacitors on a coplanar stripline. The first capacitor is A /4 away from the dipole 
feed and the second capacitor is A^/2 away from the first one. These capacitors introduce an 
RF open circuit at the dipole feed and let the IF signal pass through the coplanar stripline 
(Fig.l). The circuit is integrated on highly resistive Silicon in order to minimize any losses of 
the IF signal on the surrounding dielectric substrate. A microstrip quarter-wave transformer 
over a Duroid 5870 substrate [7] is used to match the 1.4 GHz IF diode output impedance 
to 50J1. Fig. 2 shows the structure of the integrated horn antenna receiver. The machined 
section, not shown in this figure, is attached to the front aperture of the horn antenna. Gold 



Page 340 



Third International Symposium on Space Terahertz Technology 



is evaporated on all the horn walls except on the membrane wafer walls, in order not to 
short-out the feed lines. The diode of choice to be used in this design is the UVa SC2R4 
planar Schottky diode with 2.5/zm anode diameter, a 5-6fF zero- bias junction capacitance, a 
12-13fF parasitic capacitance and a 5-6fi series resistance. A microwave model of the horn 
receiver structure shown in Fig. 2 was built at 2.55 GHz in order to find the right feed-dipole 
impedance to conjugate match the UVa diode RF impedance. A feed-dipole, which is 0.392A 
long and positioned 0.38A from the apex of the horn, has an input impedance of 75+J55 
fi with the membrane walls uncoated and with no diode chip modeled at the dipole feeds. 
The input impedance dropped to 70+jlO fi due to the capacitive effect of the diode block 
when it was modeled. A LldB-1.3dB power loss was found in the microwave model by 
measuring the difference in powers detected by the feed- dipole for the case of coated and 
uncoated membrane walls respectively. Table I shows the mixer theoretical performance for 
the UVa diode at 91.2GHz(LO) and 91.4GHz(RF) for a bias of 0.65V and an available LO 
power of 2dBm. The analysis was done using the reflection algorithm [6]. the variation in 
conversion loss over 10% bandwidth is due to the variation in the feed dipole impedance. 



Table I 



f /F (GHz) 


0.2 


fftF(GHz) 


91.4 


Zd,p /e,fiF(^) 


70+jlO 


Zd,po/e,2HF(^) 


14+jlO 


^diode.RFK^) 


62-J19 


^diodcLoify 


55-J49 


% diode, IF (fy 


86 


Diode SC2R4 SSB Conversion loss(dB) 


5.7 


Diode SSB Conversion loss(dB) over 10% BW 


5.7-6.2 



Third International Symposium on Space Terahertz Technology 



Page 341 




to IF 
Amplifier 



IF Matching network 



Figure 1: The mixer design consisting of the diode epoxied at the dipole feeds, the two 
lumped capacitors forming the RF choke, and the microstrip line IF matching network. 



35X 



Front 
wafers 




IF Matching Network 
(microstrip) 



Back 
wafers 



&$%&>*'&. 



MembraiiL i ' v '^* , *v ! ',& r v 



11 u ' yy.wgw ' « 



770nm 

\ 



Figure 2: The integrated horn antenna receiver structure. The horn walls of the membrane 
wafer are not coated with gold. 



Page 342 



Third International Symposium on Space Terahertz Technology 



RECEIVER MEASUREMENTS 



A quasi-integrated horn antenna receiver was built at 91.4 GHz with a UVa SC2R4 diode 
epoxied at the dipole feeds. Video detection measurements were done at 91.4 GHz by shining 
a known plane wave power density onto the multi-mode antenna and measuring the output 
detected diode voltage using a lock-in amplifier. The diode theoretical video responsivity 
vs. bias current is fitted to the measured data by using the parameters shown in table II to 
model the receiver. 

Table II 



^aperture 


e lossinwalls 


^dipole 


R* 


C J 


<^bi 


c P 


7? 


-2.0dB 


-1.2dB 


70+jlO ft 


6ft 


5.5 fF 


0.88 V 


12.5 fF 


1.14 



In fig. 3, the measured video responsivity is equal to the ratio of the detected voltage across 
the diode over the plane wave power incident on the aperture of the quasi-integrated horn 
antenna. The diode parameters used in the model are those provided by University of 
Virginia. Although the receiver was designed for a 1.4 GHz IF frequency, we found that 
epoxy and solder at the junction between the duroid and the silicon substrate have added 
a parasitic IF capacitance. The measurements were therefore done at 200 MHz where this 
capacitance has negligeable effect. For the SSB conversion loss measurement, a calibrated 
91.4 GHz RF plane wave and a 91.2 GHz LO were combined using a thin Mylar sheet and 
shined on the receiver. Figure 4 shows the measured planar mixer SSB conversion loss, 
defined as the received IF power divided by the RF power absorbed by the horn aperture 
(plane wave power density x horn area x horn aperture efficiency). The SSB conversion 
loss includes the 1.2dB loss in the uncoated membrane walls. An S.3dB SSB conversion 
loss is measured at 91.4 GHz with 3.5dBm estimated LO power available at the feed-dipole 
terminals. The coupling efficiency of the horn aperture to a plane wave is normalized out of 
the measurement because in a receiver system the horn has a gaussian coupling efficiency of 
97%. Also, the measured result can be directly compared to waveguide mixers performance 
which have no antennas attached. The 8.3 dB SSB conversion loss compares favorably with 
the best waveguide mixers performance (5.3±0.5dB) using the same diode [1]. 



Third International Symposium on Space Terahertz Technology 



Page 343 




10" 10"° 10" 10 -* 10 
Bias Current, Amps 



Figure 3: Measured and theoretical video responsivity at 91.4GHz. 



25 



*& 20 



(n 
O 



15 



o 



£ io 

c 

o 

PQ 5 
in 

CO 



i i i i i i i i i i i i i i i i i i i i i i i i i i i i | i i i i 



-^ SSB (Exp.) 



Q I I I I l I l l l I I I I I I I I l l l I I I I I I l i i i I i i i l 

-25 -20 -15 -10 -5 5 10 

Estimated LO Power Available 
at Feed-Dipole Terminals (dBm) 



Figure 4: Measured planar mixer SSB conversion loss for the SC2R4 diode at 91.2 GHz 
(LO). The measured values include a 1.2dB loss attributed to power loss in the horn walls. 



Page 344 • Third, International Symposium on Space Terahertz Technology 



CONCLUSION 

A 90GHz quasi-integrated horn antenna receiver has been designed and tested. The measure- 
ments show that this new receiver is a very good candidate for millimeter-wave applications. 
DSB measurements are being done on a new improved receiver design and using the UVa 
SC2T3 diode which has lower parasitic capacitance and series resistance than the UVa SC2R4 
diode. 

ACKNOWLEDGMENTS 

This work was supported by the NASA/Center for Space Terahertz Technology at the Univer- 
sity of Michigan, Ann Arbor. We thank G.V. Eleftheriades for the multi-mode horn design. 
We thank Dr. Thomas W. Crowe and William L. Bishop at the University of Virginia, for 
providing us with the diodes. 

REFERENCES 

[1] D.G. Garfield, R.J. Mattauch, and S. Weinreb,"RF Performance of a Novel Planar 
Millimeter- Wave Diode Incorporating an Etched Surface Channel," Trans Microwave 
Theory Tech., vol MTT-39, pp. 1-5, Jan 1991. 

[2] G.M. Rebeiz, D.P. Kasilingan, P.A. Stimson, Y. Guo, and D.B. Rutledge, "Monolithic 
millimeter-wave two-dimensional horn imaging arrays," IEEE Trans. Antennas Propag., 
vol. AP-28, Sept. 1990. 

[3] W.Y. Ali-Ahmad, and G.M. Rebeiz, "92 GHz dual-polarized integrated horn antennas," 
IEEE Trans. Antennas Propag., vol. AP-39, June 1991. 

[4] W.Y. Ali-Ahmad, G.V. Eleftheriades, L.P. Katehi, and G.M. Rebeiz,"Millimeter-Wave 
Integrated Horn Antennas, Part II: Experiment," IEEE- Trans. Antennas Propagation, 
vol. AP-39, pp. 1582-1587, Nov. 1991. 

[5] G.V. Eleftheriades, W.Y. Ali-Ahmad, and G.M. Rebeiz, "A 20dB Quasi- Integrated Horn 
Antenna," IEEE- Microwave Guided-Wave Lett., vol. 2, pp. 72-75, Feb. 1992. 

[6] D.N. Held and A.R. Kerr, "Conversion loss and noise of microwave and millimeter-wave 
receivers: Part I-Theory; Part II- Experiment," IEEE Trans. Microwave Theory Tech., 
vol. MTT-26, p.49-61, 197S. 

[7] Duroid is a trademark of Rogers Corporation. We thank Rogers Co. for the donation of 
the substrate. 



Third International Symposium on Space Terahertz Technology Page 345 

ZONE PLATE LENS ANTENNAS FOR /4> oSy£~ 

P /? 
MILLIMETER AND SUBMILLIMETER WAVELENGTHS / ' 

Paul F. Goldsmith 

Five College Radio Astronomy Observatory 

Department of Physics and Astronomy 

University of Massachusetts, Amherst MA 01003 

and 

Millitech Corporation 

P.O. Box 109 

South Deerfield, Massachusetts 01373 



Abstract 



Zone plate lenses are a type of focusing element which function essentially as 
differential phase shifters, having a relatively few, coarsely quantized phase delays across 
the incident beam of radiation. The major advantages are ease of fabrication and much 
reduced thickness, compared to conventional refractive focusing elements. These 
considerations are both of particular importance for the submillimeter range, in which 
manufacturing tolerances for curved optical elements can pose significant problems, and 
where the absorption of readily available dielectric materials is quite large. In this 
presentation we briefly review the theory of zone plate lens operation, present a relatively 
simple method for calculating the aperture efficiency of zone plate lenses used as antennas, 
and show some theoretical and measured results in the 100 GHz range. 



Page 346 Third International Symposium on Space Terahertz Technology 

I. Lens Operation as a Phase Transformer 

An ideal lens changes the radius of curvature of an incident beam of radiation 
without affecting its amplitude distribution. This situation is illustrated in Figure 1, which 
shows a phase transformer converting a diverging spherical wave (as might be produced by 
a feed horn) into a plane wave. In the paraxial limit, the phase variation of a spherical 
wave perpendicular to an axis from its focus is a quadratic function of the radius r from the 
axis: 

A<p(i) = tttVAR , (1) 

where A is the wavelength and R is the radius of curvature. The phase variation is defined 
in the sense that the phase delay increases with increasing distance from the axis of 
propagation. 

A lens modifies the radius of curvature of the beam by making use of different 
propagation speeds in different media; these can be dielectrics for which the speed is less 
than that in free space by a factor n, and arrays of waveguides or metal plates for which 
the propagation speed is greater than c according to the general relationship 

v = c/[l-(A/Aco)2]0' 5 • (2) 

Reduced to simplest terms, a dielectric lens modifies the phase distribution by providing a 
phase delay which decreases away from its axis according to 

A^i(r) = -7rr2/AF , (3) 

where F is the focal length of the lens. The output phase variation is just the sum of that 
of the input beam given by (1) together with that of the lens given by (3), so that we 
obtain 

TnrVARout = 7rr2/AR in - ttt2/AF , (4a) 

which leads immediately to the relationship 

1/Rout = 1/Rin - 1/F . (4b) 



Third International Symposium on Space Terahertz Technology Page 347 

With the convention that Ri n = Di n , the distance of the focal point of the input beam from 
the lens, and R ou t = — Dout (the distance to the focal point of the output beam), we 
recover the familiar expression 

1/Din + 1/Dout = 1/F • (4c) 

Equation (4b) also applies to quasioptical Gaussian beams [1], while (4c), being dependent 
on the assumption that the radius of curvature is equal to the distance from the focal point, 
applies only to geometrical optics beams. 

A converging dielectric lens of index of refraction n has a central thickness t c 
determined by the maximum phase delay that is required: 

Ap,nax= (27r/A)-t c -(n-l) . (5) 

In the simplest approximation, we find that the central thickness of a lens of diameter D 
and focal length F is 

t c = D2/8(n - 1)F . (6) 



II. LENS LOSS 

A lens which operates perfectly as a phase transformer may still suffer loss as a 
result of reflections at free space — dielectric interfaces, and absorption within the lens 
itself. Reflection losses are typically a few percent per surface if no anti— reflection 
treatment is employed. The basic technique to reduce reflections is to include a matching 
layer, which may be a natural dielectric of the required index of refraction, or an artificial 
dielectric as formed by cutting grooves. Both approaches have limitations due to 
variations in incidence angle and polarization effects, but can reduce reflections 
significantly. 

Absorption cannot be eliminated, and depends on the lens thickness, together with 
the material properties. We use the definition of the fractional power loss per unit distance 
to be a, so that the input and output power (or power density) after traversing a path 
through the dielectric of length L are related by 



Page 348 Third International Symposium on Space Terahertz Technology 

Pout = Pin-exp(-aL) , (7a) 

where 

a = 2* n tan 6/ A . (7b) 

In the preceding equations, we have employed the usual definitions of the complex 

dielectric constant c = c' + ie", the index of refraction n = ft', and the loss tangent, 
tan 5= e"/e'. 

At submillimeter wavelengths, information on dielectric properties is quite scarce 
and often not entirely consistent. Some of this may be a result of variations in sample 
properties, while measurement techniques and errors may also be playing a role. Teflon is 
a low— loss dielectric widely used for millimeter and submillimeter wavelength lenses. This 
material has an absorption coefficient which rises almost linearly with frequency, and is 
approximately 0.042 cm"i at 300 GHz and 0.09 cm-' at 600 GHz [2]. Other measurements 
give higher absorptions of 0.2 - 0.5 cnr* at a 900 GHz [3]. We adopt an absorption 
coefficient of 0.1 cm* 1 at 600 GHz and a real part of the dielectric constant of 2.0. 
Rexolite tm is often used at millimeter wavelengths due in part to its good mechanical 
properties. It is relatively lossy in the submillimeter range; different measurements give a 
= 0.70 - 1.0 cm-i at 600 GHz [4]. 

Taking (6) as giving a representative lens thickness, we find t c = 0.31 D 2 /F. For a 
F = D = 5 cm lens at 600 GHz, we find t c = 1.55 cm; a perfectly phase correcting 
plano-convex lens of the same focal length and diameter has t c = 1.475 cm. The 
absorptive loss at the center of a F = D = 5 cm teflon lens (where most of the power is 
concentrated) will thus be about 15 %. A comparable rexolite tm lens will have an 
absorption at its center of approximately 60 %! Clearly, these numbers are large enough to 
suggest the use of refractive optics. An alternative that merits serious consideration is the 
zone plate lens, which can be made far thinner and thus have negligible absorption loss. 

III. Zone Plate Lens Operation 

The monotonically decreasing phase delay as a function of distance from the axis of 



c-s- 



Third International Symposium on Space Terahertz Technology Page 349 

symmetry characteristic of typical dielectric lenses (Figure 2a) can be interrupted by a step 
change in thickness. If at a particular frequency this produces an increase in the phase 
delay equal to 2tt radians, it will nominally not have any effect on lens performance, except 
for possible shadowing by step boundaries. Such devices are generally called zoned lenses, 
or Fresnel lenses and are widely used to reduce the thickness of relatively large and thick 
lenses employed at microwave frequencies. As shown in Figure 2b, they still have at least 
one surface which has curved sections, so that manufacturing is, in fact, more difficult than 
conventional lenses since the steps are an added complication. 

Zone plate lenses represent a more radical approach, in that, as illustrated in Figure 
2c, they are designed using only surfaces perpendicular to the axis of propagation. This is 
not at all a new concept, deriving quite directly from concept of Fresnel zones in diffraction 
theory. A number of references which discuss theoretical and experimental aspects of zone 
plate lenses are given in [5]. The closely related zone plate reflector antenna is discussed in 
references [6]. 

Rather than attempting to achieve the desired phase error function (3), the zone 
plate lens allows the phase error to increase as a quadratic function of distance from the 
axis. When it has reached a certain point, the lens thickness is reduced to bring the phase 
error to zero. If we define the maximum allowed phase error to be 27r/p, the axial size of 
the step is given by 

t!=A/(n-l)p . (8) 

If this procedure were continued indefinitely, we would merely have a stepped 
approximation to a conventional lens, which would not be particularly thin. The zone 
plate lens is distinguished by the technique of increasing the phase delay by 2ir radians at 
the design frequency, at points where the previously described procedure of reducing the 
phase delay would lead to an accumulated phase error of 2t radians. The maximum 
change in thickness of the zone plate lens is 

At.ax = [(p-l)/p]-(A/(n-l) , (9) 

since at the next step the thickness returns to its original value rather than to 
p-ti = A/(n — 1). The total zone plate lens thickness is given by At max + t m i n , where the 
latter is the minimum thickness required for mechanical integrity. The total thickness is 



Page 350 Third International Symposium on Space Terahertz Technology 

thus on the order of a wavelength, far less than required for unzoned lenses. The 
absorption loss for a material with a proportional to frequency will thus be a constant, 
providing one of the most important advantages of zone plate lenses at submillimeter 
wavelengths. 

The radii at which the zones occur are obtained by requiring that the total phase for 
rays representing a plane wave converged to focal point be constant for all initial radii from 
the axis of symmetry. We ignore any phase shift of constant thickness component of lens, 
and take the phase shift of the zone plate lens to be 

A^ p i = -27Tk/p , (10) 

where k is an index which increases by unity at each zone boundary. As illustrated in 
Figure 3, the radius of zone k is denoted rk and the distance from the lens at this radius to 
the focal point Rk, so that the path phase difference between axial ray and an arbitrary ray 
is 

Aip = (27r/A)[R k - F] . (11) 

The two previous equations can be combined with the constant total phase condition to 
determine Rk. With the additional approximation of ignoring effect of changes in the lens 
thickness on Rk, we obtain Rk = [rk 2 + F 2 ]°« 5 , which gives the relation 

r k =[2kFA/p + (kA/p)2]o.5 . (12) 

Some designs for zone plate lenses are shown in Figure 4, with p = 2, 4, 10, and 50. All are 
designed for 300 GHz and have F = D = 10 cm, with an index of refraction equal to 1.4. A 
minimum thickness of 0.1 cm has been arbitrarily chosen. In practice, it is effective to 
chose the minimum thickness to make the zone plate lens central thickness resonant at the 
design frequency. 

If we restrict ourselves to the situation F/D >> 0.5, the first term in (12) 
dominates, and we obtain the nominal lens diameter 

D = [8k max fA/p]o.5 . (13a) 



Third International Symposium on Space Terahertz Technology Page 351 

Alternatively, we see that the number of zones in the lens is given by 

k max = pD 2/8fA . (13b) 

Although zone plate lens design concentrates on phase delays and ignores refraction, an 
important limitation must be borne in mind, which is that the zone width must be large 
enough that zones do not begin to act like waveguides, in the manner of matching layers on 
a conventional dielectric lens. For this reason it is useful to determine the minimum zone 
width, which is just the minimum value of Ar = rk+i — rk- This occurs at the outer radius 
of the lens where we find 

Ar min = (2/p).(f/D).A . (14) 

For a F/D = 1 zone plate lens with p = 4 we find Ar m in = A/2, which is on the borderline 
of being a problem; it is apparent that the performance of the outer portion of fast zone 
plate lenses may be compromised by this effect. Detailed calculations remain to be carried 
out. 

IV. Zone Plate Lens Efficiency 

The major issue we wish to investigate is how the wave front errors which are a 
necessary consequence of the approximate nature of the zone plate lens design affect its 
efficiency as an antenna. We consider a lens being used to transform a spherical wave into 
a plane wave and start with a wave of radius of curvature Ri n ; the phase distribution as a 
function of distance from the axis is shown in Figure 5a; in the paraxial limit this is just 
that described by (1). To this we add the differential phase shift produced by the lens 
which has thickness t at radius r 

A^ zp i = (27r/A)-(n-l)t(r) . (15) 

The lens shown in cross section in Figure 5b has p = 4 and F/D = 1. The resulting output 
or aperture phase distribution is shown in Figure 5c; note that the phase delay increases 
essentially quadratically as a function of radius, except at the zone boundaries. The p = 4 
lens has 3 successive zone boundaries at which the phase delay decreases by 27r/4 radians, 
followed by a boundary at which the phase delay increases by 27r. 



Page 352 Third International Symposium on Space Terahertz Technology 

In order to calculate the aperture efficiency, we assume that we have a Gaussian 
feed distribution which yields an aperture field distribution with magnitude of the form 

I Eap(r) | = exp Kr/w)2] , (16a) 

which defines the power edge taper 

Te(dB) = 8.69 (R/w)2 = 2.17 (D/w)2 . (16b) 

The taper efficiency is the efficiency with which the aperture is utilized, and is defined by 

ct= |// E ap -dS|2/// |E ap |2.dS • //dS , (17a) 

where all integrals extend over the aperture [7]. The spillover efficiency is the fraction of 
power in the feed pattern which is intercepted by the aperture, and is given by 

e s = // |E ap |2.dS / // |E ap |2.dS . (17b) 

aperture entire 

pat tern 

The aperture efficiency is the product of the two preceding contributions: 

c a = ft • fs • (17c) 

The integral in the numerator of (17a) includes the effects of the phase errors; any 
deviation from a uniform phase distribution reduces the taper efficiency and thus the 
aperture efficiency. 

The efficiencies can be calculated for different input beam characteristics for a given 
lens. An edge taper of approximately 10 dB yields the maximum efficiency for this type of 
illumination of an unblocked antenna [7]. The variation of taper efficiency as a function of 
input beam radius of curvature is shown in Figure 6a for a zone plate lens with p = 4 and 
F = D = 10 cm operating at a wavelength of 0.3 cm. As expected, the maximum efficiency 
occurs for Ri n = F. 

The behavior of the lens efficiency as a function of p is shown in Figure 6b, for the 
same lens conditions as above, fixing Rin = F. The values of the efficiencies for large p are 



Third International Symposium on Space Terahertz Technology Page 353 

very close to those for a perfect phase transformer with the same 10 dB edge taper: 
ft = 0.9, e s = 0.9, and e a = 0.81. The efficiencies for p = 2 and 3 are quite low, but for 
p = 4 we begin to approach the asymptotic behavior. Thus, the choice of p represents a 
compromise between obtaining the highest efficiency and ease of fabrication together with 
the requirement on minimum zone width given by (14). 

V. Zone Plate Lens Measurements 

We have fabricated and measured a zone plate lens designed for operation at 95 
GHz, where test equipment is readily available. The lens was fabricated of Rexolite tm (n = 
1.59), with p = 4, f = 12.7 cm, D = 9.53 cm, and a central thickness of 0.59 cm. For 
comparison, we used a fused silica lens which was anti— reflection coated with layers of 
polyethylene. This lens had the same diameter, but a slightly different focal length of 
14.5 cm. Both lenses were illuminated by a scalar feed horn giving a Gaussian illumination 
pattern with an edge taper of close to 10 dB. The measured patterns in one plane are 
shown in Figure 7. We see that the main lobe beamwidths are very similar. The sidelobe 
structure of the unzoned lens is essentially that predicted from the truncated Gaussian 
illumination. The zone plate lens shows more extended error pattern which is a 
consequence of the phase errors. 

The gains of the two lens antennas were also measured using a compact range; their 
absolute values are compromised by uncertainty in the gain of the reference horn but are 
consistent with expectations. What is more reliable is the difference in gain between the 
unzoned lens and the zone plate lens, which indicate that the zone plate lens has 1.0 dB 
lower gain. The calculations for the p = 4 zone plate lens predict an efficiency 0.86 dB 
below that of an ideal phase transformer lens. The reflection loss of the n = 1.6 zone plate 
lens is 0.23 dB per surface at normal incidence. Given the possible imperfections in 
matching, phase transforming, and the absorption in the unzoned lens, the measurements 
and calculations are in satisfactory agreement. 



VI. Conclusions 

We have reviewed the theory of operation and design of zone plate lenses. Their 
very small thickness makes these devices attractive for use at submillimeter wavelengths 
where absorption loss of unzoned lenses can be appreciable. We have examined the 



Page 354 Third International Symposium on Space Terahertz Technology 

efficiency of zone plate lenses as a function of interzone phase shift, and find that for zone 
boundary phase shifts < ir/2 the performance approaches that of an ideal phase 
transformer. Measurements and calculations of the efficiency of a 95 GHz lens agree quite 
well. 

We thank John Kapitzky, Chris Koh and Ellen Moore for their contributions to this 
project. 

References 

[1] T.S. Chu, "Geometrical Representation of Gaussian Beam Propagation," Bell Syst. 
Tech. J., 45, PP- 287-299, 1966. 

[2] M.N. Afsar, "Millimeter Wave Complex Refractive Index, Complex Dielectric 
Permittivity and Loss Tangent Measurements of Common Polar and Non-Polar 
Polymers," Proc. Tenth International Conference on Infrared and Millimeter Waves, 
pp. 60-61, 1985. 

[3] A.P. Sheppard, A. McSweeney, and K.H. Breeden, "Submillimeter Wave Material 
Properties and Techniques: Dielectric Constant, Loss Tangent, and Transmission 
Coefficients of Some Common Materials to 2000 GHz", Proc. Symposium 
Submillimeter Waves, Vol. XX in Microwave Research Institute Symposia Series. 
Brooklyn: New York Polytechnic Institute Press, 1970, pp. 701-705. G.W. 
Chantry, SubmiUimetre Spectroscopy. New York: Academic, 1971, p. 341. 

[4] G.J. Simonis, J.P. Sattler, T.L. Worchesky, and R.P. Leavitt, "Characterization of 
Near— Millimeter Wave Materials by Means of Non— Dispersive Fourier Transform 
Spectroscopy," Int. J. Infrared and Millimeter Waves, 5, pp. 57-72, 1984. R.H. 
Giles, A.J. Gatesman, and J. Waldman, "A Study of the Far— Infrared Optical 
Properties of Rexolitet m ," Int. J. Infrared and Millimeter Waves, 11, pp. 1299-1302, 
1990. 

[5] M. Sussman, "Elementary Diffraction Theory of Zone Plates," Amer. J. Phys., 28, 
pp. 394-398, 1960. F. Sobel, F.L. Wentworth, and J.C. Wiltse, "Quasi-Optical 
Surface Waveguide and Other Components for the 100- to 300 Gc Region," IEEE 
Trans. Microwave Theory Tech., MTT-9 . pp. 512-518, 1961. D. N. Black and J.C. 



Third International Symposium on Space Terahertz Technology Page 355 

Wiltse, "Millimeter-Wave Characteristics of Phase— Correcting Fresnel Zone 
Plates," IEEE Trans. Microwave Theory Tech., MTT-35 . pp. 1123-1129, 1987. 
J.E. Garrett and J.C. Wiltse, "Fresnel Zone Plate Antennas at Millimeter 
Wavelengths," Int. J. Infrared and Millimeter Waves, 12, pp. 195-220, 1991. This 
includes a particularly complete of references on this topic. 

[6] L.F. Van Buskirk and C.E. Hendrix, "The Zone Plate as a Radio— Frequency 
Focusing Element," IRE Trans. Antennas Propag., AP-9 . pp. 319-320, 1961. Yu. 
N. Danilov and L.A. Fedorova, "Scattering Behavior in a Zoned Reflector 
Antenna," Izvestiya VUZ. Radioelektronika, 32(2) . pp. 61-65, 1989. R. Lambley, 
"Fresnel Antenna," Electronics & Wireless World, 95, P- 1642, 1989. J.M. Franke 
and B.D. Leighty, "Reflection Zone Plate Antenna," NASA Tech. Brief 
LAR-1S5S5, 1989. M.A. Gouker and G.S. Smith, "A Millimeter-Wave Integrated 
Circuit Antenna Based on the Fresnel Zone Plate," 1991 IEEE MTT-S Digest, pp. 
157-160. 

[7] P.F. Goldsmith, "Radiation Patterns of Circular Aperture with Gaussian 
Illumination," Int. J. Infrared and Millimeter Waves, 8, pp. 771-781, 1987. 



Page 356 



Third International Symposium on Space Terahertz Technology 



INPUT BEAM 
RADIUS OF CURVATURE 




1 


OUTPUT BEAM 






i 





PHASE 

TRANSFORMER 

A¥ (r) 



FIGURE 1 - OPERATION OF LENS AS PHASE TRANSFORMER 



FOCAL 
POINT 

» 



IS 



fe 



(a) UNZONED 
LENS 



(b) ZONED 
LENS 



(c) ZONE PLATE 
LENS 



FIGURE 2 - DIFFERENT DESIGNS OF LENSES 



Third International Symposium on Space Terahertz Technology 



Page 357 




FOCAL POINT 



FIGURE 3 - ZONE PLATE LENS DESIGN PARAMETER DEFINITIONS 



GS I I 84 



Page 358 



Third International Symposium on Space Terahertz Technology 



P = 2 



P = 4 




Radius CcttO 




Radius Ccm) 



P = 10 



P = 50 




u 9.1 



c 




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ZONE PLATE LENSES WITH F = D = 10 CM ; FREQUENCY = 300 GHz ; n = 1.4 

FIGURE 4 



Third International Symposium on Space Terahertz Technology 



Page 359 



FIGURE 5 
100 GHZ ZONE PLATE LENS 

FL = 10 cm DIA = 10 cm 

n = 1.59 • p = A 

INPUT PHASE DISTRIBUTION 



a 
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Page 360 



Third International Symposium on Space Terahertz Technology 



>• 
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Third International Symposium on Space Terahertz Technology 

FIGURE 7 

94GHz LENS COMPARISON 



Page 361 










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GS1179 



Page 362 Third International Symposium on Space Terahertz Technology 

ONSET OF DISPERSION IN Nb MICROSTRIP TRANSMISSION LINES AT 
SUBMILLIMETER WAVE FREQUENCIES 



H. H. S. Javadi, W. R. McGrath, B. Bumble, H. G. LeDuc 



53^-33 

n *)fo Center for Space Microelectronics Technology 

f ^ Jet Propulsion Laboratory ftQ Q 7 7 CC 8 

California Institute of Technology N V o ~ £ * ( 
Pasadena, CA 91109 

^^ ABSTRACT 

/ ' 

! We have measured the dispersion in phase velocity of a Nb-SiO x -Nb 
, microstrip transmission line resonator over a frequency range from 50 
GHz to 800 GHz. A submicron Nb/AI-AIO x /Nb Josephson junction was used 
as a voltage-controlled oscillator to excite the high order modes in the 
resonator. The same junction is used as a direct detector resulting in a 
series of step-like structures in the DC current-voltage characteristic at 
the position of each mode frequency. The transmission line is 
dispersionless up to about 500 GHz where the phase velocity begins to 
decrease. This is well below the gap frequency f g = 700 GHz. Results 
agree qualitatively with the expected theoretical behavior near f g . This 
onset of dispersion and loss in Nb transmission lines will have a 
significant impact on the design of submillimeter wave rf circuits. 



Third International Symposium on Space Terahertz Technology Page 363 

Superconducting transmission lines have many important 
applications in high frequency analog circuits and high bit rate digital 
systems, to name a few. These transmission lines are expected to be 
nearly lossless and dispersion free up to frequencies near the 
superconducting energy gap frequency f g = 2A / h, where A is the 

superconductor energy gap parameter and h is Planck's constant. We are 
interested in superconductive microstrip transmission lines as integrated 
rf tuning elements for superconductor-insulator-superconductor (SIS) 
quasiparticle mixers at frequencies near 630 GHz. These SIS mixers 
provide near quantum-limited sensitivity throughout the millimeter wave 
band [1-3], provided they are optimized with an appropriate rf embedding 
circuit. The large geometrical capacitance Cj of the tunnel junction 
provides a susceptance which shunts the rf signal away from the nonlinear 
quasiparticle conductance channel. An open-circuited superconductive 
microstrip transmission line (stub) can provide a parallel inductance to 
resonate with the junction capacitance. This approach was first used 
successfully with an SIS mixer operating near 36 GHz [4] and since has 
been used at frequencies up to 360 GHz [5,6]. However, above 100 GHz, 
uncertainties in material, transmission line, and junction parameters 
generally lead to poor performance. 



Page 364 Third International Symposium on Space Terahertz Technology 

The phase velocity must be accurately known in order to correctly 
design the microstrip stub for a particular application. At high 
frequencies possible dispersion and loss are expected to degrade the 
performance. It is thus important to know the frequency range at which 
these effects become significant. Other groups [7,8] have measured short- 
pulse propagation on superconductive coplanar transmission lines using 
optical sampling techniques and indirectly determined loss and dispersion 
from a Fourier transform analysis of the pulse distortion. 

We have taken a different approach, using a Josephson junction as 
both a sweep-oscillator and detector to sample the resonances of a 
microstrip stub. This stub is connected in parallel across the junction as 
shown in fig. 1. The ac Josephson effect provides a voltage-controlled 
oscillator. The frequency is 2eVyh = 0.485 x Vb (u.V) GHz where Vb is the 
junction bias voltage. This monochromatic signal sets up a standing wave 
in the stub. The junction detects this standing wave and a signature is 
developed in the DC current-voltage (l-V) curve of the junction by a 
frequency downconversion process. As the bias voltage, and thus frequency 
is swept, small current steps will appear in the DC l-V curve of the 
junction at frequencies corresponding to the modes of the stub. The 



Third International Symposium on Space Terahertz Technology Page 365 

frequency spacing of the modes is given by Af = v$/2/ where v$ is the 

phase velocity and / is the physical length of the stub. Thus by measuring 
the voltage interval between steps, the phase velocity can readily be 
obtained if it is constant over this interval. In general, this technique can 
be used at high frequencies well above the energy gap frequency f g . A 
similar approach using microstrip resonators with a single resonance 
below f g has recently been reported [9]. Values of surface resistance and 
phase velocity were determined up to about 400 GHz. We have used long 
microstrip resonators with many high order modes to determine the phase 
velocity as a function of frequency up to 800 GHz, which is above the gap 
frequency of niobium (Nb). The onset of dispersion has clearly been 
observed. 



High quality Nb / AI-AlOx / Nb tunnel junctions are fabricated using 
a trilayer process [10]. The junction area, defined by electron-beam 
lithography, is between 0.3u2 and 1u.2. The DC sputtered Nb films are 2000 - 
3300 A thick. The current density is as high as 10 4 A/cm2 and the normal 
state resistance is 50Q - 70Q. SiO x serves as both a planarization layer 
for the junction trilayer and the dielectric layer for the microstrip stub. 



Page 366 Third International Symposium on Space Terahertz Technology 

The thickness is U » 1500A and the dielectric constant is taken to be e r = 

5.5 [11]. The dielectric thickness is comparable to the magnetic 
penetration depth. In this case, the phase velocity is << c and hence is 
strongly affected by the field penetration into the Nb. This makes v$ a 

sensitive probe of the superconductive properties of Nb. 

Junctions fabricated for use as SIS mixers have stubs 2ji wide by 
60u, - 70u, long to provide a fundamental broad band resonance near 630 
GHz. However, to study the phase velocity, junctions were fabricated with 
stubs 2u. wide by 51 8u. or 1064u. long. A longer stub gives a smaller 
spacing between resonant modes, thus providing a better indication of the 
phase velocity in a given frequency range. Careful examination of these 
stubs however, indicated jagged edges and undercutting of the SiO x . 

Additional junctions were fabricated with stubs 6u. and 12u. wide to 
reduce uncertainties due to these edge effects. These stubs were either 
500u,, 750{i, or 1000|i long. 

The fundamental resonance frequency of a 1000u, long stub is 
estimated to be about 50 GHz [4]. This leads to steps in the l-V curve 
which are =0.1 mV apart. The inset of fig. 2 shows one example. The 



Third International Symposium on Space Terahertz Technology Page 367 

subgap conductance of this junction is very large with an almost point- 
contact type of appearance. While junctions with much lower subgap 
conductance also showed resonant peaks, the step structure was seen 
mostly in junctions similar to that shown in fig. 2. An external magnetic 
field was used to enhance the steps in different voltage ranges [12]. In 
order to accurately locate these small steps in current, a small amplitude, 
low frequency ( ~ 200 Hz ) ac voltage was superimposed on Vt,. The 
resulting ac signal was detected with a lock-in amplifier. The interval 
between two adjacent peaks is then measured to determine v$ which is 

plotted as a function of the frequency of the higher step. Due to the 
discreteness of the data, the resolution for changes in velocity is the 
frequency spacing of the modes. Figure 2 shows the results for 4 different 
microstrip stubs. About 10 to 14 resonances were observed between 50 
GHz and ~ 750 GHz. A smooth curve was drawn through the resulting 
velocity vs frequency data. The scatter in velocity values about this curve 
is typically 5%, with only a few points deviating by ~ 10%. 

Curve 1 is an example of a junction with a 2|i wide stub. Resonances 
of these narrow stubs were usually not observable above about 500 GHz. 
This may be due to increased scattering or losses from the edge effects 



Page 368 Third International Symposium on Space Terahertz Technology 

mentioned earlier. Curves 2, 3, and 4 represent stubs "lOOOu, long with 
widths of 6u. or 12u.. At low frequencies, a slight curvature can been seen 
in the data as expected from the mode spacing (see discussion below). 
Otherwise the curves are horizontal up to about 500 GHz where they begin 
to bend down indicating the onset of dispersion. The gap frequency of a 
pristine Nb film is f g ~ 740 GHz at OK. Thus the dispersion begins at 
frequencies well below f g . This agrees with previously reported results [8] 

using an optical sampling technique. While SIS mixers are predicted to 
operate well up to f g [13], the resonant embedding circuit utilizing 

superconductive microstrip transmission lines is limited to lower 
frequencies. 

Figure 3 shows the data for a 500u. long x 12ji wide stub. In this 
case, resonances, and hence the phase velocity, were obtained up to = 
800GHz which is above the gap frequency of Nb. At low frequencies, the 
velocity is dispersionless. Around 500 GHz, the velocity begins to 
decrease and reaches its lowest value near 730 GHz. For this data point, 
the frequency resolution for velocity change is about 50 GHz. This 
minimum is expected theoretically to occur near 770 GHz (see fig. 3) using 
a gap frequency of 660 GHz as determined from the l-V curve of this 



Third International Symposium on Space Terahertz Technology p a g e 359 

junction at zero magnetic field (a more complete discussion of the theory, 
will be given in the future [14]). In addition, we have observed a small 
suppression of the energy gap with external magnetic field in these 
junctions which may describe the lower frequency of the observed phase 
velocity minimum. Other loss mechanisms may also play a role. The 
velocity begins to increase above 730 GHz as is expected theoretically, 
since the superconductor begins to behave as a normal metal for 
frequencies well above f g . The solid line in fig. 3 is the theoretical 
prediction for the velocity. We have used the approach followed by Kautz 
[15] which employs the Mattis-Bardeen [16] theory for the electrical 
conductivity. The magnetic penetration depth was adjusted to X = 900A to 

fit the theory to the low frequency asymptote of the velocity. As seen 
from fig. 3, theory and experiment show the same general trend. However, 
based on the Mattis-Bardeen conductivity, the theory predicts a 9% 
decrease in velocity at the dip whereas the experiment shows a 35% dip. 
Errors in velocity resulting from uncertainties in the mode spacings can 
result from end effects on the microstrip line and the possibility of a 
► negative resistance loop in the l-V curve at the position of the resonance 
[17], However for our geometry, the end-effect correction to the length is 
<< 1%, and the worst-case negative resistance, predicted for a lightly- 



Page 370 Third International Symposium on Space Terahertz Technology 

damped resonance, could only cause a 5% shift in the apparant position of 
the resonance. Thus neither effect can account for the large change we 
observe. At low frequencies, superconductive microstrip is a slow-wave 
transmission line due to the penetration of the electromagnetic field into 
the superconductor over a length comparable to the thickness of the 
dielectric. For frequencies well above f g , the phase velocity increases 
towards the value for a normal line. The onset of dispersion just below 
the gap frequency is due to the departure of the imaginary part of the pair 
conductivity from a 1/f frequency dependence [15]. 

Some additional insight may be gained by first considering a 
Josephson junction directly coupled to a lossless microstrip stub. Using 
the notation in reference [18], the resonant frequencies of the stub are 
solutions to the transcendental equation 



■n$± -L _ » ( -*£-) ■■ (1) 



'S «S 



where C s = e r eo/w/td is the total capacitance of the stub and f s is the 
fundamental mode of the stub. This mode occurs when the length of the 



Third International Symposium on Space Terahertz Technology Page 371 

stub,/, equals one wavelength and is given by f s = v^ / 21. 



In fig. 4 both sides of eqn. (1) are plotted vs (f/f s ) for an arbitrary 
value of Cj/C s . At low frequencies, intersections between the straight 
line ( left hand side of eqn. 1) and the tangent curves are between (m- 
1/2)f s and mf s where m is an integer. These solutions move progressively 
toward (m-1/2)f s for m » 1. Thus at high frequencies, the solutions of 
eqn. (1) (i.e.: the modes of the stub) are equidistant. If plotted in the spirit 
of fig. 2, they represent a horizontal line. 

We have extended eqn. (1) to the case of a lossy transmission line, 
yielding 



C, f 2s.n(-^- ) 

-K-r- t - ^~r- < 2 > 

°S 's (e 2a/ + e -2a/) + ^ ( 2*1) 

T s 



where a is the attenuation constant. In fig. 4, the right hand side of eqn. 



Page 372 Third International Symposium on Space Terahertz Technology 

(2) is plotted for three arbitrary cases of loss: a/ = 0.05, 0.10, and 0.15. 
As can be seen, the distance between the solutions first decreases and 
then increases as losses in the microstrip line increase. In light of this, 
our experimental observations can be interpreted as evidence for 
increased loss in either the superconductor or the dielectric layer. Losses 
due to radiation will be negligible given the cross sectional dimensions of 
the stub, 0.1 5ji x 12ji, compared to relevent free space wavelengths, X Q ~ 

500ji [19]. It is expected that absorption of water in evaporated SiO x 
films could contribute losses at microwave and millimeter wave 
frequencies. Losses due to absorbed water molecules constitute broad 
peaks in the frequency domain. The attenuation of a microstrip line is 
linearly proportional to frequency provided the loss in the dielectric is 
frequency independent. A linear dependence is much weaker than the sharp 
increase in attenuation expected near the gap frequency. Moreover, since 
we observe a strong dispersion near the Nb gap frequency, the 
superconductors are the most probable source for the losses. The 
superconducting electrodes of the microstrip line are not perfect, defect 
free, bulk crystalline materials, but are polycrystalline thin films with 
fine grains and possibly surface layers and interface defects. These 



Third International Symposium on Space Terahertz Technology Page 373 

microstructural features of real films may be responsible, in part, for the 
large dispersion we observe. 

In summary, we have presented evidence for the onset of dispersion 
and loss at submillimeter wave frequencies in Nb-SiO x -Nb microstrip 
transmission lines. This behavior is expected at frequencies approaching 
the gap frequency, but the range over which Nb microstrip lines are 
dispersionless was previously not well known. These results will have a 
direct impact on the application of Nb microstrip lines in millimeter wave 
and submillimeter wave circuits. For operation near 1 THz, higher 
temperature superconductors such as NbN or NbCN will have to be 
investigated. 

This work was supported in part by the Jet Propulsion Laboratory, 
California Institute Of Technology, under contract to the National 
Aeronautics and Space Administration and the Innovative Science and 
Technology Office of the Strategic Defense Initiative Organization. 



Page 374 Third International Symposium on Space Terahertz Technology 



REFERENCES 



1. C.A. Mears, Q. Hu, P.L Richards, A. H. Worsham, D.E. Prober, and A.V. 
Raisanen, Appl. Phys. Lett. 57, 2487 (1990). 

2. H.H.S. Javadi, W.R. McGrath, S.R. Cypher, B. Bumble, B.D. Hunt, and H.G. 
LeDuc, Digest, 15th. Int. Conf. on IR and Millimeter Waves, p. 245, Orlando, 
FL (1990). 

3. B.N. Ellison, P.L Schaffer, W. Schaal, D. Vail, and R.E. Miller, Int. J. IR and 
mm Waves 10, 937 (1989). 

4. A.V. Raisanen, W.R. McGrath, P.L. Richards, F.L. Lloyd, IEEE Trans. Microwave 
Theory Techn. MTT-33, 1495 (1985). 

5. Q. Hu, C.A. Mears, P.L. Richards, and F.L. Lloyd, IEEE Trans. Magn. 25, 1380 
(1989). 

6. W.R. McGrath, J.A. Stern, H.H.S. Javadi, S.R. Cypher, B.D. Hunt, H.G. LeDuc, 
IEEE Trans. Magn. 27, 2650 (1991). 

7. M.C. Nuss and K.W. Goossen, IEEE J. Quantum Electronics 25, 2596 (1989). 

8. C.C. Chi, W.J. Gallagher, I.N. Duling III, D. Grischkowsky, N.J. Halas, M.B. 
Ketchen, and A.W. Kleinsasser, IEEE Trans. Magn. MAG-23, 1666 (1987). 

9. B. Bi, K. Wan, W. Zhang, S. Han, J.E. Lukens, IEEE Trans. Appl. 
Superconductivity 1, 145 (1991). 



Third International Symposium on Space Terahertz Technology Page 375 

0. H.G. LeDuc, B. Bumble, S.R. Cypher, and J.A. Stern, submitted to 3rd. 
International Symposium on Space Terahertz Technology, University of 
Michigan, Ann Arbor, March 1992. 

1. H.K. Olsson, IEEE Trans. Magn. 25, 1115 (1989). 

2. I.O. Kulik, JETP Lett. 2, 84, 1965. 

3. M.J. Feldman, Int. J. IR and mm Waves 8, 1287 (1987). 

4. H.H.S. Javadi and W.R. McGrath, to be published. 

5. R.L Kautz, J. Appl. Phys. 49, 308 (1978). 

6. D.C. Mattis and J. Bardeen, Phys. Rev. 111, 412 (1958). 

7. D.B. Tuckerman and J.H. Mageriein, Appl. Phys. Lett. 37, 241 (1980). 

8. H.D. Jensen, A. Larsen, J. Mygind, IEEE Trans. Magn. 27, 3355 (1991). 

9. T.C. Edwards, Foundations for Microstrip Circuit Design , John Wiley and 
Sons, New York (1981). 



Page 376 Third International Symposium on Space Terahertz Technology 

FIGURE CAPTIONS 

Figure 1. (a) Geometry of open-circuited microstrip stub and Josephson 
junction, (b) Cross sectional view of stub and junction. 

Figure 2. Phase velocity vs. frequency for four different Nb microstrip stubs 
at 4.2K. The stub dimensions are 1: 2u. x 1064u.; 2: 12u. X "lOOOu,; 3: 6u. X 
"lOOOu.; 4: 6u, X 1000^. Inset shows current steps in DC IV curve 
associated with high order modes in the stub. 

Figure 3. Phase velocity vs. frequency for a Nb stub 12ji wide X 500u, long. The 
different symbols refer to data taken with different applied external 
magnetic fields. The solid line is the theoretical prediction. The arrow 
indicates the gap frequency as determined for the Nb junction (at zero 
magnetic field) which is used in the calculation. 

Figure 4. Graphical representation of both sides of the transcendental eqns (1) 
and (2). X-axis is frequency normalized to f s (fundamental resonance of 
the stub). Straight line is a plot of the left hand side of eqns (1) and (2) 
with arbitrary slope. Right hand side of eqn (1) is represented by tangent 



Third International Symposium on Space Terahertz Technology Page 377 

curves while right hand side of eqn (2) is plotted for values of a/ = 0.05, 
0.10, 0.15. Intersections of the straight line and the tangent curves 
represent the modes of the stub. When losses are significant ( a/ > ), the 

mode frequencies are determined by the closest approach of the two 
curves as indicated by the arrows. 



88 






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Page 382 Third International Symposium on Space Terahertz Technology 



The material presented below is intended as a review only. A full length paper has been 
submitted for publication in IEEE/MTT (May 1992). 



DOUBLE-SLOT ANTENNAS ON EXTENDED HEMISPHERICAL 

DIELECTRIC LENSES MQ^-27759 

Daniel F. Filipovic, Steve J. Gearhart, Brian K. Kormanyos and Gabriel M. Rebeiz 
/LOW? 



y 



j ^ \ "^- NASA/Center for Space Terahertz Technology 

j Electrical Engineering and Computer Science Department 

University of Michigan 
Ann Arbor, MI 48109-2122 



ABSTRACT 

An investigation of the coupling efficiencies to a gaussian-beam of a double-slot antenna on 
a hyperhemispherical lens is presented. It is shown that both lenses couple equally well to 
an appropriate gaussian beam (about 80%). The radiation patterns of both lenses with a 
double-slot antenna are computed using the ray-tracing method. The experimental radiation 
patterns are presented and show close agreement to the theoretically computed patterns. 



Third International Symposium on Space Terahertz Technology Page 383 

I. INTRODUCTION 

The use of a hemispherical lens with an attached extension length can greatly improve 
coupling to a gaussian-beam system. In optical theory, an extension length of r/n is used, 
and this extended lens is termed a hyperhemispherical lens. This extension length was chosen 
since it satisfies the sine condition, which is where first-order aberrations are removed [1]. 
The hyperhemispherical lens was borrowed into the millimeter- wave field [2,3,4], but it was 
found that radiation patterns from these lenses were very broad and even multi-lobed in 
some cases. The hyperhemispherical lens is capable of coupling well to a gaussian-beam 
system. However, it couples most efficiently to a converging beam and not to a plane 
wave. Recently, several researchers showed that a narrow, diffraction-limited beam could 
be achieved by putting the antennas on an elliptical lens [5,6]. The same effect was also 
found by taking a hyperhemispherical lens and adding a planar extension to it [7]. Figure 1 
shows that the focus of this longer extension length lens superimposes exactly on the second 
focus of an elliptical lens. It is known from optical theory that a plane wave converges to 
the second focus of an ellipse, and therefore a lens with this extension length is simply a 
close geometrical approximation to an elliptical lens. The validity of this approximation 
depends on the maximum allowed phase tolerance. For high dielectric constants (see Fig. 
1) and relatively low frequencies, the phase difference becomes small and the approximation 
is valid. Generally, for lens diameter of 12.5mm, e larger than 4, and frequencies less than 
300GHz, the approximation is very good. 



Page 384 Third International Symposium on Space Terahertz Technology 

II. THEORETICAL AND EXPERIMENTAL PATTERNS 

The theoretical radiation patterns are computed using a ray- tracing technique [9]. First, 
the feed antenna pattern into the dielectric is calculated using standard far-field methods. 
Figure 2 shows the calculated radiation patterns for a double-slot antenna with L = 0.28A at > 
and d = 0.16A at >. These parameters were chosen to result in a symmetric pattern inside the 
dielectric and a low cross-polarization in the 45° -plane. Ray-tracing is then used to calcu^ 
late the electric field distribution across the aperture plane (Fig. 3). In this method, the 
fields are decomposed into TE/TM components at the lens/air interface, and the appropri- 
ate transmission formulas are used for each mode. The power reflected into the substrate 
is neglected in this analysis. A diffraction integral over the aperture then yields the far- 
field pattern from the lens. Experimental measurements were performed at 246GHz on a 
13.7 mm diameter silicon lens (e=11.7) with the double-slot antenna as a feed. Different 
values of extension length were achieved by adding high-resistivity silicon wafers, resulting 
in 3 extension lengths: hyperhemispherical, intermediate, and elliptical (Fig. 3). Measured 
patterns at the elliptical focus (Fig. 4) demonstrate a gain of 28.6dB±0.3dB with relatively 
low sidelobes (-16dB). From the measured patterns, the resulting aperture efficiency (cou- 
pling to a plane wave) is 73%. The theoretical patterns calculated for this position are a bit 
wider than the measured patterns (Fig. 5). This discrepancy arises from the fact that rays 
at a certain angle end up hitting the critical angle at the lens/air interface, resulting in no 
transmission of rays after this point. This limits the aperture size and results in a wider theo- 
retical pattern. Note that this discrepancy is only significant at the elliptical focus for lenses 
with high dielectric constants. Measured patterns at the elliptical focus for ±10% of the 



Third International Symposium on Space Terahertz Technology Page 385 

246GHz design frequency (Fig. 6) result in nearly the same gain, and therefore the double- 
slot antenna has good pattern bandwidth. The measured power at broadside is nearly the 
same from 222GHz-270GHz, also indicating good impedance bandwidth for the double-slot 
design. The measured patterns at the intermediate focus (Fig. 7) are similar to the elliptical 
focus, but with a gain of 24dB±0.3dB. In this case, the critical angle is not a problem, and 
there is close agreement between theory and experiment (Fig. 8). At the hyperhemispherical 
focus (Fig. 9), the pattern becomes very wide with a gain of 18.1dB±0.3dB and shows a 
multi-peak behaviour, as indicated by theory (Fig. 10). As will be seen later, this has no 
detrimental effect on the coupling efficiency to a converging beam. The ratio of the 246GHz 
measured received power at broadside for an elliptical lens and a hyperhemispherical lens 
was lOdB which is the same as the difference in the measured directivities. This indicates 
that no power is coupled to substrate modes that may arise in the flat wafers. 

III. GAUSSIAN-BEAM COUPLING 

In order to match the double-slot/extended hemisphere system to a gaussian beam, one 
could compute the electric field across the aperture and match this to a gaussian beam. 
Since we had already predicted the far-field amplitude and phase distributions, we chose to 
compute the coupling efficiency to a gaussian beam in the far- field (see Appendix). In this 
calculation, the power radiated by the slot antennas to the air-side (which is 11.5% of the 
total power) is taken into account, and no lens-air interface loss is considered. The power 
loss radiated to the air side could be reflected using an appropriately designed cavity at 
the expense of impedance bandwidth. Figure 11 gives the gaussian-beam parameters which 



Page 386 Third International Symposium on Space Terahertz Technology 

yield the highest coupling efficiency, and shows that all three focus positions are capable 
of coupling equally well to a gaussian beam. However, the non-elliptical foci require a 
converging wavefront, whereas the elliptical focus couples directly to a Gaussian beam with 
an equal phase wavefront. Note that equivalent gaussian-beam parameters in the near field 
may be found through a simple inverse Fourier transform. A gaussian beam experiment was 
performed at 246GHz, in which it was attempted to couple all the power coming out of a lens 
into the double-slot antenna. For the elliptical focus, the lens was placed at the minimum 
waist position, where the radius of curvature is infinite, indicating an equal phase wavefront. 
For the hyperhemispherical position, the lens was placed closer to the lens, at a position 
where there is a negative radius of curvature. The proper negative radius of curvature 
and position were computed knowing the gaussian-beam parameters from Figure 11. It was 
found that the ratio of powers with either focus is the same within experimental error (±4%), 
indicating that both the hyperhemispherical focus and the elliptical focus will match equally 
well to an appropriately designed gaussian-beam system. Similar measurements were done 
on a log-periodic antenna from 90-250GHz. The results are similar to those presented in this 
paper and have been submitted for publication in IRMMW (May 92). 

IV. ACKNOWLEDGEMENTS 

This work was supported by the NASA/Center for Space Terahertz Technology at the Uni- 
versity of Michigan. 



Third International Symposium on Space Terahertz Technology Page 387 



APPENDIX 

The field representation of a Gaussian beam is of the form: EGauss(^) = t exp - ^^ ^ exp* 7 ^/^ 2 
The coupling efficiency between an antenna pattern and a gaussian beam is calculated using 
the formula [12,13]: 

| //[g co • F(fl, (j>f) exp-tW exp^ g /*> 2 sin fldfld<ft| 2 
VGauss - jj | F ^ ^ | 2 gin mA(j) jjexp-a («/*)» sin eded(j> 

where F(d, (f>) is the far-field pattern of the antenna, and e co is the co-pol unit vector. The 
value 0o controls the amplitude term and 0i controls the phasing term. These values are 
varied to optimize the coupling efficiency. 

REFERENCES 



[1] Born and Wolf, Principles of Optics , Permagon Press, New York, 1959, pp. 252-252. 

[2] D.B. Rutledge, D.P. Neikirk and D.P. Kasilingam, "Integrated Circuit Antennas," Infrared 
and Millimeter-Waves, Vol. 10, K.J. Button, Ed., Academic Press, New York, 1983, pp. 1- 
90. 

[3] D.B. Rutledge and M. Muha, "Imaging antenna arrays," IEEE Trans. Antennas Propa- 
gat., Vol. AP-30, 1982, pp.535-540. 

[4] J. Zmuidzinas, "Quasi-optical slot antenna SIS mixers," IEEE Trans, on Microwave 
Theory Tech., accepted for publication Jan. 1992. Also presented at the 2nd Int. Symp. on 
Space Terahertz Technology, CA, March 1991. 

[5] A. Skalare, Th. de Graauw, and H. van de Stadt,"A Planar Dipole Array Antenna with 
an Elliptical Lens," Microwave and Optical Tech. Lett., Vol. 4, No. 1, 1991, pp. 9-12. Also, 
"Millimeter and Submillimeter Studies of Planar Antennas," First Int. Symp. on Space 
Terahertz Technology, Ann Arbor, MI, March 1990, pp. 235-255. 

[6] C.J. Adler, C.R. Brewitt-Taylor, R.J. Davis, M. Dixon, R.D. Hodges, L.D. Irving, H.D. 
Rees, J. Warner, and A.R. Webb, "Microwave and Millimeter- Wave Staring Array Technol- 
ogy," IEEE MTT-S Int. Microw. Symp. Digest, June 1991, pp. 1249-1252. 

[7] T.H. Buttgenbach, "A Fixed Tuned Broadband Matching Structure for Submillimeter 
SIS Receivers," presented at the Third Int. Symp. on Space Terahertz Technology, Ann 
Arbor, MI, March 1992. 

[8] A.E. Siegman, Lasers, University Science Books, New York, 1986. 

[9] R.E. Collin, Antennas and Radiowave Propagation, McGraw-Hill, New York, 1985, pp. 
190-199. 



Page 388 

e r =11.7 



Third International Symposium on Space Terahertz Technology 



e r =4.0 



e r =2.3 






Figure 1: The synthesis of an elliptical lens from a hyperhemispherical lens and planar 
wafers. The extended hemisphere is a very good approximation to an elliptical lens at high 
dielectric constants. 



: ; V ^ 








i 

< 


•28X air ; 

* 

r - 








^_i 














.16 ?vair 









u i 












i 

c 




' ' '■■■ ' >\ 




j 




l a = 


= 0.76A m / \\ 








r 


d, 


= 0.30A m / 


- 


■ rl-plone _' 
- 15 -pi a nc 








/ 








-°i 




V 






'^ 


L 




* 




J 


c 


-4 






\ 


J 

J 
— i 








'',' 






CO 


i 




'/ 


\ 


V i 


O 
o 


i 

L 
-15 L 




//: 




■\> i 


> 








' \ ~i 


—J 


i" 








J 

i 


™ 


i 




, / 




j 








i 








-20 f- 


V 




/J 




l 








■ ■ ; ^ 
















-90 




-60 -30 30 
Angle (degrees) 




60 90 



Figure 2: The double-clot antenna (left) and its radiation patterns into a silicon (e=11.7) 
dielectric (right). 



Third International Symposium on Space Terahertz Technology 



Page 389 



Aperture 
Plane 



Critical 
Angle 



Silicon 
/ Wafers 




Silicon 
Wafer 

Feed Antenna 



Hyperhemispherical — ' | ' — Elliptical 

Intermediate 



Figure 3: The ray-tracing method. Note the three focus positions that are achieved by 
adding high-resistivity silicon wafers. 






o 

4) 
> 

CO 
0) 



-10 



-15 



-20 



-25 



I i. 



. aw 



E-plane j 

H-plane j 

— — 45-plane^ 






-30 -20 -10 10 

Angle (degrees) 



20 30 



> 



•10 



-20 



-30 



Gain=28.6<lB 



* ri I 






u 



-40 






E-plane 
H-plane 




-60 -40 -20 20 40 60 

Angle (degrees) 



Figure 4: Measured patterns at the elliptical focus at 246GHz. The patterns are diffraction- 
limited by the size of the aperture 



Page 390 



Third International Symposium on Space Terahertz Technology 



X3 



CO 
O 

> 



OS 




-20 -10 10 20 

Angle (degrees) 



.5 _10 r 

eo r 

C I"" 

F 

> -15 r 



-20 



-25 




H-Exper. 
- - H-Theory ' 



-30 -20 -10 10 

Angle (degrees) 



1 
1 



20 30 



Figure 5: Comparison of theory vs. experiment for the elliptical focus. The critical angle 
limits the size of the aperture, resulting in wider theoretical patterns. 



m 
-a 



c - 



a 
O 

> 



0) 

OS 




-30 -20 -10 10 

Angle (degrees) 



20 30 



Gain = 29.4dB 



TV ' ' ' ~ 
v E-plane 

i H-plane 

— — 45-plane. 



/ \ 



-10 h 



> -15 



-20 






.1' ' . \ 



J 



-30 -20 -10 10 20 30 

Angle (degrees) 



Figure 6: Measured patterns at the elliptical focus at 222GHz (left) and 270GHz (right). 



Third International Symposium on Space Terahertz Technology 



Page 391 



s 



— -■ 1 ' 1 r 

Gain=25.7dB 



25 I 1 L 



T ] I | I 

E-plane 

H-plane 

— — 45-plane 




' ' ' ' 



-30 -20 -10 10 20 30 

Angle (degrees) 



Figure 7: Measured patterns at the intermediate focus position at 246GHz. 



-5 - 



CO 



■rH 

efl 
O 

4) 

> -15 






-20 - 



-25 L 
-30 



1 1 1 1 1 




r 

I 

/ 


\ 

\ 

\ 
\ 




1 ; 

- E-Exper. 

- E-Theor\ 


-i 


- 


I'- 
ll 
// 






\ 

V 




- 


- 


II 






V 




- ! 




l\l 






A 


-: 


\ 


r: 






• 


\ 

u 


i 


- 


1 








\ 


-; 


" I 1 


1 
1 

1 * 1 


1 




t ! 


I 


i 



-20 -10 10 

Angle (degrees) 



20 30 




-20 -10 10 

Angle (degrees) 



30 



Figure 8: Comparison of theory vs. experiment for the intermediate focus. 



Page 392 



Third International Symposium on Space Terahertz Technology 



j- Gain = 18.1dB // I 

i if I ^ 



c -10 

3 
o 

> -15 



I An ) 






, .'■/'■' , .' 
-20 L\S - i 

M 



-25 




-60 -40 -20 20 

Angle (degrees) 



40 60 



Figure 9: Measured patterns at the hyperhemispherical focus position at 246GHz. 









CO 

o 



•10 



4) 

> -15 






-20 



-25 



T <~ 




E-Exper. - 
E-Theorv - 



'. \ 



-30 -20 -10 10 

Angle (degrees) 



20 



■10' 



;\ 



• 15 U/ ; 
l/V ; 



-20 



r ,x/ 



r ' 



-25 



/ 



^H 



H-Exper. 
Theory 




-30 -20 -10 10 20 

Angle (degrees) 



30 



Figure 10: Comparison of theory vs. experiment for the hyperhemispherical focus. Notice 
the predicted multi-peak behaviour. 



Third International Symposium on Space Terahertz Technology 

Coupling to Gaussian Beams 



Page 393 



Focus Position 
(extension) 


Gain 


Gaussian Beam Parameters 


Matching 
Efficiency 


(amplitude) 


O^phase) 


Elliptical 
(.39 radius) 


28.6dB 


5.0 8 


- 


~ 79% 


Intermediate 
(.32 radius) 


25.7dB 


8.2° 


11.3° 


~ 83% 


Hyper- 
hemispherical 
(.25 radius) 


18.1dB 


13.3° 


13.5° 


~ 81% 



Gaussian Beam Electric Field: explX©/©^ 2 ] exp[j*7t*( 0/G^ 2 ] 



Max. Power Elliptical 
Max. Power Hyperhemispherical 



= 10dB 



Figure 11: Table of Gaussian beam parameters. 



Page 394 Third International Symposium on Space Terahertz Technology 

6 ^^ EMBEDDING IMPEDANCE APPROXIMATIONS 
J6 6V8 IN THE ANALYSIS OF SIS MIXERS 

A. R. Kerr and S.-K. Pan ^ ,v 

National Radio Astronomy Observatory* ^93'"^**^^ 
Charlottesville, VA 22903 

S. Withington 

Cavendish Laboratory, University of Cambridge 
Cambridge CB3 OHE, UK 

ABSTRACT 



Future nillimeter-wave radio astronomy instruments will use arrays of many 
SIS receivers, either as focal plane arrays on individual radio 
telescopes, or as individual receivers on the many antennas of radio 
interferometers. Such applications will require broadband integrated 
mixers without mechanical tuners. To produce such mixers, it will be 
necessary to improve present mixer design techniques, most of which use 
the three - frequency approximation to Tucker's quantum mixer theory. 

This paper examines the adequacy of three approximations to Tucker's 
theory: (i) the usual three -frequency approximation which assumes a 
sinusoidal LO voltage at the junction, and a short-circuit at all 
frequencies above the upper sideband, (ii) a five -frequency approximation 
which allows two LO voltage harmonics and five small-signal sidebands, and 
(iii) a quasi five -frequency approximation in which five small-signal 
sidebands are allowed, but the LO voltage is assumed sinusoidal. These 
are compared with a full harmonic -Newton solution of Tucker's equations, 
including eight LO harmonics and their corresponding sidebands, for 
realistic SIS mixer circuits. 

It is shown that the accuracy of the three approximations depends 
strongly on the value of uRgC for the SIS junctions used. For large a>R„C, 
all three approximations approach the eight -harmonic solution. For uRgC 
values in the range 0.5 to 10, the range of most practical interest, the 
quasi five -frequency approximation is a considerable improvement over the 
three -frequency approximation, and should be suitable for much design 
work. For the realistic SIS mixers considered here, the five - frequency 
approximation gives results very close to those of the eight -harmonic 
solution. 

Use of these approximations, where appropriate, considerably reduces 
the computational effort needed to analyze an SIS mixer, and allows the 
design and optimization of mixers using a personal computer. 



*The National Radio Astronomy Observatory is operated by Associated 
Universities, Inc. under cooperative agreement with the National Science 
Foundation. 



Third International Symposium on Space Terahertz Technology Page 395 



INTRODUCTION 

The superior sensitivity of SIS mixer receivers at millimeter 
wavelengths has been clearly demonstrated in recent years [1], and SIS 
receivers are now used at almost all the world's major millimeter-wave 
radio astronomy observatories. SIS mixers appear for the most part to be 
well described by Tucker's quantum theory [2,3], which predicts strong 
non-classical behavior in a resistive mixer with an extremely sharp I-V 
nonlinearity. Tucker's theory is normally applied in its three -frequency 
approximation, in which form it is moderately tractable analytically and 
allows optimization of designs by small computer. 

At present, the most sensitive SIS mixers have one or two mechanical 
waveguide tuners which allow the RF embedding impedance to be adjusted to 
suit the particular junction (or array of junctions). While mechanically 
tuned mixers are acceptable in applications requiring one or two 
receivers, plans for future radio astronomy instruments include arrays of 
many SIS receivers operating either all in the focal plane of a single 
radio telescope , or individually on the many antennas of a radio 
interferometer. For such applications, it is highly desirable to replace 
mechanically tuned mixers with broadband integrated mixers without 
mechanical tuners. To produce such mixers will require refinement of 
present mixer design techniques, most of which use Tucker's three- 
frequency approximation. 

This paper examines the adequacy of three approximations to Tucker's 
theory: (i) the usual three -frequency approximation, (ii) a five- 
frequency approximation, and (iii) a quasi five -frequency approximation. 
These are compared with a full harmonic -Newton solution [4,5] of Tucker's 
equations up to the eighth LO harmonic for realistic SIS mixer circuits. 
Use of the approximations greatly reduces the computational effort needed 
to optimize a mixer design. 



Page 396 Third International Symposium on Space Terahertz Technology 



THE APPROXIMATIONS 

In the so-called three- frequency approximation to Tucker's theory, 
it is assumed, as indicated in Fig. 1(a), that the embedding impedance 
seen by the junction is finite at the LO frequency w p , and at w u , w, and 
w , the upper and lower sideband and intermediate frequencies. At all 
higher frequencies the junction is short-circuited. This is likely to be 
a good approximation for junctions with large capacitance. The three- 
frequency approximation implies a sinusoidal LO voltage at the junction, 
for which case Tucker gives closed- form expressions for the coefficients 
Y t j of the mixer's 3 x 3 small-signal conversion matrix as functions of 
the pumping parameter a - eV p /#w p and DC bias voltage V . 

The quasi five- frequency approximation assumes the same sinusoidal 
LO voltage waveform as above. However, while the second LO harmonic 2w p 
is short-circuited at the junction, the second harmonic sidebands 2w p ± o> 
are not. This is depicted in Fig. 1(b). Closed-form expressions for the 
elements of the 5x5 small-signal conversion matrix are given by Tucker. 
While this situation is not easy to implement practically, it is not 
physically unrealizable, and is expected to be a better approximation to 
a real mixer than the simple three -frequency approximation. 

The full five-frequency approximation allows finite embedding 
impedances at all frequencies up to 2w p + w , but requires the junction to 
be short-circuited at all higher frequencies. This is depicted in Fig. 
1(c). The LO voltage at the junction contains a second harmonic 
component, and an iterative algorithm must be used to determine the LO 
voltage waveform and thence the elements of the 5x5 small-signal 
conversion matrix. 



Third International Symposium on Space Terahertz Technology p a g e 397 



SIMULATIONS 

The three approximations are investigated using hypothetical double 
sideband SIS mixers at 115 and 345 GHz, with the desired embedding 
impedances at the LO harmonics and small-signal sideband frequencies. In 
all the examples, the junction capacitance is tuned out at frequencies w, 
and w u by the source susceptance. The embedding impedance at frequencies 
w > o> p + w is either zero or that of the junction capacitance alone, 
depending on the particular example. The RF source and IF load 
conductances are assumed equal. These assumptions are consistent with 
realistic low noise mixer designs [1] with low IF (u> « u> p ) . 

The I-V curve used in the examples is that of the 4 -junction array 
of Nb/Al-Al 2 3 /Nb junctions used in [1], and is shown in Fig. 2. The 
theoretical equivalence between a series array of junctions and a single 
junction is shown in [6] . 

Five values of w p R N C are used: 0.5, 1, 2,4, and 50. In all cases, 
the pumping parameter a - eVj/ftWp = 1.2, where V 1 is the amplitude of the 
fundamental component of the LO voltage at the junction. The junction is. 
voltage biased at the center of the first photon step below the gap 
voltage; i.e., V = V gap - hu/2e. 

The mixer's (equivalent input) noise temperature, (transducer) 
conversion gain, input return loss, and output impedance are computed for 
each case as functions of Rrf/Rn, where Rgp is the reciprocal of the RF 
source conductance at the signal frequency and R N is the normal resistance 
of the junction (or array of junctions) . For comparison, the results of 
the full eight-harmonic analysis are shown on each graph. 



Page 398 Third International Symposium on Space Terahertz Technology 



RESULTS 

Full five- frequency approximation 

The results of the full five -frequency approximation are shown in 
Figs. 3(a) -(d) for 115 GHz and 345 GHz mixers. At 115 GHz it is clear 
that the only significant deviation from the eight -harmonic results is in 
the gain for the lowest value of o>R N C (0.5), and that is only a small 
fraction of a decibel. At 345 GHz there is almost no difference between 
the full five - frequency approximation and the eight -harmonic results. 

Quasi five- frequency approximation 

Figures 4(a) -(d) show the results of the quasi five -frequency 
analysis of the same mixers as above. At both 115 and 345 GHz, the gain 
is within a decibel of that computed using eight harmonics. The mixer 
noise temperatures agree closely for wR N C - 50 and are within about 10% for 
wR H C - 4, but for «R N C - 0.5 they differ by as much as 40%. The input 
return loss and output impedance show minor deviations from the eight- 
harmonic solution. 

Three- frequency approximation 

In the three -frequency case, the value of wR N C has no effect. This 
is because all relevant frequencies above the upper sideband are short- 
circuited, and we are assuming the junction capacitance is tuned out at 
the signal and image frequencies. The three -frequency case is thus 
equivalent to the high-wR N C limit of any of the other cases we have 
considered. The results of Figs. 3 or 4 with wR„C ■= 50 are, in fact, 
indistinguishable from those of the three -frequency approximation. 



Third International Symposium on Space Terahertz Technology Page 399 



DISCUSSION 

In an earlier paper [1] we put forward design criteria for SIS 
mixers for millimeter-wave radio astronomy applications. They should have 
low noise, conversion gain near unity (to avoid saturation), and a 
reasonable RF input match. For wide frequency coverage, only double - 
sideband operation is considered. It was found that these conditions 
could be met if the mixer operated with equal RF source and IF load 
conductances , and that the output impedance of the mixer was then large 
(i.e., the mixer operates as a current source). 

The examples here likewise assume equal RF source and IF load 
conductances, and examine the mixer performance as a function of Rrf/Rh (Rrf 
is the reciprocal of the RF source conductance, and R N is the normal 
resistance of the junction or array of junctions). As expected, the mixer 
noise temperature exhibits a broad minimum as Rrf/Rh is varied (Fig. 3). 
The (transducer) gain exhibits no minimum, but increases with Rrf/Rn- 

At higher frequencies, the minimum noise temperature occurs at 
larger values of R^/R^. This has an important implication for SIS mixer 
design: If the same mixer circuit is scaled for use at two different 
frequencies (i.e., the embedding impedances are the same for the two 
designs), then the normal resistance of the junction (or array) should be 
smaller for the higher -frequency mixer. This is discussed in more detail 
in [1]. 

In designing a broadband tunerless SIS mixer, the choice of the 
value of wR N C is primarily a tradeoff between noise temperature and useable 
frequency range; large values of C will obviously limit frequency 
coverage, while too small an wR N C degrades the noise temperature as well 
as being more difficult to achieve without sacrificing junction quality. 
It is important, therefore, for the designer to have available a method of 
mixer analysis which reflects the effect of wR N C with sufficient accuracy. 
From the results in Figs. 3 and 4, it is clear that the quasi five- 



Page 400 Third International Symposium on Space Terahertz Technology 



frequency approximation describes the mixer performance quite well for 
wR N C > 4. For 0.5 < wR N C < 4, the gain, input return loss, and output 
impedance are well enough predicted by the quasi five -frequency 
approximation for most mixer design work, but the mixer noise temperature 
can be too high by as much as 40% . Figure 5 shows the dependence of the 
minimum value of T M (with respect to Rrf/Rjj) on wR N C for each of the methods 
described here . 

It may seem odd that the quasi five -frequency approximation should 
give quite accurate results for all the mixer parameters except the noise 
temperature. This is explained by considering the origin and nature of 
the noise at the output port of the mixer. This noise originates as shot 
noise from the current flowing in the SIS junction. The action of the 
local oscillator is, in effect, to amplitude modulate the shot noise 
produced by the DC current flowing in the junction, thereby generating 
correlated components at all the sideband frequencies nw p ± w , n - 0, 

1 etc. The mixing action of the time-varying junction conductance 

converts all these sideband components to the IF, preserving their 
correlation. The relative phase of each correlated IF component depends 
on the embedding impedance (including junction capacitance) at its 
sideband of origin, and on the phases of the harmonics in the LO waveform 
at the junction. The minimum noise occurs when these correlated IF noise 
components are phased so as to cancel to the greatest degree. (It is in 
this way that a classical diode mixer can, in principle, have zero noise 
temperature while operating with 3 dB conversion loss and many milliamps 
of rectified current.) The fact that, in low-noise operation, the SIS 
mixer has several quite large but strongly correlated output noise 
components which largely cancel one another, explains the sensitivity of 
the noise temperature results to the small changes in LO waveform between 
the quasi five -frequency and full five - frequency approximations. (A more 
detailed discussion of noise conversion in mixers, and additional 
references are given in [7].) 



Third International Symposium on Space Terahertz Technology Page 401 



REFERENCES 

[1] A. R. Kerr and S.-K. Pan, "Some recent developments in the design of 
SIS mixers," Int. J. Infrared Millimeter Waves, vol. 11, no. 10, 
Oct. 1990. 

[2] J. R. Tucker, "Quantum limited detection in tunnel junction mixers," 
IEEE J. of Quantum Electron., vol. QE-15, no. 11, pp. 1234-1258, 
Nov. 1979. 

[3] J. R. Tucker and M. J. Feldman, "Quantum detection at millimeter 
wavelengths," .Rev. Mod. Phys . , vol. 57, no. 4, pp. 1055-1113, Oct. 
1985. 

[4] C.-Y. E. Tong and R. Blundell, "Simulation of superconducting 
quasiparticle mixer using a five-port model," IEEE Trans. Microwave 
Theory Tech., vol. MTT-38, no. 10, pp. 1391-1398, Oct. 1990. 

[5] S. Withington and P. Kennedy, "Numerical procedure for simulating 
the large-signal quantum behavior of superconducting tunnel -junction 
circuits,"; Proc. IEE, part G, vol. 138, no. 1, pp. 70-76, Feb. 1991. 

[6] M. J. Feldman and S. Rudner, "Mixing with SIS arrays," Reviews of 
Infrared & Millimeter Waves, (Plenum, New York), vol. 1, p. 47-75, 
1983. 

[7] D. N. Held and A. R. Kerr, "Conversion loss and noise of microwave 
and millimeter-wave mixers: Part 1 - Theory," IEEE Trans. Microwave 
Theory Tech. , vol. MTT-26, pp. 49-55, Feb. 1978. 



Page 402 



Third International Symposium on Space Terahertz Technology 











- 3w p 


Junction 








short- 








circuited 
















- 2u p 












■ 










USB Up + 


«o 




- Up LD 


LSB u p - 


Uq 




— - u IF 








■4 


J- 



f 



Junction 
short- 
circuited 



not shorted 

shorted 

not shorted 



USB Up +• wo — 

LSB Up - Ug — — 



3u r 



2u c 



u p LD 



— u IF 



t j 

Junction „ 

short- -- 3u 
circuited p 

__L_„~____ 


2u p + uo — 




2u p - u — - 


- 2u p 


USB Up +■ uo — ■■ 






- u p LD 


LSB Up - uo — - 






— u IF 


-a- 



3-FREQ. APPRDX. 
Co.) 



QUASI 5-FREQ. APPRDX. 
<b) 



5-FREQ. APPRDX. 



Fig. 1 . Embedding impedance diagram indicating which frequencies are short-circuited for the three 
approximations: (a) three-frequency, (b) quasi five-frequency, and (c) full five-frequency. 



200 




HY430C5K11 
4 junctions 


4.2 


K 


/ 


I jjA 




R „* = 7 2 oh 


ns 




| 


100 


\ 






J 












. 


N 




5 


V 


10 
nV 


15 



Fig. 2. I-V curve used in the simulations. This curve is for a four-junction array of Hypres 
Nb/Al-Al 2 3 /Nb junctions at 4.2K, as used in [1]. 



Third International Symposium on Space Terahertz Technology 



Page 403 




8 Harmonics 
FuH 5-freq. 



CD 
(/) 
CO 

CD 

~o 

C 

"o 
o 



0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 

Rrf/Rn 
Fig. 3(a). Noise temperature and conversion gain as functions of Rrf/R n for a 1 15 GHz SIS mixer, 
computed using the full five-frequency approximation (solid line) and with eight LO harmonics 
(broken line). Results are shown for wR N C = 0.5, 1, 2, 4, and 50. 



20 



15 



CO 

■o 



w 

O 10 



c 

3 
^> 5 



3 

a 

c 



o- 



-5 




8 Harmonics 
Full 5-freq. 



0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 

Rrf/Rn 
Fig. 3(b). Input return loss and output impedance as functions of R RF /R N for a 1 15 GHz SIS mixer, 
computed using the full five-frequency approximation (solid line) and with eight LO harmonics 
(broken line). The output resistance is normalized to the IF load resistance R IF . Results are shown 
for wR N C = 0.5, 1, 2, 4, and 50. 



Page 404 



Third International Symposium on Space Terahertz Technology 




8 Harmonics 
Full 5-freq. 



m 
in 
</) 

m 
■o 

c 
'a 
o 



-10 



Fig. 3(c). Noise temperature and conversion gain as functions of Rrf/Rm for a 345 GHz SIS mixer, 
computed using the full five-frequency approximation (solid line) and with eight LO harmonics 
(broken lines - almost obscured by the solid lines). Results are shown for wR N C = 0.5, 1 , 2, 4, and 
50. 




8 Harmonics 
Full. 5-freq. 



or 



3 
O 



Fig. 3(d). Input return loss and output impedance as functions of Rrf/R n for a 345 GHz SIS mixer, 
computed using the full five-frequency approximation (solid line) and with eight LO harmonics 
(broken lines - almost obscured by the solid lines). The output resistance is normalized to the IF 
load resistance R, F . Results are shown for «R N C = 0.5, 1,2,4, and 50. 



Third International Symposium on Space Terahertz Technology 



Page 405 




8 Harmonics 
Quasi 5-freq. 



CD 
(A 
(/) 

CD 

■o 

C 

'o 
o 



0.0 0.2 0.4 



0.8 1.0 1.2 1.4 

Rrf/Rn 



-10 



Fig. 4(a). Noise temperature and conversion gain as functions of R RF /R N for a 115 GHz SIS mixer, 
computed using the quasi five-frequency approximation (solid line) and with eight LO harmonics 
(broken line). Results are shown for wR N C = 0.5, 1 , 2, 4, and 50. 




8 Harmonics 
Quasi 5-freq. 



0C 



3 
O 

cc 



0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 

Rrf/Rn 

Fig. 4(b). Input return loss and output impedance as functions of R„f/R n for a 115 GHz SIS mixer, 
computed using the quasi five-frequency approximation (solid line) and with eight LO harmonics 
(broken line). The output resistance is normalized to the IF load resistance R IF . Results are shown 
for wR N C = 0.5, 1 , 2, 4, and 50. 



Page 406 



Third International Symposium on Space Terahertz Technology 




8 Harmonics 
Quasi 5-freq. 



CD 
CD 

■o 

c 
o 
o 



Fig. 4(c). Noise temperature and conversion gain as functions of R RF /R N for a 345 GHz SIS mixer, 
computed using the quasi five-frequency approximation (solid line) and with eight LO harmonics 
(broken line). Results are shown for wR N C = 0.5, 1 , 2, 4, and 50. 




8 Harmonics 
Quasi 5-freq. 



or 



3 
O 

or 



Fig. 4(d). Input return loss and output impedance as functions of Rrf/R n for a 345 GHz SIS mixer, 
computed using the quasi five-frequency approximation (solid line) and with eight LO harmonics 
(broken line). The output resistance is normalized to the IF load resistance R IF . Results are shown 
for wR N C = 0.5, 1, 2, 4, and 50. 



Third International Symposium on Space Terahertz Technology 



Page 407 




100 



u>RnC 



Fig. 5. Minimum mixer noise temperature as a function of wR N C for: (i) the full eight-harmonic and 
full 5-frequency solutions (broken line), (ii) the quasi 5-frequency approximation (solid line), and (iii) 
the 3-frequency approximation (horizontal lines). 




Page 408 Third International Symposium on Space Terahertz Technology 



SUB MICRON AREA Nb/A10 x /Nb TUNNEL JUNCTIONS FOR SUBMM MIXER 

APPLICATIONS 

H.G. LeDuc, B. Bumble, S.R. Cypher, A. J. Judas*, and J A.. Stern 

Center for Space Microelectronics M Q Q C) ry ty „ 
Jet Propulsion Laboratory i\ & O m /£{ • 6 1 

California Institute of Technology 
Pasadena, CA 91109 

*Present Address: Stanford University 
Palo Alto, CA 

Abstract 



In this paper, we report on a fabrication process developed for submicron area tunnel 
junctions. We have fabricated Nb/A10 x /Nb tunnel junctions with areas down to 0.1 nm* 
using these techniques. The devices have shown excellent performance in receiver systems 
up to 500 GHz and are currently in use in radio astronomy observatories at 115, 230, and 
500 GHz. 

The junctions are fabricated using a variant of the self-aligned-liftoff trilayer process* with 
modifications for electron beam lithographic patterning of junction areas. In brief, the 
technique involves patterning submicron holes in PMMA using electron beam lithography. 
The negative of this pattern is formed by thermal deposition and liftoff of chromium metal 
using this PMMA stencil. The chromium pattern is transferred to an underlying polyimide 
film using oxygen RIE. Junctions are formed by RIE using a gas mixture containing 
CQ2F2 and electrically isolated with thermally evaporated silicon monoxide. Contact 
wiring and coupling/tuning structures are patterned by RIE. 

Introduction 

SIS tunnel junctions can be modeled as a nonlinear resistor in parallel with a shunt 
capacitor. A good figure-of-merit of the high frequency performance of these devices is 



Third International Symposium on Space Terahertz Technology p a o e 409 

the ratio of the capacitive reactance to the real resistance (coRC). The RC product, for 
SIS tunnel junctions, is determined by the tunnel barrier thickness and is independent of 
the device area. The junction area is chosen to provide the best impedance match to the 
mixer embedding circuit and is usually a compromise between minimizing the capacitance 
while maintaining a reasonable real impedance. In the best case, the embedding circuit 
can tune out the capacitance and the junction area is chosen to make the rf-resistance 
match the real part of the embedding circuit impedance (approximately 50 - 100 CI). For 
small RC devices, the resistance-area product is small so that achieving the appropriate 
resistance using a single junction requires submicron areas. Series arrays or other novel 
coupling mechanisms may relieve the constraint on submicron areas, however, designing 
these elements may require a greater understanding of the high frequency characteristics 
of devices and materials than is currently available. We have chosen to use single junctions 
in the hope that the simplicity in understanding the high frequencies behavior of the 
mixers may outweigh the complexity associated with the fabrication of submicron devices. 
Since their development 2 , high quality Nb/AlOx/Nb tunnel junctions represent the only all 
refractory SIS technology in use in radio astronomy receiver systems. This is primarily due 
to their nearly ideal tunneling characteristics and physical robustness. In this paper we 
describe techniques for fabricating submicron devices. 

Tunnel Junction Fabrication 

The tunnel junction fabrication process is similar to the self-aligned-liftoff process used 
to fabricate refractory tunnel junctions employing optical lithography 1 ' 3 . The primary 
difference arises from the need to use higher resolution lithography in the tunnel junction 
patterning and to maintain this resolution throughout the fabrication process. The process 
steps are shown schematically in figure 1. 



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Third International Symposium on Space Terahertz Technology 



Chromium 




PMMA 

Polyfmide 

Nb 

Insulator 

Nb 





Chromium 
Polyfmide 

i Nb 
Insulator 

2 Nb 



m 




Figure 1. Submicron fabrication process schematic, (a) After trilayer deposition , wafers are spin coated 
with 400-600 nm of polyimide and 120 nm PMMA. (b) Chromium metal is thermally deposited, (c) Oxygen 
RIE of polyimide. (d) RIE of Nb counter electrode in CCI2F2+CF4+O2 gas mixture, (e) Deposition of 
SiO, lift-off, and wire electrode deposition and patterning. 



a. Nb/AJO x /Nb Trilayer Deposition 



The Nb/A!O x /Nb trilayer is deposited in-situ in a high vacuum system (base pressure 1.3 x 
10 "7 Pa) by magnetron sputtering. The substrates are oxidized silicon or quartz and are 
heat sunk to a thermal mass but not actively cooled during deposition. The large scale 
features of the trilayer are formed by lift-off using AZ5214 photoresist (AZ Hoechst) and 
image reversal. The Nb base and counter electrodes are approximately 160 nm and 120 
nm respectively. The barrier is formed by depositing 6-10 nm of aluminum followed by an 
in-situ oxidation in an argon/oxygen gas mixture in a manner similar to that described by 
Morohashi et al. . During this step the total process gas pressure is maintained constant by 
throttling the vacuum pump. A dc-plasma is formed during the oxidation process by 
applying approximately -500V to an aluminum ring placed in the system. This plasma has 



Third International Symposium on Space Terahertz Technology Page 411 

been found to reduce oxidation times, but does not effect the quality of the barrier. After 
the Nb counter electrode deposition, 30 nm of gold is deposited on the trilayer to act as a 
contact layer. 

b. Junction Patterning 

The etch mask used to form the tunnel junction is patterned by electron beam (e-beam) 
lithography using a JEOL JBX-5 lithography system with a minimum spot size of 8 nm. 
The lithographic stencil must be robust enough to withstand Reactive Ion Etching (RIE) 
and provide a means to subsequently lift-off the SiO isolation layer. The high resolution e- 
beam resist, PMMA, is not suitable as the final RIE mask because it lacks the required 
etch resistance. Techniques have been developed which transfer the e-beam written 
pattern into polyimide while maintaining the required resolution 4 . The wafer is spin 
coated with a polyimide 5 film approximately 400 - 600 nm thick. Following a hot plate 
bake to drive the solvents from the polyimide, the wafer is spin coated with 120 nm of 
PMMA. It is then exposed in the e-beam lithography system to form holes in the PMMA 
film with the required junction dimensions. Chromium metal is thermally evaporated onto 
this stencil and the PMMA is removed in acetone, leaving metal where there were holes 
(polyimide is not soluble in acetone). The resulting pattern is etched in a parallel plate 
RIE system using oxygen gas to remove polyimide from areas of the wafer not protected by 
chromium. The RIE of polyimide is highly anisotropic, however, it is sensitive to surface 
contamination such as dust or material resputtered from the electrodes of the etcher and 
care must be taken to provide a clean environment for this process step. An SEM 
micrograph of an etch test pattern is shown in figure 2. The square etch stencils consisting 
of Cr(30nm) on Polyimide(550nm) have dimensions of 1.5, 1.0, 0.5, and 0.25 urn on a side. 
The minimum area is .06p.m2 . 



Page 412 



Third International Symposium on Space Terahertz Technology 




Figure 2. Test patterns etched in polyimide using oxygen RIE . The smallest 
features are 0.06u,m2 . 



c. Junction Etch 



The tunnel junction is formed using RIE by first etching the gold contact layer and then 
the Nb counter electrode. The gold is sputter etched using argon gas. Techniques for 
aniso tropically etching Nb had to be developed. An etch profile for a submicron line 
patterned in an Nb film using a standard etch process (CF4+20% O2 ,4 Pa pressure, and 
0.27 W/cm^ power density ) is shown in figure 3. The isotropic component of this etch 
mixture is clearly too large to be used in the fabrication of submicron devices. Anisotropy 
occurs in RIE when the etch mechanism requires predominantly normal incident ion 
impact energy to proceed 6 . Etching of Nb in CF4/O2 , however, occurs via a spontaneous 
rather than ion assisted reaction of fluorine and fluorine radicals with Nb. We have found 
a technique which achieves the required anisotropy. Etching with a gas mixture containing 
CQ2F2 produces very good etch anisotropy, which may be attributed to the a nonvolatile 
NbCl x product which forms on the sidewalls. Figure 4 shows the etch rate of Nb and NbN 
using mixtures of CCI2F2+CF4+O2 • For these measurements, the total pressure was 4 



Third International Symposium on Space Terahertz Technology 



Page 413 




Figure 3. Submicron Nb lines etched by RIE using CF4+O2. The large undercut of the Nb 
line below the 0.4(im chromium etch stencil is evident. 

Pa, the power density was 0.27 W/cm^ and the oxygen flow was constant at 2 seem, while 
the CQ2F2/CF4 ratio was varied. The etch is highly anisotropic for mixtures containing 
greater than 60% CQ2F2 in CQ2F2+CF4. Mixtures rich CF4 exhibited isotropic etching. 
The region with approximately 20% to 50% CCI2F2 content was characterized by low etch 
rates and polymer formation. Shown in figure 5 is the etch profile of Nb achieved using 
62% CQ2F2 in (CCI2F2+CF4) and similar sample etched in CF4+O2. Structures etched 
in the CCI2F2 gas mixture show very little undercut while CF4+O2 produced a large 
undercut. 

d. Electrical Isolation 



Following the etch the counter electrode to form the junctions, a electrical isolation layer 
of SiO is deposited with the etch mask in place . The SiO is thermally deposited from a 
baffled source. To achieve good edge coverage, the samples are placed at a fixed angle 



Page 414 



Third International Symposium on Space Terahertz Technology 



0.2 



0.4 



0.6 



Nb 
NbN 




0.8 



Figure 4. RIE etch rate for Nb and NbN as a function of gas composition. The 
etch gas consists of 85%(x CCl2F2+(l-x)CF4)+15%02. 

relative to SiO flux and rotated during the deposition. Flux angles for normal incidence to 
approximately 60 degrees have been evaluated. Angles of 5-15 degrees have been found to 
provide a good compromise between side wall coverage and clean lift-off. SiO film 
thicknesses are typically 150-250 nanometers depending on the application. The polyimide 
and SiO are removed from the junction areas using dichloromethane solvent. A short RIE 
etch in oxygen is used to remove polyimide residues after the lift-off step. 



e. Contact Wiring 
Mixer elements are completed by depositing 250-350 nm of Nb by magnetron sputtering. 
The wire layer is patterned lithographically and etched using a RIE process similar to the 
one used for the junction etch. A typical current-voltage characteristic for a tunnel junction 
fabricated by this process is shown in figure 6. This device is 0.25 \xxa^ in area and has a 
critical current density of 7.7 kA/cm^. 



Third International Symposium on Space Terahertz Technology Page 415 



Summary/Conclusions 

In this paper, we have described techniques developed for the fabrication of submicron 
area tunnel junctions in refractory materials. The process described is applied specifically 
to the fabrication of Nb/A10 x /Nb tunnel junctions, however, much of the technology has 
also been used to fabricate NbN/MgO/NbN tunnel junctions 7 and is relevant to other 
submicron fabrication tasks. This process extends the self-aligned lift-off process used to 
fabricate refractory tunnel junctions using optical lithography. The primary new features 
are the use of electron beam lithography to form a submicron pattern in PMMA and the 
transfer of this pattern into chromium by lift-off. The chromium pattern is transferred into 
polyimide using oxygen RIE and the resulting Cr/polyimide is used to etch the trilayer 
counter electrode using a highly anisotropic RIE gas mixture containing CQ2F2. 
Nb/A10 x /Nb tunnel junctions with areas down to 0.1 \im^ have been fabricated using these 
techniques. Mixer elements have been fabricated using this process for both wave 
guide 8 » 9 » 10 and quasi optically coupled 11 * 12 ' 13 receiver systems. In wave guide receiver 
systems with operating frequencies up to 500 GHz, the capacitance associated with the 
submicron area Nb/A10 x /Nb devices is small enough so that the mixer block rf-embedding 
circuit provides enough tuning to achieve excellent performance (receiver noise 
temperatures, Tr(DSB) = 180K at 485 GHz) 14 without integrated tuning structures. In 
principle junction areas can be scaled down further, however, in order to do so the junction 
relaxation times must also be scaled down so that the real part of the junction impedance in 
the correct range. The junction relaxation time (RC) is determined by the insulator barrier 
thickness, with thinner barriers producing smaller RCs. The limit for a given insulator 
barrier is determined by the thinnest barrier that can be achieved while maintaining 
suitable junction characteristics. It has been our experience with Nb/A10 x /Nb tunnel 
junctions, that the I-V characteristics degrade significantly for critical current densities of 



Page 416 Third International Symposium on Space Terahertz Technology 

greater than 15kA/cm2 (RA= 12 Q u.m2). For junctions with this current density, a 100 Q 
junction has an area of = 0.12 um^ 

Acknowledgements 
The research described in this paper was performed by the Center for Space 
Microelectronics Technology, Jet Propulsion Laboratory , California Institute of 
Technology, and was jointly sponsored by the Strategic Defense Initiative Organization / 
Innovative Science and Technology Office and the National Aeronautics and Space 
Administration / the Office of Aeronautics and Space Technology. We would also like to 
acknowledge P.D. Maker and R.E. Muller for the excellent electron beam lithography 
support and technical discussions. 

References: 



1 A. Shoji, F. Shinoki, S. Kosaka, M. Aoyagi, and H. Hayakawa, " New Fabrication Process for 
Josephson Tunnel Junctions with (Nobium Nitride, Niobium) Double-Layered Electrodes", Appl. Phys. 
Lett., 41,1097, (1982). 

2 M. Gurvitch, M.A. Washington, and H.A. Huggins, "High Quality Refractory Tunnel Junctions 
Utilizing Thin Aluminium Layers", Appl. Phys. Lett., 42, 472 (1983). 

3 S. Morohashi, F. Shinoki, A. Shoji, M. Aoyagi, and H. Hayakawa, "High Quality NblAl-AlOxINb 
Josephson Junction", Appl. Phys. Lett. 46, 1179, (1985). 

4 D.M. Byrne, AJ. Brouns, F.C. Case, R.C. Tiberio, B.L. Whitehead, and E.D. Wolf, "Infrared Mesh 
Filters Fabricated by Electron-Beam Lithography", J. Vac. Sci. Technol., B3, 268 (1985) and references 
within: M. Hatzakis, B.J. Canavello, and J.M. Shaw, IBM J. Res. Dev., 24, 452 (1980). 

5 Olin Ciba-Geigy, Probimide 200 series. 

6 See for instance: J.W. Coburn, Plasma Etching and Reactive Ion Etching. AVS Monograph Series, 
Ed. N. Rey Whetten. 

7 J.A. Stern, H.G. LeDuc, and AJ. Judas, "Fabrication and Characterization of High Current-Density, 
Submicron, NbN/MgO/NbN Tunnel Junctions", this conference. 

8 RES. Javadi, W.R. McGrath, S.R. Cypher, B.D. Hunt, and H.G. LeDuc, Digest 15th Int. Conf. on 
IR and Millimeter Waves, p245, Orlando, FL (1990). 

9 J.W. Kooi, M. Chan, T.G. Phillips, B. Bumble, and H.G. LeDuc, "A Low Noise 230 GHz Heterodyne 



Third International Symposium on Space Terahertz Technology Page 417 



Receiver Employing .25\un^ Area NblAlOJNb Tunnel Junctions", IEEE Microwave Theory and 
Techniques Journal, to be published. 



10 



C.K. Walker, M. Chen, P.L. Shafer, H.G. LeDuc, J.E. Carlstrom, and T.G. Phillips, "A 492 GHz SIS 
Waveguide Receiver for Submillimeter Astronomy", Int. J. of IR and Millimeter Waves, to be 
published. 



11 T.H. Buttgenbach, H.G. LeDuc, P.D. Maker, and T.G. Phillips, "A Fixed Tuned Broadband Matching 
Structure for Submillimeter SIS Receivers", IEEE Trans. Appl. Superconductivity, to be published. 

12 P.A. Stimson, RJ. Dengler, P.H. Siegel, and H.G. LeDuc, "A Planar Quasi-Optical SIS Receiver for 
Array Applications", this conference. 

13 J. Zmuidzinas, H.G. LeDuc, and J.A. Stem, "Slot Antenna SIS Mixers for Submillimeter Wavelengths", 
this conference. 

14 Private communication, C.K. Walker. 



Page 418 



Third International Symposium on Space Terahertz Technology 
















Figure 5. SEM micrographs showing etch comparrisons between CF4+O2 (left) and 
CCl 2 F2+CF 4 +02(right). The RIE mask is Cr(30 nm)/Polyimide(550 nm) patterned by e-beam 
lithography and oxygen RIE. The Nb film (400 nm thick) etchs anisotropically in the CCI2F2 containing 
etch gas. 




1mV 



Figure 6. Typical tunneling current-voltage characteristic for a Nb/AlOx/Nb junction taken ar 4.2K. The 
junction area is 0.25 mm^ and the critical current density is 7.7 kA/cm^ . 



Third International Symposium on Space Terahertz Technology Page 419 

Noise in Josephson Effect Mixers and the RSJ Model 

R. Schoelkopf*, T. Phillips*, and J. Zmuidzinas* ^ " «* ^-X7 6 2 

Josephson effect mixers have previously been observed to display "excess" noise r J 
/ both in experiments with point contacts and in numerical simulations using the resistively 
shunted junction (RSJ) model. This excess noise causes the mixer noise temperature to be 
a factor of typically 20- 100 times^ the physical temperature of the device. Previously, this 
excess was ascribed to conversion from unwanted sidebands of the local oscillator and 
Josephson frequencies and their harmonics^. Our numerical modeling of the RSJ 
equations has led to a new understanding of the excess noise, which is simply due to the 
intrinsic Josephson oscillations of the device. In addition, we have extended the modeling 
to include the previously ignored case of finite device capacitance (i.e. RSJ capacitance 
parameter (3c ^ 0), which is more realistic for lithographically defined Josephson such as 
shunted tunnel junctions or SNS bridges. For some cases, this yields an improvement of a 
factor of two in noise temperature from the zero capacitance models. We will discuss the 
device parameters which optimize the mixer performance for frequencies approaching the 
characteristic frequency of the device, which is given by the Josephson frequency at the 
IcR n voltage (v = 2eIcRn/h). These modeling results predict good conversion efficiency 
and a noise temperature within a factor of a few of the physical temperature. Experiments 
are in progress to determine the accuracy of this modeling using a waveguide mixer at 
100 GHz with optimized, resistively shunted Nb tunnel junctions. If the modeling results 
are valid, they are particularly encouraging for mixers in the submillimeter regime, given 
the possibility of obtaining non-hysteretic Josephson devices with IcR n products in excess 
of a millivolt, using, for instance, high-T c SNS bridges. We discuss the modifications to 
the classical RSJ model which are necessary in the quantum regime (hv > kT), and 
conclude the Josephson mixers may attain noise temperatures less than ten times the 
quantum limit at high frequencies. 



*R. Schoelkopf, T. Phillips, and J. Zmuidzinas are with California Institute of 
Technology. 

MY. Taur, IEEE Transactions on Electron Devices, ED-27, p. 1921, 1980. 

t^K.K. Likharev, Dynamics of Josephson Junctions and Circuits, New York: 
Gordon and Breach, p. 423, 1986. 



Page 420 Third International Symposium on Space Terahertz Technology 

FABRICATION AND CHARACTERIZATION OF HIGH CURRENT-DENSITY, 
^ -y^p/L SUBMICRON, NbN/MgO/NbN TUNNEL JUNCTIONS 

/hoSer/ N93-27763 

Oy /. A. Stern " ** 

U ^ H. G. LeDuc 

A. J. Judas* 

Center for Space Microelectronics Technology 

Jet Propulsion Laboratory 

California Institute of Technology 

Pasadena, California 91109 

* Present Address: Stanford University 

Palo Alto, California 



Abstract 
At near-millimeter wavelengths, heterodyne receivers based on SIS tunnel junctions are the most 
sensitive available. However, in order to scale these results to submillimeter wavelengths, 
certain device properties should be scaled. The tunnel-junction's current density should be 
increased to reduce the RC product. The device's area should be reduced to efficiently couple 
power from the antenna to the mixer. Finally, the superconductor used should have a large 
energy gap to minimize RF losses. Most SIS mixers use Nb or Pb-alloy tunnel junctions; the 
gap frequency for these materials is approximately 725 GHz. Above the gap frequency, these 
materials exhibit losses similar to those in a normal metal. The gap frequency in NbN films is 
as-large-as 1440 GHz. Therefore, we have developed a process to fabricate small area (down 
to 0.13 n 2 ), high current density, NbN/MgO/NbN tunnel junctions. 



Third International Symposium on Space Terahertz Technology Page 421 

In this paper, we describe a process used to fabricate submicron NbN junctions. Low-leakage 
current-voltage (I-V) characteristics are achieved for current densities up to 40 kA/cm 2 . 
However, the quality of the I-V characteristics degrades significantly for higher current densities. 
Junction areas are patterned by lifting off a Cr stencil defined by electron beam lithography. 
This image is transferred to a polyimide stencil using reactive ion etching (RIE). The junctions 
are then etched and isolated using a self-aligned liftoff technique. The limitations of this 
technique and the quality of the resulting I-V characteristics will be discussed. 

There are several device and materials parameters which must be known to properly design 
mixer circuits. The optimal imbedding impedance for the mixer is determined by the 
capacitance and I-V characteristic of the tunnel junction. If microstrip line circuits are used to 
achieve this impedance, the magnetic penetration depth of the NbN must be known to calculate 
the microstrip line impedance and propagation velocity. We have measured the junction 
capacitance versus current density and microstrip line inductance using superconducting quantum 
interference devices (SQUIDs). The propagation velocity was measured using long open-ended 
microstrip-lines connected to Josephson tunnel junctions. The magnetic penetration depth can 
be calculated from either the microstrip line inductance or the propagation velocity. The 
implications of these measurements will be discussed. 

Introduction 
There is a growing need for sensitive submillimeter (smm) wavelength detectors for both ground 
based and space based applications. Heterodyne receivers based on superconductor-insulator- 
superconductor (SIS) tunnel junctions are the most sensitive at millimeter and near-millimeter 



Page 422 Third International Symposium on Space Terahertz Technology 

wavelengths [1][2]. At wavelengths longer than 3 mm, the noise in these detectors has 
approached the quantum limit [3]. If these mixers designs could be redesigned for shorter 
wavelength operation, they would fill the need for smm wavelength mixers. For good 
performance to be achieved at smm wavelengths, a number of SIS tunnel junction properties 
must be adjusted. We have fabricated and characterized tunnel junctions which are suitable for 
smm use. 

The junction capacitance is the property which has the largest impact on smm wave mixers. An 
SIS tunnel junction can be modeled as a nonlinear tunneling element in parallel with a shunt 
capacitance. The product of the shunt capacitance (C) and the RF resistance of the junction 
(approximately the normal state resistance, RJ is one limit on the high frequency response of 
a tunnel junction, because the capacitance tends to shunt RF currents away from the nonlinear 
tunneling element. To reduce the R n C time constant, the tunnel junction barrier is made 
thinner; this increases the junction capacitance 

C=e e r A/d, 
where e is the dielectric constant of free space, e r is the relative dielectric constant of the 
barrier, A is the junction area and d is the barrier thickness, but it decreases the resistance more 
rapidly 



„ _ h 2 d-Q\p{4ird\/2m(f) /h} 



e 2 



■A\Jlm<\> 



where h is Plank's constant, m is the mass of the quasi-particle, <t> is the barrier height and e the 
charge of the quasiparticle. Because the barrier cannot be made thinner than one monolayer, 
the R,jC product is ultimately limited by the barrier material properties (e r , and the lattice 



Third International Symposium on Space Terahertz Technology Page 423 

constant). The R n C product is independent of junction area, but the junction resistance must be 
roughly 100 Q for it to couple efficiently to the antenna. Therefore, as the barrier is made 
thinner, the tunnel junction's area must also be reduced. At smm wavelengths, junctions with 
submicron areas are needed. Since the barrier thickness is difficult to measure, R„A products 
can be used as an area-independent measure of junction relaxation time. The product of the 
critical current density (J c ) and R„A is proportional to the energy gap, so the current density is 
also frequently used as a measure of the relaxation time. 

The embedding circuit of a mixer can be designed to resonate out the junction capacitance over 
a frequency band. In waveguide mixer mounts, this can be accomplished using a back short and 
E-plane tuner; however, at high frequencies these tuners do not work as well as at lower 
frequencies. Additionally, quasi-optical mixers do not have these tuning elements. Monolithic 
embedding circuits can also be fabricated to resonate out the junction capacitance 
[4] [5] [6]. Many of these circuits use superconducting microstrip lines. To accurately 
design these circuits, the propagation velocity and impedance of the microstrip line must be 
known. Both of these parameters depend critically on the magnetic penetration depth of the 
superconductor [7]. 

The energy gap of the superconductor (2A) limits the use of SIS mixers at smm wavelengths. 

* 

Above the gap frequency (2A/h), the superconductor has losses similar to those in a normal 
metal; therefore, any embedding circuits will have losses[8]. Embedding circuits could be made 
of a larger gap superconductor or a low-loss normal metal (such as gold or copper).. In this 
case, the SIS mixer will perform well up to twice the gap frequency. Most mixers currently use 



Page 424 Third International Symposium on Space Terahertz Technology 

niobium based tunnel junctions. The gap frequency of niobium is 725 GHz. Above this 
frequency, other superconductors will probably be needed. 

Based on the small gap in niobium, we have developed a process for fabricating junctions with 
NbN electrodes, which has a much larger gap frequency (up to 1440 GHz). The goal was to 
fabricate high current-density, submicron tunnel junctions with large energy gaps. The 
limitations of our process on both current density and junction area are discussed below. 
Additionally, we have measured several device properties necessary to design mixer circuits 
(NbN magnetic penetration depth, junction uniformity, yield and specific capacitance.) 

Junction Fabrication 
The tunnel junctions are fabricated by depositing the NbN/MgO/NbN trilayer over the entire 
substrate [9][10]. The NbN films are deposited by reactive DC-magnetron sputtering from a 
niobium target using a 5 cm diameter US inc. sputter gun in an argon and nitrogen atmosphere. 
The gas flow rates are 150 and 15 seem respectively and the total pressure with the plasma off 
is 14.4 mTorr. The gun is typically biased at -232 Volts and 600 mAmps. The substrate is not 
heated and is approximately 6 cm from the sputter gun. Deposition rates are 45 nm/min. The 
thickness of the base and counter electrode are 315, 135 nm respectively. The superconducting 
transition temperature of the films is usually between 14.5 and 15.0 K. 

The barrier layer is formed by depositing MgO and performing a pinhole cure in an oxygen 
plasma. MgO is deposited by RF sputtering from a US inc. 5 cm diameter source in a pure 
argon atmosphere. The Ar pressure is 10.0 mTorr and the power is 50 Watts. To promote 



Third International Symposium on Space Terahertz Technology Page 425 

uniform growth of the barrier, deposition occurs intermittently as the substrate is rotated past 
the source in a circular orbit. After the barrier deposition, the sample is exposed to an oxygen- 
plasma glow discharge for 1 minute at 100 mTorr. The plasma is maintained with a high purity 
aluminum ring below the substrate; the ring is biased at -400 Volts and 2.0 mA. Barrier 
deposition times vary between 5.5 and 6.7 min. for current densities between 40 and 10 kA/cm 2 . 

The trilayer is completed by covering the entire NbN/MgO/NbN structure with 30 nm of gold. 
The gold prevents the top surface of the junction from oxidizing in air. In some cases, there 
is a thin (40 nm) layer of Nb between the counter electrode and the gold. The purpose of the 
Nb interlayer will be discussed below. 

Tunnel junctions are fabricated using a submicron, self-aligned lift-off technique. This process 
is shown in figure 1 . The trilayer is patterned using RIE and a photoresist stencil. The etcher 
used is a Semi Group System 1000, and the electrode area is 730 cm 2 . The gold and MgO layers 
are etched using a straight argon sputter etch. The NbN layers are etched in a CF 4 -0 2 mixture. 
The conditions for this and other reactive ion etches are given in table 1. After the etch, the 
photoresist is stripped in organic solvents. 

The submicron tunnel junction stencil is formed in a polyimide layer [11]. The etched 
trilayer is coated in Ciba Geigy Probimide 286 and thinner 2:1 at 4000 rpm resulting in a 310 
nm polyimide layer. The polyimide is cured at 150° C on a hot plate for 15 minutes and flood 
exposed with 315 nm radiation. A KTI Chemicals Inc. 950K 2% PMMA layer is spun on top 
of the polyimide at 4000 rpm and baked at 150° C. A thin aluminum layer is evaporated on top 



Page 426 Third International Symposium on Space Terahertz Technology 

of the PMMA to prevent charging during the exposure. The sample is exposed in a JEOL model 
JBX5 electron beam lithography system and subsequently developed to reveal square holes in 
the PMMA. A thin (35 nm) chromium layer is thermally evaporated and lifted off to define 
the junction area. Next an AZ 5214-E stencil is patterned using image reversal [12] to reveal 
an open area around the chromium dot. This stencil acts to protect the majority of the trilayer 
from the junction etch. The polyimide layer is etched using an oxygen plasma. During the etch, 
a polyimide (Kapton) sheet is about 3 cm above the sample to protect against particles falling 
on the sample during the etch. The oxygen etch is very anisotropic and an over etch is 
normally used to insure that the trilayer surface is completely clean. 

The junction etch consists of an Ar sputter-etch of the gold and one of several NbN etches 
[13]. For micron size junctions, the CF 4 -0 2 mixture can be used, however this etch is isotropic 
and it will undercut NbN mesa. A straight CF 4 etch can be used for submicron junctions. This 
etch is largely anisotropic, because a fluorinated carbon layer forms on the sidewall of the 
junction as it etches; however, there is about 50 nm of undercut. A mixture of CCl 2 F2-CF 4 -0 2 
produces an anisotropic etch, however it reacts at the Au-NbN interface and produces a 
superconductor-normal metal-superconductor type I-V characteristic in series with the SIS I-V 
characteristic. This series weak link can be avoided by inserting a niobium layer between the 
gold and NbN layers and by rinsing the sample in water after the etch. The niobium buffer 
removes the series weak link which seams to occur at the NbN-Au interface Regardless of the 
etch used, a monitor sample is used to detect the endpoint of the junction etch. After the 
junction etch, a 1 minute argon sputter etch is done to improve the adhesion of the dielectric 
isolation layer. 



Third International Symposium on Space Terahertz Technology 



Page 427 



Table 1 . Summary of Reactive Ion Etch Conditions 



Material 


Etch 
Gases 


Flow Rates 
(seem) 


Pressure 
(mTorr) 


Power 
(Watts) 


Etch Rate 
(nm/min) 


Polyimide 


o 2 


20 


30 


130 


100 


Gold 


Ar 


20 


30 


118 


15 


NbN 


CF 4 ,0 2 


20,2 


30 


118 


65 


NbN 


CF 4 


20 


30 


118 


52 


NbN 


CC1 2 F 2 , 
CF 4 ,0 2 


23,7,2 


30 


118 


58 



The junction is electrically isolated by thermally evaporating 150 nm of Silicon monoxide on the 
sample. The wafer is rotated at a 5 to 15° angle during the evaporation to improve SiO 
coverage. The photoresist protection layer is then lifted off in acetone, followed by the 
polyimide in dichloromethane. The polyimide liftoff is done in an ultrasonic cleaner followed 
by a mechanical scrub with a q-tip to remove the SiO flags that form on the side- walls of the 
polyimide pillar. The liftoff yield, even for the smallest junctions, is excellent. Following the 
polyimide liftoff, an oxygen etch is done to remove any polyimide residue which will lead to a 
contact resistance. 

The NbN wiring layer is deposited on top of the electrically isolated junction. To avoid breaks 
at step edges, the wiring is typically 600 nm thick. The wiring is patterned and etched like the 
trilayer, although if small features are desired, one of the anisotropic etches is used. 



Results 



Page 428 



Third International Symposium on Space Terahertz Technology 



The quality of tunnel junction I-V characteristics depends strongly on the critical current density. 
As the current density increases, the ratio of the subgap resistance (measured at 3 mV) to the 
normal state resistance decreases. Figure 2 shows a plot of this ratio as a function of critical 
current density. Above 50 kA/cm 2 , the ratio R S g/Ro is reduced to two; however, for current 
densities up to 40 kA/cm 2 the I-V characteristics are still reasonable. An example of a 0.42x0.42 
(i 2 , 35 kA/cm 2 junction is shown in figure 3; this junction was fabricated for 626 GHz mixer 
tests. 

A second form of I-V degradation occurs at high current densities. For a fixed area junction, 
the gap voltage decreases as the current density increases. This is caused by a non-thermal 
distribution of quasi-particles near the barrier. As the tunneling current increases, the number 
of quasi-particles injected into the superconducting electrode increases; this represents a local 
heating of the quasiparticle bath and leads to a reduction in the gap voltage. The quasi-particle 
heating can be minimized by reducing the junction area. Table 2 shows the gap for different 
area junction on the same wafer; the current density is 28 kA/cm 2 . 

Table 2. Energy Gap as a Function of Junction Size 



size (microns) 


Energy Gap (meV) 


1.0 


4.15 


0.7 


4.3 


0.5 


4.4 


0.36 


4.45 



Third International Symposium on Space Terahertz Technology Page 429 

Junction uniformity and yield was measured by fabricating series arrays of 100 tunnel junctions. 
Typical I-V characteristics for arrays of 0.49, 0.25 and .13 ^junctions are shown in figure 4. 
For these size junctions, the standard deviation of the critical currents is typically ±5.4, 6.6 and 
7.7% respectively. For comparison, the standard deviation for optically defined, 4 y? junctions 
is 3.9%. Typical minimum-to-maximum uniformity is ±17% for 0.25 n 2 junctions. When the 
fabrication process was successful, junction yields were better than 99 %. The uniformity and 
yield for this process were high enough that a monolithic array of mixers could be fabricated. 

SQUIDs have been fabricated to measure the magnetic penetration depth of NbN films and the 
specific capacitance of NbN/MgO/NbN junctions. The design of the SQUIDs is based on that 
of Magerlein [14] and is described in detail elsewhere [15]. Briefly, the SQUID design 
is a microstrip line over a ground plane with two tunnel junctions connecting the microstrip to 
the ground plane. The inductance of the microstrip line, and therefore the magnetic penetration 
depth, can be determined by observing the critical current of the two parallel-junctions versus 
control current passing along the microstrip line. The loop of the SQUID and the junction 
capacitance form an LC tank circuit. The Josephson oscillations will interact with this tank 
circuit to form a current step at a voltage proportional to this resonant frequency. The junction 
capacitance can be determined from the resonance voltage and the critical-current modulation 
curve. 

The magnetic penetration depth can also be determined using an open-ended microstrip-line 
connected to a Josephson junction. The microstrip line has an impedance which is periodic with 
frequency. The Josephson oscillations interact with the microstrip line circuit to yield a series 



Page 430 Third International Symposium on Space Terahertz Technology 

of resonances whenever the microstrip line reflects an inductance that cancels out the junction 
capacitance. An example of these resonances is shown in figure 5. The propagation velocity of 
the microstrip line can be calculated to be 

V p =e-5V//i-2L, 
where 5V is the spacing or the resonances and L is the length of the microstrip line stub. The 
penetration depth can be calculated from the propagation velocity. These microstrip line stubs 
are a good method of verifying the SQUID results and directly measuring the microstrip line 
propagation velocity. 

The specific capacitance of our junctions as a function of J c is shown in figure 6 for critical 
current densities ranging from 500 to 50,000 A/cm 2 . Much of this data has been published 
previously 14 ; however, we have extended the data to include the highest current densities shown 
here (10 to 40 kA/cm 2 ). Also shown in figure 6 is the R„C frequency as a function of current 
density. At 40 kA/cm 2 , the roll off frequency is only 140 GHz; therefore, at smm wavelengths, 
the capacitance must be tuned out to achieve optimal performance. 

The magnetic penetration depth of our NbN films has also been measured using SQUIDs. The 
penetration depth varies significantly with film quality and is typically between 270 and 380 nm. 
The propagation velocity of microstrip lines with 150 nm SiO dielectric layers is typically 0.16 
to 0.18 times the speed of light in a vacuum. The penetration depth calculated from these 
numbers is 300 to 365 nm, which agrees well with the SQUID results. Because the penetration 
depth depends critically on film quality, SQUID and microstrip line circuits are usually added 
to mixer mask-designs, so the penetration depth can be measured for each set of tunnel junctions 



Third International Symposium on Space Terahertz Technology Page 431 

that are fabricated. These circuits are small (2.5 mm x 2.5 mm) and 12 test dies take up 
roughly one sixth of a 25 mm diameter quartz wafer. 

Conclusions 
We have developed a process for fabricating high critical current, submicron NbN/MgO/NbN 
tunnel junctions. The yield and uniformity of this process is good and should be sufficient for 
most mixer needs. However, the RJ2 frequency can only be made 140 GHz without seriously 
degrading the I-V quality. Submillimeter wave mixers will either require high Q circuits to 
resonate out the junction capacitance or a new barrier material with a lower dielectric constant 
or barrier height. Although NbN films should have low RF losses even at smm wavelengths, the 
actual losses need to be measured in order to evaluate potential high Q tuning circuits. In 
addition, the effects, if any, of quasi-particle heating on mixer performance need to be 
investigated. 

Work Supported by NASA and SDI/IST 

References 



[1] C. K. Walker, M. Chen, P. L. Shafer, H. G. LeDuc, J. E. Carlstrom and T. G. Phillips, 
"A 492 GHz SIS Waveguide Receiver for Submillimeter Astronomy," Int. J. of IR and 
Millimeter Waves, Submitted 1992. 

[2] T. H. Buttgenbach, H. G. LeDuc, P. D. Maker and T. G. Phillips, "A Fixed Tuned 
Broadband Matching Structure for Submillimeter SIS Receivers," IEEE Trans. Appl. 
Superconductivity, Submitted Feb. 1992. 

[3] C. A. Mears, Qing Hu, P. L. Richards, A. H. Worsham, D. E. Prober and A. V. 
Raisanen, "Quantum Limited Quasiparticle Mixers at 100 GHz," IEEE Trans, on Magn., vol. 
27, no. 2, 1991. 



Page 432 Third International Symposium on Space Terahertz Technology 



[4] L. R. D'Addario, "An SIS Mixer for 90-120 GHz with Gain and Wide Bandwidth," Int. 
J. ofIR and Millimeter Waves, vol. 5, no. 11, 1419-1442, 1984. 

[5] A. V. Raisanen, W. R. McGrath, P. L. Richards and F. L. Lloyd, "Broad-Band Match 
to a Millimeter- Wave SIS Quasi-Particle Mixer," IEEE Trans, on Microwave Theory and 
Technique, vol. 4, no. 12, December 1985. 

[6] S. K. Pan, A. R. Kerr, M. J. Feldman, A. W. Kleinsasser, J. Stasiak, R. L. Sandstrom 
and W. J. Gallagher, "An 85-116 GHz SIS Receiver Using Inductively Shunted Edge- Junctions 
," IEEE Trans, on Microwave Theory and Technique, vol. 37, no. 3, 580-592, March 1989. 

[7] W. H. Chan, "The Inductance of a Superconducting Strip Transmission Line, " J. Appl. 
Phys., vol.50, no. 12, Dec. 1979. 

[8] R. L. Kautz, "Picosecond Pulses on Superconducting Striplines," J. Appl. Phys., vol. 
49, 308-314, 1978. 

[9] H. G. LeDuc, J. A. Stern, S. Thakoor and S. Khanna, "All Refractory NbN/MgO/NbN 
Tunnel Junctions," IEEE Trans. Magn. vol. 23, March 1987. 

[10] J. A. Stern, B. D. Hunt, H. G. LeDuc, A. Judas, W. R. McGrath, S. R. Cypher and 
S. K. Khanna, "NbN/MgO/NbN SIS Tunnel Junctions for Submm Wave Mixers," IEEE Trans. 
Magn., vol 25, 1989. 

[11] A. W. Lichtenberger, D. M. Lea, C. Li, F. Lloyd, M. J. Feldman and R. J. Mattauch, 
"Fabrication of Micron Size Nb/Al-Al 2 3 /Nb Junctions with a Trilevel Resist Liftoff Process, " 
IEEE Trans on Magnetics, vol. 27, no. 2, March 1991. 

[12] M. Spak, D. Mammato, S. Jain and D. Durham, "Mechanism and Lithographic 
Evaluation of Image Reversal in AZ 5214 Photoresist," As Presented at: Seventh International 
Technical Conference on Photopolymers, Ellenville, New York, Reprints Available from 
American Hoechst Corporation, AZ Photoresist Products. 

[13] H. G. LeDuc, A. Judas, S. R. Cypher, B. Bumble, B. D. Hunt and J. A. Stem, 
"Submicron Area NbN/MgO/NbN Tunnel Junctions For SIS Mixer Applications," IEEE Trans, 
on Magnetics, vol. 27, no. 2, March 1991. 

[14] J. H. Magerlein, "Specific Capacitance of Josephson Tunnel Junctions," IEEE Trans. 
Magn., vol. 17, no. 2, 286-289, 1981. 

[15] J. A. Stem and H. G. LeDuc, "Characterization of NbN Films and Tunnel Junctions," 
IEEE Trans. Magn., vol 27, no. 2, March 1991. 



Third International Symposium on Space Terahertz Technology Page 433 

Fi gure Captions 
Figure 1 Submicron, self-aligned liftoff process. 



Figure 2 Subgap resistance divided by the normal state resistance as a 

Junction of critical-current density 



Figure 3 I-V characteristic of a .42x.42 p. 2 , 35 kA/cm 2 , tunnel junction. The 
current scale is 20 pA/div. and the voltage scale 1 mV/div. 



Figure 4 I-V characteristics for series arrays of 100 tunnel junctions. The 
voltage scale is 50 mV/div. The area of the junctions is 0.13, 0.25, 
0.49 p. 2 , and the current scale is 10 20 and 50 pA/div for figures 
a, b and c respectively. 



Figure 5 I-V characteristic of a 1 p 2 tunnel junction connected to a 6 p by 
750 p open-ended microstrip-line stub. The current scale is 5 
pA/div, and the voltage scale is 100p.V/div. The resonance spacing 
is 65 p.V leading to a propagation velocity of 0.16 times the speed 
of light. 



Figure 6 Junction specific capacitance (a) and R n C roll-off frequency 
(l/2irR„C) (b) as a function of critical-current density. 



Page 434 



Third International Symposium on Space Terahertz Technology 



Figure 1. 



[ a ) 



\\\\\\\\\\\\\\\\\\\\\^^^ 


NbN 


mtzmmmm^m^. 


NbN 


V N \ N \ V \ N \ N \ N \ N \ N \ N \ X \ N \ N \ N \ N \ N \ N \ N \ N \ N \ N \ X \ N \ N \ N \ X 

Y\\y\\\\\\Y\\\a\\\\\\\\Y 

\\ v\ \\ 0\ \\ s\ n\ v\ \\ v\ \\ 0\ \\ \\ s\ \\ \\ v\ s\ \\ v\ n\ s\ v\ \ 



[ b ) 




SiO 




Third International Symposium on Space Terahertz Technology 



Page 435 



Figure 2. 
Rsg/Rn vs. J c 



12. 



8.0 - 



0.0 




4.0 L 



5.0 - 10.0 



20.0 50.0 


100.0 


200.0 


kA/cm 2 






Figure 3. 








Page 436 



Third International Symposium on Space Terahertz Technology 



Figure 4. 



lOmV 




Third International Symposium on Space Terahertz Technology 



Page 437 



Figure 5. 




■4 
'i 



i 

i 



Page 438 



Third International Symposium on Space Terahertz Technology 
Figure 6. 



(a) 



200.0 



fF/u 



Specific Capacitance vs. Current Density 




0.2 0.5 1.0 2.0 5.0 10.0 20.0 50.0 100.0 

kA/cm 2 



(b) 



RC Roll— Off Frequency vs. Current Density 



160.0 




2.0 5.0 10.0 20.0 50.0 100.0 

kA/cm 2 



Third International Symposium on Space Terahertz Technology Page 439 

A Quasioptical Resonant-Tunneling-Diode Oscillator " , 

Operating Above 200 GHz fj 9 ^S^-^^T^ 4 

E. R. Brown*, C. D. Parker*, K. M. Molvar*, A. R. Calawa*, and M. J. Manfra* /&$^ 



We have fabricated and characterized a quasioptically stabilized resonant-tunneling- 
diode (RTD) oscillator having attractive performance characteristics for application as a 
radiometric local oscillator. The fundamental frequency of the oscillator is tunable from 
about 200 to 215 GHz, the instantaneous linewidth is between 10 and 20 kHz, and the 
output power across the tuning band is about 50 H.W. The narrow linewidth and fine 
tuning of the frequency are made possible by a scanning semiconfocal open cavity which 
acts as the high-Q resonator for the oscillator. The cavity is compact, portable, and 
insensitive to vibration and temperature variation. The total dc power consumption (RTD 
plus bias supply) is only 10 mW. 

The present oscillator provides the highest power obtained to date from an RTD 
above 200 GHz. We attribute this partly to the use of the quasioptical resonator, but 
primarily to the quality of the RTD. It is fabricated from the Ino.53 Gao.47As/AlAs 
materials system, which historically has yielded the best overall resonant-tunneling 
characteristics of any material system. The RTD active area is 4 (im 2 , and the room- 
temperature peak current density and peak-to- valley current ratio are 2.5xl0 5 A cnr 2 and 9, 
respectively. The RTD is mounted in a WR-3 standard-height rectangular waveguide and 
is contacted across the waveguide by a fine wire that protrudes through a via hole in a 
Si3N4 "honeycomb" overlayer. We estimate that the theoretical maximum frequency of 
oscillation of this RTD is approximately 1.1 THz, and that scaled-down versions of the 
same quasioptical oscillator design should operate in a fundamental mode up to frequencies 
of at least 500 GHz. 

This work was sponsored by NASA-OAST through the Jet Propulsion Laboratory and by the U.S. 
Army Research Office. 



*E. R. Brown, C. D. Parker, K. M. Molvar, A. R. Calawa, and M. J. Manfra are 
with Lincoln Laboratory, Massachusetts Institute of Technology. 



a-Q 



( 



r 



Page 440 Third International Symposium on Space Terahertz Technology 

Transit-Time Devices as Local Oscillators for Frequencies Above 100 GHz *) 

-5JST-35 N93-27765 



H. Eisele. C. Kidner, G. I. Haddad 
/& O 5^3 

. \ V Center for Space Terahertz Technology 

~^ Department of Electrical Engineering & Computer Science 

2231 EECS Building 

The University of Michigan 

Ann Arbor, Michigan 48 109-2122 



Abstract: 

Very promising preliminary experimental results have been obtained from GaAs IMP ATT diodes at F- 
band frequencies (75 mW, 3.5 % @ 111.1 GHz and 20 mW, 1.4 % @ 120.6 GHz) and from GaAs 
TUNNETT diodes at W-band frequencies (26 mW, 1.6 % @ 87.2 GHz and 32 mW, 2.6 % @ 93.5 GHz). 
These results indicate that IMPATT, MIT ATT and TUNNETT diodes have the highest potential of deliv- 
ering significant amounts of power at Terahertz frequencies. As shown recently, the noise performance of 
GaAs W-band IMPATT diodes can compete with that of Gunn devices. Since TUNNETT diodes take 
advantage of the quieter tunnel injection, they are expected to be especially suited for low-noise local 
oscillators. This paper will focus on the two different design principles for IMPATT and TUNNETT 
diodes, the material parameters involved in the design and some aspects of the present device technology. 
Single-drift flat-profile GaAs D-band IMPATT diodes had oscillations up to 129 GHz with 9 mW, 0.9 % 
@ 128.4 GHz. Single-drift GaAs TUNNETT diodes had oscillations up to 112.5 GHz with 16 mW and 
output power levels up to 33 mW and efficiencies up to 3.4 % around 102 GHz. These results are the best 
reported so far from GaAs IMPATT and TUNNETT diodes. 



*) This work was supported by NASA under contract No. NAGW 1334. 



Third International Symposium on Space Terahertz Technology Page 441 



1. Introduction 

GaAs IMPact ionization Avalanche Transit-Time (IMP ATT) diodes have long been thought to be limited 
to frequencies below 60 GHz. Little has been reported regarding the operation of GaAs IMP ATT or 
MITATT diodes above 100 GHz [1,2]. Experimental results of W-band IMPATT diodes (up to 320 mW, 
6.0 % @ 95 GHz) [3] with excellent noise performance [4] clearly indicate that IMPATT diodes are one 
prime candidate to fulfill the growing need for local oscillators above 100 GHz. TUNNE1 injection 
Iransit-Time (TUNNETT) diodes were already proposed in 1958 and are considered another prime can- 
didate for low-noise, medium power sources at millimeter and submillimeter frequencies. Although 
pulsed oscillations were demonstrated up to 338 GHz in 1979 [5], CW power has only recently been 
obtained from devices with low impact ionization carrier multiplication [6]. This significant progress is 
mainly due to the fact that refined epitaxial growth techniques have become widely available. Despite the 
impressive progress in oscillators with three-terminal devices at mm-wave frequencies [7] two-terminal 
devices hold the highest potential in delivering significant amounts of power with clean spectra above 
100 GHz. 



2. Design of single-drift flat-profile IMPATT diodes 

In GaAs, the first derivative of the ionization rates of electrons and holes with respect to the electric field 
saturates around 500 kVcnr 1 [8-10]. Together with dead space effects in the avalanche zone [11] this sat- 
uration phenomenon favors a flat-profile structure for frequencies above V-band (50 - 75 GHz). The per- 
formance of a single-drift structure is the least sensitive to doping profile variations. The design of this 
structure is based on the assumption that the center of the avalanche region occurs where the electron 
concentration equals the hole concentration for the applied bias voltage [12] and that such a defined 
avalanche region is electrically equivalent to an avalanche region of the same width / a but constant elec- 
tric field and ionization rates [10]. The drift region - where ionization is to be neglected - is defined in its 
length / d by the maximum in the well known transit-time function [13] 

3v s 

/ d = • CD 

8/o 

where v s is the average saturated drift velocity (4.5 x 10 6 cms 1 in GaAs for 7j = 500 K) [9,10] and/ the 
operating frequency. Several structures for operating frequencies between 130 GHz and 160 GHz have 
been designed and the nominal doping profile of such a p ++ nn + structure is given in Figure 1. A bias cur- 
rent density Joe °f 60 kAcnr 2 extrapolated from the experimental results in W-band [3] was taken into 
account. 



Page 442 third International Symposium on Space Terahertz Technology 



3. Design of single-drift TUNNETT diodes 

The design of the TUNNETT diode structure is based on a first order large signal theory [14] and experi- 
mental studies of highly doped MBE grown p ++ n + structures. The carrier generation rate due to tunneling 
does not depend on the current density, but does strongly depend on the electric field. Therefore, a sharp 
pulse of electrons is injected at the p ++ n + junction when the RF field reaches its maximum. 

Under these assumptions for the p ++ n + nn + structure the first order large signal theory predicts a maximum 
in RF output power and DC to RF conversion efficiency [13,14] for 

5v s 
'i + 'd = . (2) 

where /j is the length of the n + region in the p ++ n + junction, / d is the length of the n region, v s is the aver- 
age saturated drift velocity (4.6 x 10 6 cms 1 for 7] = 500 K) [9,15] and/ the operating frequency. Since 
the design is based on considerably lower electric fields in the drift region compared to the ones in the 
IMPATT diodes above, a slightly higher value for v s is appropriate. Further details of the design proce- 
dure are given in References 16 and 17. The carrier concentration due to a current density /qc of 
25 kAcnr 2 is taken into account in the doping profile. The nominal doping profile of this p ++ n+nn + 
TUNNETT diode structure is depicted in Figure 2. 



4. Device technology 

The operating current density of 60 kAcnr 2 in a single-drift flat-profile GaAs D-band IMPATT diode 
requires a diamond heat sink to keep the operating junction temperature below 250 °C. Therefore all 
IMPATT diodes were fabricated using a well established selective etching technology for substrateless 
diodes on diamond heat sinks which gives up to 600 diodes per cm 2 wafer area with high uniformity [18]. 
This technology implements an Al 55 Ga 45 As etch-stop layer between the substrate and the epitaxial layers 
for the device. In order to get the steep transitions for doping profiles in the submicron range, all wafers 
were grown by MBE. Figure 3 shows the flow chart of this technology process. The epitaxial side of the 
wafer is metallized with Ti/Pt/Au for a p + ohmic contact, then selectively plated with gold to form a grat- 
ing for mechanical support and glued on a ceramic carrier. In the next step the substrate is removed by 
selective wet chemical etch and subsequently the etch-stop layer in another selective wet chemical etch. A 
Ni/Ge/Au contact metallization is evaporated on top of the n + layer and plated with gold to ease bonding. 
Contact patterns and diode mesas are defined by standard positive photoresist technology and wet chemi- 
cal etching. The diodes outside the supporting grating are tested and selected for good DC characteristics 



Third International Symposium on Space Terahertz Technology Page 443 



and thermocompression bonded on diamond heat sinks which are embedded in plated copper blocks. 
Electrical contact to the diode is provided by four metallized quartz stand-offs thermocompression 
bonded onto the heat sink and tapered gold ribbons bonded on the diode and the stand-offs. 

The TUNNETT diodes were designed to operate at a maximum current density of 25 kAcnr 2 and a DC 
bias voltage comparable to the one of the IMP ATT diodes. This allows fabrication of TUNNETT diodes 
with an integral heat sink. The wet chemical etching in the previously described technology limits the 
choice of materials and the minimum diameter for the n + ohmic contact. Therefore a different process has 
been developed, which likewise implements an Al 55 Ga 45 As etch-stop layer between the substrate and the 
epitaxial layers for the device. Its flow chart is given in Figure 4 and further details are discussed in 
Reference 19. 

Before the epitaxial side of the MBE-grown wafer is metallized with Ti/Pt/Au for a p + ohmic contact, 
grooves are selectively etched down to the AlGaAs etch-stop layer to divide the device layers into square 
shaped islands. This reduces the stress in the device layers during annealing. Furthermore, it shapes the 
Ti/Pt/Au layers and the plated gold layer of the integral heat sink thus providing additional mechanical 
strength for the subsequent processing steps after the substrate has been removed. The contacts are 
defined by standard lift-off technology and an additional metallization and photolithography step gives 
holes on top of the n + ohmic contact through which up to 3 urn of gold is electroplated. The mesas are 
formed by a wet chemical etch. After annealing the sample is diced into individual diodes and diodes with 
the desired size and DC characteristic are soldered or glued to a gold plated copper block. Electrical con- 
tact to the diode is provided by four metallized quartz stand-offs thermocompression bonded onto the 
plated block and tapered gold ribbons bonded on the diode and the stand-offs. 



5. Experimental results 

RF testing is performed in full height waveguide cavities with a resonant cap on top of the diode. 
IMP ATT diodes are tested both in W-band (WR-10 waveguide) and D-band (WR-6 waveguide). 
TUNNETT diodes are only tested in W-band. 

Figure 5 shows RF output power, DC to RF conversion efficiency and oscillation frequency of the best 
IMP ATT diode in a W-band cavity as a function of the bias current. At each bias point the short plunger 
of the cavity and at some bias points also the E-H-tuner were adjusted for maximum output power. As can 
be seen from Figure 5, the efficiency reaches its maximum of 3.8 % at an output power of 72 mW. 

An output power of 85 mW at 102.0 GHz with an efficiency of 2.5 % in WR-10 waveguide cavity and 
20 mW at 120.6 GHz with an efficiency of 1.4 % in a WR-10 waveguide cavity were obtained from other 



Page 444 Third International Symposium on Space Terahertz Technology 



IMPATT diodes. The highest oscillation frequency of 128.4 GHz could be observed in a WR-6 wave- 
guide cavity. At this frequency the output power was 9 mW and the efficiency 0.9 %. Table 1 summarizes 
the experimental results obtained from these diodes. The operating junction temperature was limited up to 
Tj = 550 K in order to ensure reliable long-term operation. 



Frequency [GHz] 


102.0 


111.1 


111.5 


120.6 


128.4 


Output power [mW] 


85 


75 


48 


20 


9 


Efficiency [ % ] 


2.5 


3.5 


2.3 


1.4 


0.9 


Cavity (W/D) 


W 


W 


W 


D 


D 



Table 1: Experimental results of IMPATT diodes in W-band and D-band cavities. 

To verify the mode of operation the DC I-V characteristics are measured at room temperature (7 = 300 K) 
and an elevated temperature (7 = 370 K). As shown in Figure 6a for low bias currents and Figure 6b for 
high bias currents, the breakdown of the D-band IMPATT diode is sharp and the bias voltage always 
increases with increasing temperature due to the decreasing ionization rates. The breakdown voltage at 
7= 300 K agrees well with breakdown voltages that were calculated from ionization rates evaluated in 
Reference 9, and which are plotted in Figure 7 together with the peak electric field strength. The sharp 
breakdown also proves that tunneling is significant only for electric field strengths above 1.0 MVcnr 1 . 

Figure 8 shows RF output power, DC to RF conversion efficiency and oscillation frequency of two 
TUNNETT diodes in W-band cavities as a function of the bias current. At each bias point the short 
plunger and the E-H-tuner were adjusted for maximum output power. As can be seen from Figure 8a and 
8b, neither output power nor efficiency saturate up to the highest applied bias currents. The oscillation 
frequency varies only slightly and is mainly determined by the resonant cap. An output power of 33 mW 
at 93.5 GHz with an efficiency of 2.65 % and an output power 31.5 mW at 107.36 GHz with an efficiency 
of 3.35 % were obtained. The highest oscillation frequency of another diode was 112.5 GHz with an 
output power of 16 mW and the corresponding efficiency of 2.55 %. Table 2 summarizes the 
experimental results obtained from the so far best diodes. The operating junction temperature was well 
below 7j = 550 K in each case. 



Frequency [GHz] 


87.22 


93.50 


102:66 


107.30 


112.50 


Output power [mW] 


27 


33 


33 


31.5 


16 


Efficiency [ % ] 


1.75 


2.65 


3.35 


3.35 


2.55 



Table 2: Experimental results of TUNNETT diodes in W-band cavities 



Third International Symposium on Space Terahertz Technology Page 445 



A plot of the output power and efficiency of the W-band diodes that have been mounted and tested to date 
is given in Figure 9. There appears to be a broad peak in the RF output power and DC to RF conversion 
efficiency around the nominal design frequency of 100 GHz. This peak confirms that the first order 
design rules accurately predict the operating frequency of the TUNNETT diodes. It also indicates that the 
high field, high temperature electron average drift velocity in GaAs TUNNETT diodes is close to 
4.6 x 10 6 cms -1 . The power levels and efficiencies above 100 GHz are comparable to the ones obtained 
from Gunn devices in this frequency range [20-22]. 

To verify the mode of operation the DC I- V characteristics are measured at room temperature (T = 300 K) 
and elevated temperatures (T = 470 K). The I-V curves of a 25 ujn diameter W-band TUNNETT diode 
shown in Figure 10 clearly demonstrate that the injection mechanism is predominantly tunneling. For 
comparison the I-V curves of a 55 urn V-band Mixed Tunneling and Avalanche Transit-Time (MIT ATT) 
diode are also given in Figure 10. At room temperature the MIT ATT diode has a sharp increase in current 
at about 18 V due to the onset of impact ionization [16]. The TUNNETT diode I-V curve at room temper- 
ature exhibits no sign of this behavior. Tunneling as the dominant breakdown mechanism also explains 
the temperature dependence of the TUNNETT diode I-V curves. Increasing the junction temperature of 
the device enhances tunneling and suppresses impact ionization as can be seen in the temperature behav- 
ior of the MIT ATT diode. For low bias voltages the current increases, thus indicating tunneling. The volt- 
age for the sharp increase in current has a positive temperature coefficient, characteristic of impact ion- 
ization as previously shown in Figure 6. For the range of applied bias voltages the current in the 
TUNNETT diode always increases as a function of temperature implying that impact ionization is not 
significant. 

Figure 1 1 shows the measured spectra of a free running W-band IMP ATT diode oscillator with 42.8 mW 
at 89.2 GHz (Figure 11a) and a free running W-band TUNNETT diode oscillator with 9.2 mW at 
92.2 GHz (Figure 1 lb), and proves that the oscillations have clean spectra. The spectrum of another free 
running TUNNETT diode oscillator in Figure 12 was taken using the same settings (vertical scale, scan 
width and resolution bandwidth) of the spectrum analyzer as in Reference 23 for an free running InP 
Gunn device oscillator and it compares favorably to the spectrum of this Gunn device oscillator. 



6. Device simulation 

In order to determine the capabilities of GaAs IMPATT diodes at D-band frequencies and in order to find 
an explanation for the significant decrease in output power above 110 GHz, the device structures were 
simulated using two IMPATT diode simulation programs, a drift-diffusion (DD) model [24] and an 
energy-momentum (EM) model [25]. Table 3 shows calculated output power and efficiency at/= 95 GHz 
as preliminary results for both programs. The data for the device area A D and current density Jpc are 



Page 446 



Third International Symposium on Space Terahertz Technology 



taken from Reference 3. The energy-momentum program shows slightly higher breakdown voltages and 
higher efficiency and output power. If a series resistance R s = 0.18 Q is taken into account for this diode, 
the calculated output power and efficiency are much closer to the measured values. This series resistance 
is comparable to the value obtained from small signal impedance measurements in forward direction at 
32 MHz [9,10]. 





W-band IMPATT diode 




Frequency: 95 GHz Area A D : 8x 


lO^cm 2 Current density: 


50 kAcm- 2 


Model Voltage 


Power Efficiency 


Power Efficiency 






(/? 5 = on) 


(fl s = 0.18Jl) 




[V] 


[mW] [%] 


[mW] [%] . 




DD 12.2 


550 11.3 


320 6.5 




EM 12.5 


700 14.0 


510 10.2 







D-band IMPATT diode 




Frequency: 140 GHz Area/l D : 5x 


10' 6 cm 2 Current density: 60 kAcm- 2 


Model Voltage 


Power Efficiency 


Power Efficiency 


Power Efficiency 




(* S = 0Q) 


(/? s = o.20 n) 


(R s = 0.288 £2) 


[V] 


[mW] [%] 


[mW] [%] 


[mW] [%] 


DD 10.4 


120 3.8 


35 1.1 


12 0.4 


EM 10.9 


215 6.5 


80 2.4 


18 0.5 



Table 3: Calculated results for GaAs single-drift flat-profile IMPATT diodes. 



The results for the D-band structure in Table 3 were calculated for no series resistance and two different 
values of the series resistance. R s = 0.288 fl assumes that the series resistance is mainly due to the contact 
resistances of the p + and n + layers and scales with reciprocal area, i.e. it is equivalent to R s = 0. 18 £2 of the 
W-band diode. For this series resistance the output power is reduced to about one tenth of the output 



Third International Symposium on Space Terahertz Technology 



Page 447 



power for the case of no series resistance taken into account. Since the calculated output power agrees 
with the experimental value of 9 mW at 128.4 GHz, the series resistance is believed to be the main reason 
for the significant rolloff in performance above 1 10 GHz. As can be seen also from Table 3, the output 
power reduction is only about one third and therefore much less pronounced, if a slightly smaller 
# s x ^d (1 x 10" 6 ftcm 2 ) is assumed. This demands better technology for contacts on both p + - and n + - 
type GaAs. 

Neither the drift-diffusion model nor the energy-momentum model consider any losses in the cavity. 
These losses are due to the large transformation ratio (up to 500) from the low impedance level between 
the contacts of the diode and the high impedance level of the waveguide. 

The simplified large-signal model for TUNNETT diodes [ 14] which was employed in the design was also 
used to determine how strongly the series resistance influences output power and efficiency of these 
devices. The above mentioned drift velocity v s and the actual device dimensions (mesa height and 
diameter, heat sink thickness, etc.) were used for the simulation. In Table 4 the specific contact resistance 
was assumed to be p c = 1 x 10 7 Qcm 2 for the p + ohmic as well as for the n + ohmic contact. In this case 
the predicted RF output power into a load of R\ = 1 Q, is 251 mW for experimentally investigated 
diameters around 25 urn. 



Freq 
(GHz) 


Drift Length 
(/mi) 


Drift Field 
(kV/cm) 


V DC 
(Volts) 


Vrf 
(Volts) 


Vdc /Vrf 


Jdc 
kA/cm 2 


100.0 


0.345 


309.8 


12.29 


10.89 


0.886 


32.84 



DIAM 
(/im) 


AREA 
(/im 2 ) 


R, 

(Ohm) 


Rd 
(Ohm) 


Ri 
(Ohm) 


Vrf 
(Volt) 


Idc 
(mA) 


Pdc 
(W) 


Prf (Gen) 
(mW) 


Prf (Load) 
(mW) 


15 


177 


0.33 


2.36 


2.02 


10.89 


58 


0.71 


134 


115 


20 


314 


0.21 


1.33 


1.12 


10.89 


103 


1.27 


238 


201 


25 


491 


0.15 


1.15 


1.00 


8.40 


161 


1.98 


287 


251 


30 


707 


0.11 


1.11 


1.00 


6.35 


232 


2.85 


313 


281 



DIAM 
(/im) 


Pdc 
(W) 


Prf 
(mW) 


Rth (Cu) 
(°C/W) 


Rth (Di) 
(°C/W) 


AT (Cu) 

CO 


AT (Di) 
(°C) 


15 


0.71 


115 


213 


140 


127 


84 


20 


1.26 


201 


140 


86 


149 


91 


25 


1.98 


251 


103 


59 


178 


102 


30 


2.85 


281 


80 


44 


207 


113 



Table 4: Performance of TUNNETT diodes at 100 GHz for p c = 1 x 10 7 Qcm 2 and V RF /V DC < 0.886. 



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Third International Symposium on Space Terahertz Technology 



As described above for the IMP ATT diodes, present GaAs technology, however, gives a specific contact 
resistance closer to p c = 7.5 x 10 7 ficm 2 . As a result, the RF output power into 1 Q decreases to 171 mW. 
For the predicted results in Table 4 and 5 the maximum RF voltage was limited to 88.6 % of the applied 
DC bias voltage. Since IMP ATT diodes at millimeter wave frequencies operate at an RF voltage around 
or less than 50 % of the DC bias voltage, this case was also investigated for the TUNNETT diodes. As 
can be seen from Table 6 the RF output power drops to 158 mW for experimentally investigated diame- 
ters around 25 urn. 



Freq 
(GHz) 


Drift Length 
(/im) 


Drift Field 
(kV/cm) 


V DC 
(Volts) 


v RF 

(Volts) 


Vdc /Vrf 


Jdc 
kA/cm 2 


100.0 


0.345 


309.8 


12.29 


10.89 


0.886 


32.84 



DIAM 
(urn) 


AREA 
(/*m J ) 


R. 

(Ohm) 


R* 
(Ohm) 


R, 
(Ohm) 


Vrf 
(Volt) 


Idc 
(mA) 


Pdc 
(W) 


Prf (Gen) 
(mW) 


Prf (Load) 
(mW) 


15 


177 


1.07 


2.36 


1.29 


10.89 


58 


0.71 


134 


73 


20 


314 


0.62 


1.62 


1.00 


9.15 


103 


1.27 


200 


124 


25 


491 


0.41 


1.41 


1.00 


7.05 


161 


1.98 


241 


171 


30 


707 


0.30 


1.30 


1.00 


5.61 


232 


2.85 


276 


213 



DIAM 
{fim) 


Pdc 
(W) 


Prf 

(mW) 


Rth (Cu) 
(°C/W) 


Rth (Di) 
(°C/W) 


AT (Cu) 
(°C) 


AT 1 (Di) 
(°C) 


15 


0.71 


73 


213 


140 


136 


90 


20 


1.27 


124 


140 


86 


160 


98 


25 


1.98 


171 


103 


59 


186 


107 


30 


2.85 


213 


80 


44 


212 


116 



Table 5: Performance of TUNNETT diodes at 100 GHz for p c = 7.5 x 10 7 Qcm 2 and V^V^ < 0.886. 



The calculated values of the thermal resistance R A and temperature rise AT for a copper (Cu) or diamond 
(Di) heat sink are also included in Tables 4, 5 and 6. It should be noted that experimental values for the 
thermal resistance always are higher than calculated. Therefore, a diode with a diameter of 30 urn on a 
copper heat sink will be operated at a lower DC input power Pqq and therefore reduced RF output power 
Prf to achieve an operating junction temperature below 250 "C. 



Similar to the IMPATT diode simulation, the simplified large-signal TUNNETT diode simulation does 
not account for any losses due to the large transformation ratio from the low impedance level between the 
contacts of the diode and the high impedance level of the waveguide. Since the structure of the resonant 
cap full height waveguide cavity has been optimized for IMPATT diodes its impedance transformation 
losses are expected to be higher for the TUNNETT diodes. 



Third International Symposium on Space Terahertz Technology 



Page 449 



Freq 
(GHz) 


Drift Length 
(fim) 


Drift Field 
(kV/cm) 


V DC 
(Volts) 


Vrf 
(Volts) 


Vdc /Vrf 


Jdc 
kA/cm 2 


100.0 


0.345 


309.8 


12.29 


10.89 


0.500 


32.84 



DIAM 
(fim) 


AREA 
(/im 2 ) 


Rs 
(Ohm) 


Rd 

(Ohm) 


Ri 
(Ohm) 


Vrf 
(Volt) 


Idc 
(mA) 


Pdc 
(W) 


Prf (Gen) 
(mW) 


Prf (Load) 
(mW) 


15 


177 


1.07 


4.63 


3.56 


6.14 


58 


0.71 


76 


58 


20 


314 


0.62 


2.61 


1.99 


6.14 


103 


1.27 


134 


103 


25 


491 


0.41 


1.67 


1.26 


6.14 


161 


1.98 


210 


158 


30 


707 


0.30 


1.30 


1.00 


5.61 


232 


2.85 


276 


213 



DIAM 
(fim) 


Pdc 
(W) 


Prf 
(mW) 


Rth (Cu) 
(°C/W) 


Rth (Di) 
(°C/W) 


AT (Cu) 
(•C) 


AT (Di) 
PC) 


15 


0.71 


58 


213 


140 


136 


90 


20 


1.27 


103 


140 


86 


162 


98 


25 


1.98 


158 


103 


59 


186 


107 


30 


2.85 


213 


80 


44 


212 


116 



Table 6: Performance of TUNNETT diodes at 100 GHz for p c = 7.5 x 10" 7 Qcm 2 and V RF /V l3C < 0.50. 



7. Conclusions 

The experimental results clearly show that GaAs IMP ATT diodes are powerful devices not only for 
frequencies below 60 GHz, but also above 100 GHz. The predicted results of two different simulation 
programs agree with the experiment. These simulations also reveal that the contact technology is very 
crucial for high output power and efficiency and must be improved considerably for GaAs D-band 
IMPATT diodes. The results from D-band GaAs IMPATT diodes and from W-band GaAs TUNNETT 
diodes are the best reported to date. Both IMPATT and TUNNETT diodes exhibit clean spectra for local 
oscillator applications. The TUNNETT diodes demonstrate useful power levels and efficiencies compara- 
ble to Gunn devices. Since RF output power and DC to RF conversion efficiency do not saturate up to the 
highest applied DC bias currents, still higher output power levels and efficiencies can be expected from 
TUNNETT diodes on diamond heat sinks in optimized cavities. 



References 

[1] Elta, M. E., Fettermann, H. R., Macropoulos, W. V., and Lambert, J.: "150 GHz GaAs MITATT 
source", IEEE Electron Device Letters, EDL-1, 1980, pp. 115-1 16. 



[2] Chang, K., Kung, J. K., Asher, P. G., Hayashibara, G. M., and Ying, R. S.: "GaAs Read-type 
IMPATT diode for 130 GHz CW operation", Electronics Letters, 17, 1981, pp. 471-473. 



Page 450 Third International Symposium on Space Terahertz Technology 



[3] Eisele, H., and Grothe, H.: "GaAs W-band IMPATT diodes made by MBE", Proc. MIOP '89, 
Sindelfingen, FRG, Feb. 28th - March 3rd 1989, Session 3A.6. 

[4] Eisele, H.: "GaAs W-band IMPATT diodes for very low-noise oscillators", Electronics Letters, 26, 
1990, pp. 109-110. 

[5] Nishizawa, J., Motoya, K., and Okuno, Y.: "Submillimeter Wave Oscillation from GaAs TUNNETT 
Diode", Proceedings of the 9th European Microwave Conference, 1979, pp. 463-467. 

[6] PObl, M., Freyer, J.: "Characterization of W-Band CW TUNNETT Diode", Proceedings of the 21st 
European Microwave Conference, Stuttgart, FRG, 1991, pp. 1496-1501. 

[7] Kwon, Y., Pavlidis, D., Tutt, M., Ng, G. I., Lai, R., and Brock, T.: "W-Band Monolithic Oscillator 
Using InAlAs/InGaAs HEMTs", Electronics Letters, 26(18), 1990, pp. 1425-1426. 

[8] Rolland P. A., Friscourt M. R., Lippens D., Dalle C., and Nieruchalski, J. L.: "Millimeter Wave 
Solid-State Power Sources", Proceedings of the International Workshop on Millimeter Waves, 
Rome, Italy, April 2-4, 1986, pp. 125-177. 

[9] Eisele, H.: "Electron properties in GaAs for the design of mm-wave IMPATTs", International 
Journal of Infrared and Millimeter Waves, 4, 1991, pp. 345-354. 

[10] Eisele, H.: "GaAs W-Band IMPATT diodes: The first step to higher frequencies", Microwave 
Journal, 34, 1991, pp. 275-282. 

[11] Okuto, Y., and Crowell, C. R., "Threshold energy effects on avalanche breakdown voltage in semi- 
conductor junctions", Solid-State Electronics, 18, 1975, pp. 161-168 

[12] Hulin, R.: "GroBsignalmodell von Lawinenlaufzeitdioden", Ph.D. Thesis Techn. University Braun- 
schweig, Braunschweig, 1973. 

[13] Harth., W., Claassen, M.: "Aktive Mikrowellendioden", Springer- Verlag, Berlin, 1981. 

[14] Haddad, G. I., East, J. R., and Kidner, C.: "Tunnel Transit-Time (TUNNETT) Devices for Terahertz 
Sources", Microwave and Optical Technology Letters, 4, 1991, pp. 23-29. 

[15] Allam, R., and Pribetich, J.: "Temperature Dependence of Electron Saturation Velocity in GaAs", 
Electronics Letters, 26, 1990, pp. 688-689. 

[16] Kidner, C., Eisele, H., and Haddad, G. I.: "Tunnel Injection Transit-Time Diodes for W-Band Power 
Generation", Electronics Letters, 28, 1992, pp. 511-513. 



Third International Symposium on Space Terahertz Technology 



Page 451 



[17] Kidner, C., Eisele, H., East, J., and Haddad, G. I.: "Design, Fabrication and Evaluation of Tunnel 
Transit-Time Diodes for V-Band and W-Band Power Generation", to be presented at the 1992 IEEE 
MTT-S International Microwave Symposium, June 1 - June 5, 1992, Albuquerque, New Mexico. 

[18] Eisele, H.: "Selective etching technology for 94 GHz GaAs IMPATT diodes on diamond heat sinks", 
Solid-State Electronics, 32, 1989, pp. 253-257. 

[19] Kamoua, R., Eisele, H., East, J. R., Haddad, G. I., Munns, G., Sherwin, M.: "Modeling, Design, 
Fabrication, and Testing of InP Gunn Devices in the D-Band (110 GHz - 170 GHz), these 
Proceedings of the 3rd International Symposium on Space Terahertz Technology, March 24-26, 
1992, Ann Arbor, Michigan. 

[20] Wandinger, L.: "mm-Wave InP Gunn Devices: Status and Trends", Microwave Journal., 24(3), 
1981, pp. 71-78. 

[21] Eddison, I. G., et al.: "Efficient fundamental frequency oscillation from mm-wave InP n + -n-n + 
TEOs", Electronics Utters, 17(20), 1981, pp. 758-760. 

[22] Teng, S. J. J., Goldwasser, R. E.: "High Performance Second-Harmonic Operation W-Band GaAs 
Gunn Diodes", IEEE Electron Device Letters, EDL-10(9), 1989, pp. 412-414. 

[23] Perrin, O., et al.: "380 GHz Receiver Front-End for the Balloon-Borne Radioastronomical 
Experiment", Proceedings of the 2nd International Symposium on Space Terahertz Technology, 
February 26-28, 1991, Pasadena, California, pp. 622-640. 

[24] Bauhahn, P. E., and Haddad, G. I.: "IMPATT device simulation and properties", IEEE Transactions 
on Electron Devices, ED-24, 1977, pp. 634-642. 

[25] Mains, R. K... Haddad, G. I., and Blakey, P. A.: "Simulation of GaAs IMPATT Diodes Including 
Energy and Velocity Transport Equations", IEEE Transactions on Electron Devices, ED-30, 1983, 
pp. 1327-1338. 



5 x 1 1 9 cm" 3 




5x I0 ,8 cm" 3 




Sx I0 l9 cm" 3 




5x lO^cm" 3 




p" 


• 3 x I0 17 cm" 3 






p" 






n* puffer 




3xlO ,8 cm' 3 


n* puffer 


HP 
12 


7.3 x lO^cm" 3 


n 


n" 


— fc .. 








_ 


"300 nm 


_ 


240 nm 


• 270 nm 


1500 nm 




240 run 


40 nm 


1500 nm 





Fig. I: Nominal device structure of a GaAs 
D-band single-drift flat-profile IMPATT 
diode. 



Fig. 2: Nominal device structure of a GaAs 
W-band single-drift TUNNETT diode. 



Page 452 



Third International Symposium on Space Terahertz Technology 



FLOW DIAGRAM FOR ETCH-STOP GaAs IMPATT DIODE FABRICATION PROCESS 



Awimmmmmmw 



pt contact (Ti/Pt/Au) 
active layer ■ 




holder 



supporting 
grating (Au) 



etch -stop layer 
GaAs substrate 



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1) Metal evaporation for the 
p+- ohmic contact 





wsssssssssssssssssssssssssssss*. 



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down to the etch-stop layer 




W^ 




photoresist 



5) Etching the contact and the mesa 



2) Electroplating (Au) for the supporting 
grating and gluing sample to holder 




n + -contact 
(Au/Ge/Ni) 



4) Removing the etch-stop layer 
metallizing & electroplating for the 
rf"- ohmic contact 

supporting 
grating (Au) 

P- ohmic contact 




n -ohmic contact 



6) Annealing ohmic contacts 
► Final device 



Fig. 3: Flow chart for IMPATT diodes device fabrication. 



Third International Symposium on Space Terahertz Technology 



Page 453 



Plated Au 
Ti/Pt/Au 

p*GaAs 



n + A!GaAs 




GaAs 

aSSSSSSSSSBBSSBBSsS 



:B55S5cg£c5gccBc£cc55cS5cccge5geo£eccBSSc££5c&S 









a) Island definition, p-ohmic evaporation, and 
gold plating (~ IS um) 



n-ohmic 
Ni/Ge/Au/Ti/Au 




b) Substrate thinning, etch stop layer removal, 
and n-ohmic evaporation 



Ti/Au/Tl 




Photoresist 



c) Gold plating of ohmic contacts 




d) Final diodes after annealing and mesa etch 



Fig. 4: Flow chart for TUNNETT diodes device fabrication. 



125 



too 



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75 


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JB. 






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112 



50 100 150 200 
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0) 

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Fig. 5: Output power, efficiency and oscillation 
frequency as a function of bias current 
for a GaAs single-drift flat-profile 
IMPATT diode. 



Fig. 7: Breakdown voltage V hT and peak electric 
field £ max of an abrupt p + n-junction. 
■ * : measured 
: calculated. 



Page 454 



Third International Symposium on Space Terahertz Technology 



IF 

(jiA) 

SO. 00 



10.00 

/dlv 



7=300K -» 



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Fig. 6: Current-voltage characteristics of a GaAs single-drift flat-profile IMPATT diode at room 
temperature (300 K) and an elevated temperature (370 K). 



J. 



40 



30 



20 



10 



+ Frequency 
■ Efficiency 
• Power 

c 

V 

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94 



93 



50 75 100 125 

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30 50 70 90 

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3 
c* 
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£ 



(a) (b) 

Fig. 8: Output power, efficiency and oscillation frequency as a function of bias current for two W-band 
GaAs single-drift TUNNETT diodes. 



Third International Symposium on Space Terahertz Technology 



Page 455 



I 

u 

I 




100 
frequency [GHz] 



120 



i 

9) 

1 



Fig. 9: Output power and efficiency of GaAs 
single-drift TUNNETT diodes in 
W-band. 



20 mA 



200 pA 



20|iA 



2|iA 





/ 7 




y-+ 


TUNNETT A 


y 1 

MFTATT J 


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470K— -V/— JOOK 


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Fig. 10: Reverse bias current-voltage character- 
istics for pure tunnel injection 
(TUNNETT) and mixed tunnel injection 
and impact ionization (MITATT) at 
room temperature (300 K) and elevated 
temperatures (470 K and 400 K, 
respectively). 



l*wl ranuBcv 

-1006M ca B9.158 71GHZ 

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200KHZ 




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10KHZ 

LUTIO 
WIDTH 



(a) 
Fig. 11: Spectrum of a W-band IMP ATT diode 
free running oscillator, power level 
42.8 mW, center frequency 89.16 GHz, 
vertical scale 10 dB/div, horizontal scale 
200 kHz/div, BW 10 kHz. 



TEX 
Z755P 



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92.210 916HZ 



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FT 




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(b) 
Fig. 1 1 : Spectrum of a W-band TUNNETT diode 
free running oscillator, power level 
9.2 mW, center frequency 92.21 GHz, 
vertical scale 10 dB/div, horizontal scale 
200 kHz/div, BW 10 kHz. 



Page 456 



Third International Symposium on Space Terahertz Technology 



LEVEL 

ref -iODBM 
hkh -16.4DBM 



TEK 
2755P 



FREQUENCY 
cen 94.068GHZ 
hkr 94.06BGHZ 



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500KHZ 




DBH 
-10 



-20 



-30 



-40 



-00 



-60 



-70 



-BO 



-90 



10DB/ 
VERTICAL 
DISPLAY 



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RF FREQ 

ATTENUATION RANGE 



3KHZ 100KHZ 

VIDEO RESOLUTION 
FILTER BANDHIDTH 



Fig. 12: Spectrum of a W-band TUNNETT diode free running oscillator, power level 8.8 mW, center 
frequency 94.07 GHz, vertical scale 10 dB/div, horizontal scale 500 kHz/div, BW 100 kHz. 



Third International Symposium on Space Terahertz Technology Page 457 

Negative Differential Resistance (NDR) 5f0%33 

Frequency Conversion with Gain /^>OSSj 

R. J. Hwu, R. W. Aim, and S. C. Lee 

Department of Electrical Engineering 

University of Utah 

Abstract—The dependence of the I-V characteristic of the negative differential resistance 
(NDR) devices on the power level and frequency of the rf input signal has been 
theoretically analyzed with a modified large- and small-signal nonlinear circuit analysis 
program [1,2]. The NDR devices we used in this work include both the tunnel diode 
(without the antisymmetry in the I-V characteristic) and resonant-tunneling devices (with 
the antisymmetry in the I-V characteristic). Absolute negative conductance can be found 
from a zero-biased resonant tunneling device when the applied pump power is within a 
small range. This study verifies the work of Sollner et al. [3]. Variable negative 
conductances at the fundamental and harmonic frequencies can also be obtained from both 
the unbiased and biased tunnel diodes. The magnitude of the negative conductances can be 
adjusted by varying the pump amplitude— a very useful circuit property. However, the 
voltage range over which the negative conductance occurs moves towards the more positive 
side of the voltage axis with increasing frequency. Furthermore, the range of the pumping 
amplitude to obtain negative conductance varies with the parasitics (resistance and 
capacitance) of the device. The theoretical observation of the dependence of the I-V 
characteristic of the NDR devices on the power and frequency of the applied pump signal is 
supported by the experimental results. In addition, novel functions of a NDR device such 
as self-oscillating frequency multiplier and mixer with gain have been experimentally 
demonstrated. The unbiased oscillator have also been successfully realized with a NDR c •> 



Page 458 Third International Symposium on Space Terahertz Technology 

device with an antisymmetrical I-V characteristic. Finally, the applications of these device 
functions will be discussed. % - 

INTRODUCTION 
There have been increased interest in the study of resonant tunneling devices due to 
the fact that the characteristics of these devices can be engineered to have properties for 
very high-speed applications. In particular, their ability to exhibit negative differential 
resistance (NDR) regions lead to their potential use as gain elements in circuits and offers a 
new opportunity for circuit design. The presence of peaks and valleys in the I-V curve 
combined with the overall antisymmetry of the I-V curve about the origin [i.e., I(V) = -I(- 
V)], also offers the potential for efficient odd-harmonic generation with an unbiased 
resonant tunneling device [4,5]. The key lies in pumping the device so that the peak 
amplitude of the voltage across the device occurs above the resonant current peaks. This 
will produce more than three local maxima in the device current waveform over one cycle, 
corresponding to third or higher odd harmonic generation. The resonant tunneling 
frequency multiplier, therefore, has several distinct advantages over existing resistive 
multipliers, which are usually based on Schottky barrier diodes. The antisymmetrical 
response provides the potential for efficient odd harmonic frequency multiplication with an 
unbiased resonant tunneling device due to cancellation of the even harmonics, therefore 
greatly simplifying the circuit design. The maximum harmonic generation efficiency of a 
resonant tunneling device is significantly higher than the 1/n^ (n is the harmonic number) 
value that applies to standard resistive multipliers because of its negative resistance [4,5] 
(i.e., nonmonotonically increasing function I-V characteristic). The resonant tunneling 
device also has the ability to act as an efficient mixer due to the rapid variation of the 
dynamic conductance with voltage near the NDR region of the I-V curve. The resonant 
tunneling mixer has the potential to displace the Schottky diode in many microwave and 
millimeter-wave applications. The most intriguing aspect of the resonant tunneling 



Third International Symposium on Space Terahertz Technology Page 459 

frequency multiplier and mixer is its intrinsic capability to achieve conversion gain 
(efficiency > 1). 

LARGE- AND SMALL-SIGNAL NONLINEAR-CIRCUIT ANALYSIS 

A large- and small-signal analysis program has been developed to analyze the 
behavior of dc and microwave negative conductance of a NDR device. The analysis 
technique was developed by T. Kerr [6] and the computer program was implemented for 
analyzing ideal Schottky barrier diodes by Siegel et. al. [1]. The analysis program has 
been modified to take into account the negative resistance of the NDR device [2]. The I-V 
characteristics measured from the NDR devices as can be seen in Fig. 1 have been used in 
the nonlinear-circuit analysis. Since the devices were mounted on a 50 Q. microstrip line 
for the measurements in this work, the embedding impedance of 50 Q. at every harmonic 
frequency has been used. A simple experiment has been carried out to verify that the 
embedding impedance at higher harmonic frequencies is equal to 50 Q [7]. 
Tunnel Diode— Without Antisymmetry in the I-V Characteristic 

The differential conductance of a tunnel diode biased at zero voltage and in the 
positive differential resistance (PDR) region (close to the current peak) under different rf 
pumping conditions has been studied in this work. No negative conductance has been 
observed at dc for a tunnel diode biased at zero voltage and in the PDR region. However, 
negative conductances at the fundamental and different harmonic frequencies have been 
observed from a tunnel diode biased at zero voltage and in the PDR region. The magnitude 
of the negative conductance varies with the pump amplitude (see Fig. 2). The pump 
amplitude region required to achieve the negative conductance moves toward the more 
positive side of the power axis with increasing pumping frequency (also see Fig. 2). This 
can be easily explained using the equivalent circuit model of a tunnel diode (see Fig. 1 (c)). 
Since the impedance of the parallel circuit section decreases with the increasing frequency, 
more voltage will be distributed on the series resistance and less voltage on the parallel 
circuit section for the higher pumping frequency. The negative conductance observed at the 



Page 460 Third International Symposium on Space Terahertz Technology 

fundamental and different harmonic frequencies can be used as the basis for harmonic 
oscillators. It should be pointed out that the magnitude of the negative conductance at the 
fundamental is much higher than those at other harmonic frequencies with a runnel diode 
biased in the PDR region, close to the current peak (see Fig. 2 (b) and (c)). At this bias 
point, near the region of greatest curvature, the Fourier series of the conductance 
waveform has a predominant coefficient at the oscillation frequency. In addition, the 
negative conductances at the odd harmonic frequencies are higher than those at the even 
harmonic frequencies for the tunnel diode biased in the PDR region, close to the current 
peak due to the antisymmetrical conductance-voltage (G-V) characteristic at this bias point. 
The power levels at which the maximum negative conductances at the fundamental and 
second harmonic frequency occur are smaller than those for a unbiased tunnel diode. 

The differential conductance of a tunnel diode biased at the center of the NDR 
region has also been studied. From the results shown in Fig. 3 (a), an absolute negative 
conductance to dc has been obtained from a tunnel diode biased in the NDR region when 
the applied pump amplitude is within a small range. That is, the conductance of the 
resonant tunneling device will be negative at any frequency when the pump amplitude is 
within this small range. The value of the absolute negative conductance is approximately 
the same as that found in the NDR region. The magnitude of the negative conductances 
changes with the pumping power level (see Fig. 3). That the absolute negative 
conductance occurs for the tunnel diode biased in the NDR region asserts that oscillation 
can occur at any frequency if the pumping power is within the region that the negative 
conductance occurs. The magnitude of the negative conductance at the second harmonic is 
higher than those at dc, the fundamental and third harmonic frequencies due to the 
symmetrical G-V characteristic at this bias point (see Fig. 3 (b)). The variable absolute 
negative conductance observed can be used as the basis for oscillators and harmonic 
oscillators up to the cut-off frequency of the diode. The self-oscillation capability of a 
tunnel diode biased in the NDR region can, therefore, finds applications as biased self- 



Third International Sytnposium on Space Terahertz Technology Page 461 

oscillating mixers and frequency multipliers. The self-oscillating frequency multiplier and 
mixer discussed here do not require a large-signal rf pump. 

It should be noted that the self-oscillation at the fundamental generates its own 
harmonics using the nonlinearity of the NDR device; this will be referred to as the self- 
oscillating frequency multiplier. While the harmonic oscillator refers to the case that the 
NDR device oscillates at a particular harmonic frequency using the negative conductance at 
that harmonic frequency. It should be pointed out that the conversion gain (efficiency > 1) 
can be achieved from the biased self-oscillating frequency multiplier and mixer. 
Resonant Tunneling Device- With Antisymmetry in the I-V Characteristic 

From the nonlinear-circuit analysis results, an absolute negative conductance to dc 
can be found from a resonant tunneling device at zero bias when the applied pump power is 
within a small range (see Figs. 4 (a) and (b)). The value of the negative conductance is 
approximately the same as that found in the NDR region. As can be seen from the results 
in Figs. 4 (a) and (b), the magnitude of the negative conductance can be adjusted by 
varying the pump amplitude. This study verifies the work of Sollner et al. [3]. However, 
the voltage range over which the negative conductance occurs is strongly dependent on the 
pumping frequency. This region moves towards the more positive side of the power axis 
when the pumping frequency increases (see Figs. 4 (a) and (b)). The reason for this can, 
again, be explained with a equivalent circuit model (discussed for the tunnel diode in 
previous section). 

We found that the range of the applied pump power to obtain absolute negative 
conductance varies with the parasitics (series resistance and capacitance) of the device. As 
can be seen from Fig. 5, the pumping power region over which the negative conductance 
occurs move towards the more positive side of the power axis with increasing capacitance 
and series resistance of the device. This again can be seen from the equivalent circuit 
model of a NDR diode. When the series resistance and/or capacitance of the device 
increases, more voltage drops across the series resistance and, therefore, less power is 



Page 462 Third International Symposium on Space Terahertz Technology 

developed across the parallel circuit section of the equivalent circuit. It should be pointed 
out that the dependence of the conductance on the parasitics of the device is similar to that 
on the rf input frequency. 

The conductance of the resonant tunneling diode will be negative at any frequency 
when the pump amplitude is within a small range. Figures 4 (c) and (d) show the 
differential conductance at the second harmonic frequency versus pumping power and 
pump amplitude, respectively. The magnitude of the negative conductance at the second 
harmonic frequency is larger than the negative conductances at dc and at fundamental 
frequency. This is due to the symmetrical G-V characteristic of the resonant tunneling 
device. The variable absolute negative conductance observed can be used for oscillators up 
to the cut-off frequency of the diode. 

From these studies, one can expect to find absolute negative resistance whenever a 
material with negative differential conductance and an I-V curve that is antisymmetrical is 
driven with a pump of the right amplitude and frequency. The resonant tunneling device 
can also perform the same functions such as the self -oscillating frequency multiplier and 
mixer discussed for a tunnel diode if it is biased in the NDR region. It should be pointed 
out that the biased tunnel and resonant tunneling device (in the PDR region) requires less 
pumping power to achieve self-oscillation than the unbiased device. The biased self- 
oscillation tunnel diode and resonant tunneling frequency multipliers and mixers have the 
intrinsic capability of conversion gain. For the unbiased oscillator operations, it should be 
noted that little in the way of negative conductance or dynamic range has been sacrificed 
with this operation, and the advantage of operating with zero DC bias voltage has been 
gained. It should be further noted that the increase in negative conductance at a specific 
frequency (depending upon the operating point) could simplify frequency selection for 
oscillator designs. 

EXPERIMENTAL RESULTS 



Third International Symposium on Space Terahertz Technology ^ge 463 

During the experimental measurements, it was observed that the dc I-V 
characteristic of the NDR device is very strongly dependent on the power level of the rf 
input signal. The dc I-V characteristics of the NDR devices (with and without the 
antisymmetrical I-V characteristics) measured at different rf input power levels are shown 
in Figs. 6 and 7, respectively. The dc I-V characteristics of the tunnel diode measured at 
different rf input frequencies are shown in Fig. 8. Based on these results, the dc I-V 
characteristics of the NDR device changes dramatically with the increasing input power 
level and frequency. The dependence of the I-V characteristics of the NDR device on the 
frequency of the rf input signal can easily be seen from the equivalent circuit model of the 
NDR device. The frequency dependence of the impedance across the parallel circuit section 
of the equivalent circuit results in the power dependence of the I-V characteristics. The 
power dependence of the negative conductance complicates the dependence of the negative 
conductance on the frequency. The nonlinear circuit analysis was used to theoretically 
verify this observation. The I-V curves measured at different pumping power levels as 
shown in Figs. 6 and 7 and the I-V curves measured at different pumping frequencies as 
shown in Fig. 8 compare favorably to the simulation results. 

Based upon this study, the design, operation, and performance of the NDR 
frequency multiplier, self-oscillating frequency multiplier and mixer and harmonic oscillator 
can be complicated. For a constant rf input frequency, the biasing and pumping conditions 
and output power of the self oscillation of the NDR device vary with the rf input power 
level. In addition, the onset of self oscillation of a NDR device biased in the NDR region 
also depends upon the rf input power level. For example, the self oscillation can be 
suppressed by changing the rf input power level (tuning the I-V characteristic) in the 
frequency multiplication operation of a NDR device. However, this will cause the 
conversion efficiency of the device to change as well since the nonlinearity is not the same. 
In addition, the self-oscillation with a constant rf input frequency may disappear for a given 
dc bias depending upon the rf input power level for the self-oscillating frequency 



Page 464 Third International Symposium on Space Terahertz Technology 

multiplier, mixer and harmonic oscillator applications. Determination of the biasing and 
operating conditions and performance of the NDR frequency multiplier, self-oscillating 
frequency multiplier and mixer, and harmonic oscillator, therefore, requires complete 
information of the I-V characteristics at different rf input frequencies and power levels. 
This can be accomplished by extensive simulations of the NDR device under different 
pumping conditions using the modified large- and small-signal nonlinear circuit analysis 
program as mentioned above. 

A NDR biased in the NDR region can be used for the self-oscillating frequency 
multiplier and mixer. Both the self-oscillating frequency multiplier and mixer have been 
successfully demonstrated using a tunnel diode biased in the NDR region. The results 
from a self-oscillating frequency multiplier can be seen in Fig. 9. The highest tripling 
efficiency has been obtained at the center of the NDR region while the highest doubling 
efficiency has been obtained at the edges of the NDR region of the tunnel diode as 
expected. This is due to the I-V characteristic being antisymmetrical when biased at the 
center of the NDR region, and the I-V characteristic being almost symmetrical when biased 
close to the current peak. It should be pointed out that the circuit used does not allow 
independent tuning of the harmonics. 

APPLICATIONS 

The wide use of resonant tunnel devices is limited, to a considerable extent, by the 
low level of their output power. Power combining techniques are employed to increase the 
output power of the resonant tunneling devices. The device-grid array approach is a 
potentially attractive way to spatially combine the output power of large numbers of 
resonant tunneling devices. In this approach, a grid is monolithically integrated with 
thousands of devices thereby overcoming the power limitations of a single device since the 
power is distributed among the many devices making possible watt- level CW output power 
throughout the microwave and millimeter- wave region [8,9]. This kind of array can find 
applications as a high frequency, high power solid-state rf power source. All the 



Third International Symposium on Space Terahertz Technology Page 465 

interconnections of the high- and low-frequency leads of each port of each device 
(especially, three-terminal devices) present an extremely difficult problem for the 
development of such arrays. The demonstration of novel unbiased oscillators is most 
useful for the development of monolithic planar wafer-scale device arrays since no dc bias 
lines are required, which greatly simplify the grid design. 

In addition, the pumping power for a device grid is significantly higher than that for 
a single device (proportional to the number of the devices). Therefore, it is important to 
minimize the amount of input power required to pump each individual device. Based upon 
the theoretical and experimental work which have been performed in this study, the 
possibility of biasing a NDR diode to minimize the amount of power required to pump each 
individual diode into the desired operation point has been verified. The bias lines can be 
easily employed in the design of diode grid (two terminal device grid) to provide dc bias 
and minimize the required pumping power [9]. In addition, frequency multiplication and 
mixing with gain can be obtained from these biased NDR diodes as discussed in this paper. 

CONCLUSION 

This work employs a modified large- and small- signal nonlinear-circuit analysis [2] 
to verify the previous work of Sollner et al. using a simple mathematical model [3]. The 
absolute negative conductance can be obtained from an unbiased resonant tunneling device 
when the applied pump power is within a small region. The variable absolute negative 
conductance can be used as the basis for oscillators up to the cutoff frequency of the 
device. Furthermore, a NDR device biased in the NDR region can be used as the basis for 
the self-oscillating frequency multiplier and mixer. The biased self-oscillating frequency 
multiplier and mixer can achieve conversion gain (efficiency > 1). These functions have 
been experimentally demonstrated in this work. The advantage of a unbiased oscillator 
using a resonant tunneling device comes from the fact that the negative conductance can be 
adjusted by varying the pump amplitude-a very useful circuit property. In addition, the 
negative conductance is larger at even harmonic frequencies which could simplify 



Page 466 Third International Symposium on Space Terahertz Technology 

frequency selection of an oscillator design based upon this effect. The advantage of 
operating with zero dc bias voltage is also gained. Through this study, the power 
dependence of the negative conductance of a NDR device on the rf input signal has been 
observed. The biasing and pumping conditions and performance of the frequency 
multiplier, self-oscillating frequency multiplier and mixer, and harmonic oscillator requires 
complete information of the I-V characteristics of a NDR device at different input 
frequencies and power levels. This information can be obtained using the modified large- 
and small-signal nonlinear-circuit analysis as discussed in this paper. 

References 

[1] H. Siegel, A. R. Kerr, and W. Hwang, "Topics in the Optimization of MM Wave 

Mixers," NASA Technical Paper #2287, 1987. 
[2] R. J. Hwu and N. C. Luhmann, Jr., "Quantum Well Diode Frequency Multiplier 

Study," Second International Symposium on Space Terahertz Technology, 

Proceedings, pp. 226-237, 1991. 
[3] T. C. L. G. Sollner, E. R. Brown, and W. D. Goodhue, Picosecond Electronics 

and Optoelectronics II, Editors: F. J. Leonberger, C. H. Lee, F. Capasso, and H. 

Morkoc, Springer-Series in Electronics and Photonics, Vol. 24, pp. 102-108, 

1987. 
[4] P. D. Batelaan and M. A. Frerking, 13th Int'l Conf. on Infrared and Millimeter 

Waves, Proceedings, 1988. 
[5] T. C. L. G. Sollner, E. R. Brown, W. D. Goodhue, and C. A. Correa, J. Appl. 

Phys., Vol. 64, P. 4248, 1988. 
[6] D. N. Held and A. R. Kerr, "Conversion Loss and Noise of Microwave and 

Millimeter- Wave Mixers: Part 1 - Theory," IEEE Trans, on Microwave Theory and 

Tech., MTT-26, PP. 49-55, 1978. 
[7] P. P. Huang, "I-V Characterization of Negative Resistance Device by Microwave 

Reflection Coefficients," M.S. thesis, UCLA, pp. 49-51, 1989. 
[ 8 ] D. B. Rutledge and S. E. Schwarz, "Planar Multimode Detector Arrays for Infrared 

and Millimeter Waves Applications," IEEE J. Quantum Electronics, QE-17, pp. 

407-414, 1981. 
[9] R. J. Hwu, C. F. Jou, N. C. Luhmann, Jr., M. Kim, W. W. Lam, Z. B. Popovic, 

D. B. Rutledge, "Array Concepts for Solid-State and Vacuum Microelectronics 



Third International Symposium on Space Terahertz Technology Page 467 

Millimeter- Wave Generation," IEEE Trans, on Electron Device, ED-36, No. 11, 
pp. 2645-2650, 1989. 

Figure Captions 

Fig. 1 The I-V curves of a (a), tunnel diode and a (b). resonant tunneling device used in 

the nonlinear circuit analysis, (c). The equivalent circuit of a NDR device. 

Fig. 2 The differential conductance of a tunnel diode biased in the PDR region (close to the 

current peak) at the (a), dc, (b). fundamental, and (c). third harmonic frequency 

from the nonlinear circuit analysis. 
Fig. 3 The differential conductance of a tunnel diode biased at the center of the NDR 

region at the (a), dc, and (b). second harmonic frequency from the nonlinear circuit 

analysis. 
Fig. 4 The differential conductance at dc versus (a), pump power level, and (b). pump 

amplitude of a zero-biased resonant tunneling device. The differential conductance 

at the second harmonic frequency versus (c). pump power level and (d). pump 

amplitude of a zero-biased resonant tunneling device from the nonlinear circuit 

analysis. 
Fig. 5 The differential conductance at dc of a zero-biased resonant tunneling device with 

different (a), capacitance values of 1 pF and 1 fF with three rf input frequencies of 

0.7, 12, and 90 GHz, and (b). series resistance values of 12.5 Q, 625 Q, and 1.25 

kQ with an rf input frequency of 0.7 GHz from the nonlinear circuit analysis. 
Fig. 6 The measured I-V curves of a tunnel diode with different rf input power levels at 

two rf input frequencies of (a). 0.7 GHz, and (b). 2.5 GHz. 
Fig. 7 The measured I-V curves of a NDR device (with an antisymrnetrical I-V 

characteristic) with two different rf input power levels at an rf input frequency of 

10 MHz. 
Fig. 8 The measured I-V curves of a tunnel diode with different rf input frequencies at an 

rf input power level of 2 mW. 
Fig. 9 The ratio of the output power to the fundamental power of a self-oscillating 

frequency multiplier using the tunnel diode of Fig. 6 versus different bias points in 

the NDR region with a rf input frequency of 0.7 GHz. 



Page 468 



Third International Symposium on Space Terahertz Technology 



0.05 

0.04 

0.03 

0.02- 

0.01 - 

0.00- 

-0.01 

-0.02 

-O.03 

-0.04 - 

-0.05 




-1.0 -0.8 -0.6 -0.4 -0.2 0.0 0.2 0.4 0.6 0.8 1.0 



V (V) 



Fig. 1 (a) 



< 

E 




-i — |— h — | — i — i — i — | — i — | — i — | — i — i — r— , — i — | — i — | — t — | — i- 
.6-0.5-0.4-0.3-0.2-0.1 0.0 0.1 0.2 0.3 0.4 0.5 0.6 



Fig. 1 (b) 



V (V) 



R. 



G(V) 



$ 



Fig. 1 (c) 



Third International Symposium on Space Terahertz Technology 



Page 469 



0.10 




100 



Fig. 2 (a) 



0.00 



-0.01 



-0.02 - 



-0.03 



-0.04 



-0.OS 



-0.06 



Fig. 2 (b) 



Power (mW) 




Power (mW) 




-a Gdc - 0.7 GHz 

» Gdc - 2.5 GHz 

-» Gdc - 5.S GHz 

-o Gdc - 8.5 GHz 



Grf1 - 0.7 GHz 
Grfl - 2.5 GHz 
Grfl - S.5 GHz 
Grfl - 8.5 GHz 



-• Grf3 -0.7 GHz 

-a Grf3 - 2.5 GHz 

-* Grf3 - 5.5 GHz 

-o Grf3 - 8.5 GHz 



Fig. 2 (c) 



Power (mW) 



Page 470 



Third International Symposium on Space Terahertz Technology 



0.020 



tn 



o 

•D 

o 



0.018- 
0.016 
0.014- 
0.012- 
0.010- 
0.008 - 
0.006 
0.004 - 
0.002 - 
0.000 4 
-0.002 





100 



Power (mW) 



Gdc - 0.7 GHz 
Gdc - 2.5 GHz 

Gdc - 5.5 GHz 
Gdc - 8.5 GHz 
Gdc - 20 GHz 



Fig. 3 (a) 



tn 



O 




Power (mW) 



Grf2 - 0.7 GHz 
Grf2 - 2.5 GHz 
Grf2 - 5.5 GHz 
Grf2 - 8.5 GHz 



Fig. 3 (b) 



Third International Symposium on Space Terahertz Technology 



Page 471 




Pumping Power (mW) 




Fig. 4 (a) 



Fig. 4(b) 



0.6 GHz 
93 GHz 
ITHz 
1.5 THz 
2.2 THz 






12.00- 
11.00- 
10.00- 

9.00- 

8.00- 

7.00 

6.00- 

5.00- 

4.00 

3.00 

2.00 

1.00 

0.00 -* 
-1.00- 
-£00 



.1 



«sS££§SF 




a~2=fi^o— q^s=SSq< 




Pumping Power (mW) 



Voltage (V) 



Fig. 4 (c) 



Fig. 4 (d) 



Page 472 



Third International Symposium on Space Terahertz Technology 



E 

u 

O 



E 



u 
(3 




Fig. 5 (a) 

10.00' 



9.00 H 
8.00 
7.00- 
6.00- 
5.00 
4.00- 
3.00- 
2.00- 
1.00 
0.00 
-1.00 



;1 



Pump Power (mW) 




1 ' ■ i 

1 



■ i i i i 



-o 90 GHz, lpF 

-• 90 GHz, 10 fF 

■^ 12 GHz, 1 pF 

12 GHz, 10 fF 

0.7 GHz, 1 pF 
0.7 GHz, 10 fF 



-• Gdc-Rs=12.5n 

-a Gdc - Rs=625 £2 

-a Gdc-Rs=1.25KQ 



1 



Power (mW) 



Fig. 5 (b) 



Third International Symposium on Space Terahertz Technology 



Page 473 



0.003 



0.002 - 



< 0.001 - 



c 
o 



0.000 - 



-0.001 




0.002 i ■ T*-> 1 ' 1 ■ 1 ' 1 ' 1 ' 1 "- 

-0.2 -0.1 0.0 0.1 0.2 0.3 0.4 0.5 0.6 



Voltage (V) 



Fig. 6 (a) 

0.003 



0.002 



< 0.001 - 



c 

4) 

o 



0.000 - 



-0.001 - 



-0.002 H r 




Voltage (V) 



■a — 5nW 
-* — 2.96 mW 
49.8 mW 



4^W 
-* — 0.56 mW 
16.7 mW 



Fig. 6 (b) 



Page 474 



Third International Symposium on Space Terahertz Technology 



< 



G 
* 

L. 

3 

o 




10 mW 




OmW 



— i — i — ■ — i — ■ — i — i — i — ■ — i — ■ — i — i — i — i — i — ■ — i — r- 
.5 -1.2 -0.9 -0.6 -0.3 -0.0 0.3 0.6 0.9 1.2 1.5 



Voltage (V) 



Fie. 7 



Third International Symposium on Space Terahertz Technology 



Page 475 



15.00 



< 
E 



c 
© 

3 

o 



10.00 



5,00- 



0,00- 



-5.00 - 



-10.00- 




-15.00 | ■ i • i — i 1 1 | i | i i 

-0.2 -0.1 0.0 0.1 0.2 0.3 0.4 0.5 0.6 



-o I-.2 GHz 

-* — 1-3.62 GHz 

-o — 1-8.45 GHz 

-® — l-t1.95GHz 



Voltage (V) 



Fig. 8 



Page 476 



Third International Symposium on Space Terahertz Technology 



-o 







0.1 0.15 0.16 0.19 0.21 0.22 0.28 



Bias (V) 



P2/P1 
♦ P3/P1 



Fig. 9 



Third International Symposium on Space Terahertz Technology p a g e 477 

Modeling, Design, Fabrication, and Testing of InP Gunn /&03-OO 
Devices in the D-band (110 GHz - 170 GHz) l f- / ^ 

R. Kamoua 2 , H. Eisele, J. R. East, G. I. Haddad, 
G. Munns, and M. Sherwin 



Solid-State Electronics Laboratory 

Department of Electrical Engineering and Computer Science 

The University of Michigan, Ann Arbor, MI 48109-2122 



Abstract 

The development of fundamental Gunn sources for D-band frequencies requires improve- 
ments of doping profiles, processing technology, and circuit design. We have developed a 
' technology for fabricating InP Gunn diodes using an InGaAs etch-stop layer between the 
InP substrate and the device layers. The epitaxial layers were grown by CBE. During 
device processing, the substrate is completely removed. Substrateless devices with an n + 
InGaAs cap layer are expected to have reduced contact and series resistances, and skin 
effect losses. This technology gives better uniformity and control of the device geometry 
across the processed chip. 

InP Gunn devices with a 1.7 fim long active region (doping : 9 x 10 15 cm -3 ) have been 
mounted on copper heat sinks. Two tapered leads were then bonded to the diode and to 
four quartz standoffs. As a preliminary result, an output power of 13 mW at 82 GHz was 
obtained. Based on these RF measurements, we determine appropriate material parameters 
to be used in the Ensemble Monte Carlo model. Subsequently, we use this model to design 
and evaluate the performance of InP Gunn Devices for D-band frequencies. Using the same 
technology, we are currently processing Gunn devices with a 1/j.m long active region for 
operation at higher frequencies. 



'This work was supported by the Center for Space Terahertz Technology under Contract No. NAGW- 
1334 

2 R. Kamoua is currently with the Department of Electrical Engineering, State University of New York 
at Stony Brook, Stony Brook, NY 11794-2350 



Page 478 Third International Symposium on Space Terahertz Technology 



1 Introduction 

Gunn devices are widely used as local pump oscillators in the W-band frequency region. 
At these frequencies, the technology is well developed. In particular at 94 GHz, GaAs and 
InP Gunn devices are available with very good performance. At frequencies above 100 
GHz, the situation is quite different. Although there is a strong demand for sources at 
these frequencies, fundamental Gunn devices are generally not available. There are two 
reasons. First, most of the research effort has been focused at 94 GHz. Second, extending 
the fundamental oscillation frequency of Gunn devices into the submillimeter region is 
exceedingly difficult because the Gunn effect is being pushed to its high frequency limit. 

This paper discusses a systematic approach toward the modeling, design, fabrication, 
and testing of InP Gunn devices in the D-band region. Compared with GaAs, InP material 
parameters are more favorable for operating Gunn devices in the D-band. The approach 
taken in this work is both experimental and theoretical. Experimentally, the conventional 
processing technology is improved by incorporating etch-stop layers in the wafer design. 
The RF results obtained from devices fabricated using this technology are then used to 
improve the accuracy of the theoretical model. Subsequently, the improved model is used 
to design the optimum Gunn structure for the desired frequency of operation. 

2 Simulation Model 

The self consistent Ensemble Monte Carlo model is used to estimate the performance of 
InP Gunn devices at high frequencies. This model is an extension of the one particle 
Monte Carlo technique [1]. In order to describe the transport process in a structure with 
nonuniform doping and with time varying fields, an ensemble of electrons needs to be 
simulated simultaneously. The electric field has to be updated regularly since it is evolving . 
as the electrons redistribute in the structure. The analysis is carried out assuming the 
device behavior is mainly one dimensional which is justified for two terminal devices. A 
diode structure of length L is divided into cells of equal length (Ax = 50 A). Any attributes 
of the electrons are averaged over each cell and assigned to the midway position of the cell. 
The cell size should be smaller than the smallest Debye length in the structure which occurs 
at the highly doped regions. 

The simulation algorithm consists of monitoring the evolution in real space and momen- 
tum space of an ensemble of electrons. The simulation time is partitioned into time steps 
(At = 5xl0 -15 sec) each terminated by a call to a Poisson solver in order to update the 
field. In each time step every electron is submitted to successive free flights terminated by 
a scattering process which is selected using a random number generator. Electrons cross- 
ing cell boundaries are temporarily stopped at that boundary and then resumed with the 
electric field in the new cell. An analogous procedure is followed when it is time to update 
the electric field and the electron is in the middle of a free flight. In this case the remaining 
flight time is stored, and the flight is resumed when all other electrons are simulated for one 
time step, the carrier density is calculated, and the electric field is updated. The predicted 
performance of a particular Gunn structure is estimated by applying an RF voltage across 
the device and simulating the current response over many RF periods (about 10). The 
resulting particle current density is Fourier analyzed and the fundamental component is 



Third International Symposium on Space Terahertz Technology 



Page 479 



N + 

3 x 10 17 cm- 3 

InP 


9 x 10 15 cm -3 
InP 


3 x 10 17 cm- 3 
InP 



O.ltim 1.7/xm 0.2pm 

Figure 1: Dimensions and doping profile of Gunn structures. 

used to determine the device admittance. The RF output power is estimated by consider- 
ing the Gunn device in a resonant circuit represented by a load resistance and a resonating 
inductance. A series resistance is included in the equivalent circuit that takes into account 
effects of contact resistances, any substrate resistances, and skin effect losses. 

The accuracy of the Monte Carlo model is strongly dependent on the accuracy of the 
material parameters used. Unfortunately, one finds a wide range of values that are being 
used in the literature. In particular, some of the material parameters that are important 
to the Gunn effect have the following range of values ([2, 3, 4, 5]): 



r - L valley separation (eV) 

L valley effective mass ratio ( j^- ) 

V - L coupling constant (xlO 9 eV.cm -1 ) 

r — X coupling constant (xlO 9 eV.cm -1 ) 



0.4 


«=> 


0.832, 


0.26 


<^> 


0.4, 


0.1 


<=3> 


2.5, 


0.43 


<=> 


1.0. 



There is more than an order of magnitude uncertainty in the T to L intervally coupling 
constant. In the next section, the material parameters that yield the best agreement with 
the RF measured results will be identified. 



3 Method for Extracting Accurate InP Material Parame- 
ters for a 3- Valley Nonparabolic EMC Model 

The appropriate material parameters are determined by comparing measurements at high 
frequencies with results predicted by the model. The structure considered for comparison 
is shown in Figure 1. It has 1.7 /xm long active region doped at 9xl0 15 cm -3 , a 0.1/xm 



17 cm" 3 . 



cathode region doped at 3x10 cm and a 0.2 /im anode region doped at 3x10 

An InP wafer with this structure has been processed. The relevant fabrication technol- 
ogy will be described later. Diodes with various sizes have been mounted on copper heat 
sinks. Tapered ribbons were then used to bond a diode to four quartz standoffs. A 40 /xm 
diode was tested in a W-band resonant cavity with the following results: 



Page 480 Third International Symposium on Space Terahertz Technology 



Bias voltage 4.0 V to 4.5 V, 

Oscillation frequency 82 GHz, 

output power 13 mW, 

conversion efficiency 1 %, 

DC current 350 mA. 

The structure shown in Figure 1 is simulated using the model described above. The 
DC bias is 4.5 V, the RF voltage is 1.0 V, and the operating temperature is assumed 
to be 400 K. The material parameters are considered to be accurate if oscillations at 82 
GHz are predicted with a performance comparable to experimental results. The starting 
combination of parameters is listed in Table 1 and is referred to as the initial parameter 
set. No oscillations occurred with this set of values for frequencies ranging from 75 GHz to 
120 GHz. It appears that the T - L intervalley energy separation of 0.832 eV is too large. 
As a result, the following modifications to the initial set are considered: 



• 



Case 1. The same parameters as the initial set are used except 17 — L valley separation 
= 0.6 eV. No oscillations occurred at 82 GHz. 



• Case 2. The same parameters as the initial set are used except T — L valley separation 
= 0.4 eV. No oscillations occurred at 82 GHz. 

• Case 3. The same parameters as the initial set are used except T — L energy separation 
= 0.4 eV and T — X energy separation = 0.7 eV. No oscillations occurred at 82 GHz. 



• 



• 



Case 4. The same parameters as the initial set are used except L valley effective mass 
ratio = 0.4 and X valley effective mass ratio = 0.4. No oscillations occurred at 82 
GHz. 

Case 5. The same parameters as the initial set are used except T — L coupling constant 
= 1.0 x 10 9 eV.cm -1 . No oscillations occurred at 82 GHz. 



• Case 6. The same material parameters as the initial set are used except T — L energy 
separation = 0.5 eV, T - X energy separation = 0.8 eV, T — L coupling constant = 
1.0 XlO 9 eV.cm -1 and T - X coupling constant = 1.0 xlO 9 eV.cm -1 . Very weak 
oscillations were obtained. The efficiency was 0.032 % and the output power was 0.22 
mW. 

• Case 7. The same parameters as the initial set are used except T — L energy separation 
= 0.45 eV, T — X energy separation = 0.775 eV, L valley effective mass ratio = 0.4 eV, 
X valley effective mass ratio = 0.4 eV, T — L coupling constant = 1.0 xlO 9 eV.cm" 1 , 
T — X coupling constant = 1.0 xlO 9 eV.cm -1 and acoustic deformation potential 
= 5 eV. Oscillations were obtained at 82 GHz even though the DC bias was 4.0 V 
instead of 4.5 V. With 1.0 V RF voltage, the predicted output power for a 40 fj,m 
diode was 7.2 mW with 0.52 % efficiency. 

The last parameter set appears to be promising and was considered in more detail. At an 
RF voltage of 1.5 V, the predicted efficiency was 1.1 % and the predicted output power 



Third International Symposium on Space Terahertz Technology 



Page 481 



Energy Separation (eV) 


r-L 


0.832 


r-x 


1.5 


Effective Mass (g) 


r 


0.082 


L 


0.26 


X 


0.325 


Nonparabolicity factor ([eV] -1 ) 


r 


0.83 


L 


0.23 


X 


0.38 


Intervalley Coupling Constant 
(K^eV.cm- 1 ) 


r-L 


0.506 


r-x 


0.498 


L-X 


0.468 


L-L 


0.575 


x-x 


0.28 


Acoustic Deformation Potential (eV) 


r 


7 


L 


7 


X 


7 


LO Phonon Energy (eV) 


r 


0.043 


L 


0.0423 


X 


0.0416 


Static Dielectric Constant 




12.61 


Optical Dielectric Constant 




9.61 



Table 1: InP initial material parameter set. 



Page 482 



Third International Symposium on Space Terahertz Technology 



n: 



1.5 - 



0.5 - 



60 



I 


1 1 " 


- 


/ 


/ • Vrf = 05 V \ 
J a Vrf = 10 V 
/ ♦ Vrf - 15 V 


y - 


*r i 


i i 





80 



100 



120 



140 



Frequency [GHz] 



Figure 2: Efficiency vs. frequency for a 1.7 /zm long InP Gunn device doped at 9xl0 15 
cm" 3 . Diameter = 40 /im, Vdc = 4.0 V, T = 450 K. 

was 14.9 mW for a 40 /xm diode. The necessary load resistance for matching the diode 
was 6.5 SI. The simulated DC current density has a value of 2.7xl0 4 A. cm -2 which results 
in a current of 340 mA. The simulated device performance is in good agreement with the 
measured RF results. In summary, we found it necessary to decrease the intervalley energy 
separation and increase the electron effective mass in the upper valleys and the intervalley 
deformation potentials in order to obtain oscillations at 82 GHz. The actual device was 
operated at higher temperature than room temperature. This suggests that the lower values 
for the intervalley separation and the higher values for the deformation potentials might be 
caused by temperature effects on the band-structure. The last parameter set will be used 
to analyze various InP Gunn structures. In the next section, the 1.7 ^m structure will be 
considered in more detail. 



4 Simulation of 1.7 ^m InP Gunn Devices 

The results presented in this section correspond to a DC bias voltage of 4.0 V and an 
operating temperature of 450 K. For comparison purposes, the load resistance was adjusted 
so that the device area is 40 /im. Figure 2 shows a comparison of the conversion efficiency as 
a function of frequency for three RF voltage amplitudes: 0.5 V, 1.0 V and 1.5 V. The peak 
efficiency for a 1.5 V RF voltage is 1.86 % at 100 GHz. The corresponding comparison of 
the output power is shown in Figure 3. A peak power of 23 mW at 100 GHz was obtained 
for the case where the RF voltage is 1.5 V. 

In general, the operating temperature is 80 K to 90 K above the room temperature. 
Therefore, it is possible to extract more power by either increasing the device area or the 
DC bias or a combination of both. Increasing the device area is limited by thermal effects 



Third International Symposium on Space Terahertz Technology 



Page 483 






25 



20 - 



15 - 



10 - 



5 - 



60 



I 


, , , 




1 


/ • Vrf = 05 V \ 
/ o Vrf = 1.0 V \ 


- 


1 


/ ♦ Wf - 15 V 


\ - 


- ,„. -.1 


i i 





80 



100 



120 



140 



Frequency [GHz] 



Figure 3: Power vs. frequency for a 1.7 /im long InP Gunn device doped at 9xl0 15 cm -3 . 
Diameter = 40 /im, Vdc = 4.0 V, T = 450 K. 

and by the minimum load provided by the resonant circuit. On the other hand, increasing 
the DC bias voltage is limited by thermal effects and breakdown due to the large electric 
field at the anode. For example, increasing the diode diameter from 40 ^m to 60 \xxa. results 
in 50 mW output power at 100 GHz with an RF voltage of 1.5 V. The required matching 
load is 2 Q compared to 3.75 $7 for the smaller device and the temperature increase is 120 
K compared to 86 K. 

5 Simulation of 1.0 /im InP Gunn Devices for Operation 
in D-band Frequency Region 

The 1.7 fj,m device considered in the previous section had an optimum operating frequency 
around 100 GHz. For higher frequencies in D-band, structures with shorter active regions 
need to be considered. In this section, simulation results of a 1 /jm long InP device are 
presented. 



5.1 Flat Doping Profile 

Two flat doping profiles in the active region lxlO 16 cm -3 and 2xl0 16 cm -3 are considered. 
Figure 4 compares the corresponding efficiency as a function of frequency for a DC bias of 
4.0 V, an RF voltage of 0.5 V, and an operating temperature of 450 K. The structure with 
2xl0 16 cm* 3 doping results in a higher efficiency at all frequencies. The output power into 
a 2 ft load is shown in figure 5. Near 160 GHz, the predicted output power from the highly 
doped structure is more than 5 times higher than the output power from the low doped 
structure. However, for a meaningful comparison, thermal effects and current density levels 



Page 484 



Third International Symposium on Space Terahertz Technology 



0.7 







0.0 



120 140 160 180 200 220 

Frequency [GHz] 

Figure 4: Efficiency versus frequency for a 1.0 ^m long InP Gunn device. Vdc = 4.5 V, Vrf 
= 0.5 V,^ = 2fl,T = 450 K. 

need to be considered. 

The DC current density in the structure with a lower doping is about 3.95 xlO 4 A. cm -2 
at 160 GHz whereas it is about 6.7xl0 4 A. cm -2 for the other structure. The device with 
the higher doping has a very large current density which results in an operating temperature 
approaching the limit for a 30 /xm diameter device. For smaller diodes, bonding becomes 
very difficult. Therefore, there is a need for improving the efficiency while keeping the 
current density from increasing rapidly. It will be shown in the next section that a graded 
doping profile in the active region provides such an improvement. 

5.2 Graded Doping Profile 

This section examines methods of improving the efficiency of InP Gunn devices and opti- 
mizing the design for operation around 160 GHz. In addition to the structure with a flat 
doping of lxlO 16 cm -3 , three linearly graded doping profiles are considered: 





Ni (xlO 16 cm -3 ) 


N 2 (xlO^cm" 3 ) 


Structure 1 


1.0 


1.0 


Structure 2 


0.8 


1.5 


Structure 3 


0.8 


3.0 


Structure 4 


0.8 


4.0 



In the above table, Ni is the doping density in the active region at the cathode side and 
N2 is the corresponding doping at the anode side. The results presented in this section 
correspond to a bias of 4.0 V, an RF voltage of 1.0 V, an operating temperature of 450 K, 



Third International Symposium on Space Terahertz Technology 



Page 485 




220 



Frequency [GHz] 



Figure 5: Power versus frequency for a 1.0 /im long InP Gunn device. Vdc 
0.5 V, R L = 2 ft, T = 450 K. 



4.5 V, Vrf = 



and a matching load of 2 ft. 



Figure 6 shows the efficiency versus frequency for the three structures. The efficiency 
is increasing as the doping profile becomes steeper. At 180 GHz, structure 3 results in 
2 % efficiency which is twice the efficiency obtained from structure 1. A comparison of the 
output power is shown in Figure 7. The optimum frequency for power generation is 160 
GHz where structure 3 resulted in 73.4 mW compared with 8.6 mW for structure 1. The 
DC current density in structure 3 is 5.18xl0 4 A. cm -2 compared with 3.84xl0 4 A. cm -2 
in structure 1 at 160 GHz. This increase in the current density is much smaller than 
the corresponding increase in the 2xl0 16 cm -3 doped structure. Structure 4 has higher 
efficiencies than structure 3 but the current is also higher. For diodes with a mesa diameter 
of 30 fim or larger, structure 3 is likely to be the optimum design for operation at 160 GHz. 

The graded doping profile results in a higher electric field near the cathode and a lower 
field near the anode due to electron diffusion toward the lower doped region. This change in 
the field has two consequences: first, a higher cathode field results in a larger fraction of the 
electrons transferring to the upper valleys, second, a lower anode field permits application of 
a larger DC bias without breakdown. The electric field in structure 1 peaks at 125 kV.cm -1 
near the anode side whereas the field in structure 4 is less than 100 kV.cm -1 at the anode. 

In addition to improving the efficiency and reducing the field at the anode, a graded 
doping profile provides a modest increase in the current density. The graded doping profile 
can provide the same performance as the flat doped profile with a smaller current density. 
The reason is that the higher fraction of the electron population transferring to the upper 
valleys in the graded structure results in a reduction of the average velocity. This is not 
the case for the structure with a flat doping where an increase in the carrier density does 
not alter the distribution of the field across the structure. Figure 8 compares the current 



Page 486 



Third International Symposium on Space Terahertz Technology 



2.5 



1.5 - 






0.5 




Frequency [GHz] 



200 



Figure 6: Efficiency versus frequency for a 1.0 //m InP Gunn device. Vdc = 4.0 V, Vrf '= 
1.0 V, RL = 2 fi, T = 450 K. 



CD 



80 
70 
60 
50 
40 
30 
20 
10 




- 


1 1 1 


- 


- 


/• flat 10 x 10" on" 1 \ 

/ ON2 = 15«10* cm'' \ 


- 


-/ 


♦ N2 - 3 x 10" cm" 1 \ 


- 


- . 


i i i 


\ - 



120 140 160 180 

Frequency [GHz] 



200 



Figure 7: Power versus frequency for a 1.0 //m InP Gunn device. Vdc = 4.0 V, Vrf = 1.0 
V, RL = 2 n, T = 450 K. 



Third International Symposium on Space Terahertz Technology 



Page 487 



'$2 
•3 



7.5 



6.5 



5.5 



4.5 



3.5 



2.5 




- 4.5 



3.5 



125 



250 275 
Time (ps) 



625 



o 

3 



Figure 8: Comparison of the current density in structure 1 and structure 3 at 160 GHz. 
Vdc = 4.0 V, Vrf = 1.0 V, T = 450 K. 

waveforms in one RF period as obtained from structure 1 and structure 3. The current 
density in structure 3 resembles a pulse 180 degrees out of phase with the RF voltage 
whereas in structure 1, it is more sinusoidal and is not perfectly out of phase. This shows 
that structure 3 is more efficient and the space charge layers are more developed. 

To verify that the highly doped side of the active region should be at the anode side for 
best performance, structure 3 was considered with the opposite polarity. Figure 9 shows a 
comparison of the output power as a function of frequency. The structure with the doping 
decreasing toward the anode contribute to much smaller output power levels when compared 
with structure 3. An examination of the electric field revealed a peak electric field near the 
anode close to 200 kV.cm -1 compared with 100 kV.cm -1 in structure 3. In addition the 
current density is higher, at 160 GHz it has a average value of 6.9 XlO 4 A. cm -2 compared 
with 5.2xl0 4 A. cm -2 in structure 3. 



6 InP GUNN DEVICE TECHNOLOGY 

Among the many techniques used for fabricating Gunn devices, the most common are based 
on the Integral Heat Sink process (IHS) [7, 8] or the flip-chip process [9]. In the IHS process, 
the heat sink is formed as an integral part of the diode. The wafer front side is metallized 
and plated with copper, silver, or gold to a thickness of several thousands of an inch. The 
substrate is chemically or mechanically thinned to a thickness of 10 fim to 15 /xm . Next, 
ohmic contacts are metallized and standard photoresist techniques are used to define the 
mesas. Individual chips are then mounted in standard packages. In the flip-chip process 
the mesas are defined on the epitaxial side. The mesa chip is then flipped and mounted on 
a heat sink pedestal. Finally, the substrate is thinned to about 100 /zm . 



Page 488 



Third International Symposium on Space Terahertz Technology 



100 c 



cu 



0.1 




200 



Frequency [GHz] 



Figure 9: Comparison of the power versus frequency for structure 3 with different bias 
polarities . Vdc = 4.0 V, Vrf = 1.0 V, RL = 2 Q, T = 450 K. 



These two processing techniques have been used successfully for fabricating Gunn de- 
vices in the W-band. At higher frequencies, it is necessary to reduce further the substrate 
thickness to minimize the series resistance. In addition, the smaller size of the mesas 
presents new challenges to the bonding procedure. In this chapter a new fabrication tech- 
nology allowing the complete removal of the substrate is developed. This process is an 
extension of the integral heat sink technique with the additional step of plating the top 
contact to facilitate bonding. 

Figure 10 shows the epitaxial layers of an unprocessed InP wafer. The different layers, 
starting from the n + doped substrate consist of 

• an n + InP substrate, 

• a 0.5 /xm n + InGaAs layer doped at 2.0xl0 18 cm -3 , 

• a 0.6 /xm n + InP contact layer doped at 2.0xl0 18 cm -3 , 

• a 1.0 /xm n InP active region doped doped at l.OxlO 16 cm -3 , 

• a 0.2 /xm n + InP contact layer doped at 2.0xl0 18 cm -3 , and 

• a 0.1 /xm n + InGaAs cap layer doped at 2.0xl0 18 cm' 3 . 

Prior to processing the wafer, the doping profile in the active region is characterized through 
C-V measurements. To perform these measurements, the top n + InGaAs and InP layers 
are chemically etched from a small sample so that Schottky contacts can be formed. The 
processing sequence for Gunn device fabrication is shown in figure 11 and described below. 



Third International Symposium on Space Terahertz Technology Page 489 



6.1 Island Definition and Integral Heat Sink Formation 

The first step consists of defining square islands approximately 400 //m x 400 /an in size 
separated on all sides by 100 /xm wide trenches. These trenches are etched down to the 
InGaAs etch-stop layer. An n-ohmic contact (Ni/Ge/Au/Ti/Au) is evaporated over the 
whole surface and then gold is plated to a thickness of 25 /xm to form the integral heat 
sink. The top InGaAs cap layer reduces the contact resistance [10] of the ohmic contact 
because InGaAs has a lower bandgap than GaAs. The isolation provided by the trenches is 
helpful in reducing the cracking of the semiconductor epilayers during the annealing process. 
Cracks occur because the gold heat sink and the InP semiconductor have different thermal 
expansion coefficients. Figure 11(a) shows a cross section of the sample at the end of the 
gold plating. 

6.2 Substrate Thinning and Top Contact Definition 

The InGaAs layer, referred to as an etch-stop layer, permits the complete removal of the 
substrate by chemical etching. The chemical solution HCl:H20 (4:1) selectively etches InP 
and does not etch InGaAs. Once the substrate is removed, the InGaAs etch-stop layer 
is etched away using HzPO4.H2O2.H2O (1:1:8) which does not attack the InP n + region. 
Standard lift-off techniques are used to define circular diodes with sizes varying from 30 to 
65 /im in diameter which are then metallized to form n-ohmic contacts. Figure 11(b) shows 
a cross section of the sample after the substrate thinning and the heat sink formation. 

6.3 Gold Plating of Top Contacts 

In the standard IHS technique, the next step would be to etch the mesa. However problems 
in bonding have been encountered due to the thin ohmic contact. A thick ohmic contact is 
obtained by plating gold on top of the evaporated ohmic contacts. A conductive metal layer 
is needed to electroplate uniformly over all the contacts. A Ti/Au/Ti layer is evaporated 
over the whole surface, then an alignment over the ohmic contacts is used to open holes in 
a thick photoresist (3 /jm ). The photoresist is removed from a small region at the edge of 
the sample. The exposed Ti layer is also removed in Buffered HF and the sample is plated 
for a thickness of 2 fim - 3 fim. Figure 11(c) shows a cross section of the sample after the 
plating step. 

6.4 Mesa Definition and Annealing 

The final step before mounting individual chips consists of etching the mesas and annealing 
the ohmic contacts. The final structure is shown in figure 11(d). 



Page 490 



Third International Symposium on Space Terahertz Technology 



InGaAs cap layer 



) l l l|l lll l l l|l ll |l l l|l l l|lll| l ll|l ll )llll ll lj ll l|ll l | l l l |lip 



nlnP 



InGaAs etch -stop - 
layer 




InP Substrate 



Figure 10: InP Gunn structure 



7 Conclusions 

A new method has been developed for estimating the material parameters used in the 
Monte Carlo model. By comparing simulation and experimental results in the W-band, 
we obtained more accurate material parameters. Lower values for the intervalley energy 
separation and higher values for the deformation potentials than stated in the literature 
were used. A possible explanation for these trends is the high operating temperature of the 
Gunn device which perturbs the band-structure. 

Using these parameters, it was shown that it is possible to operate fundamental mode 
InP Gunn devices in the D-band. The performance of a flat doped structure can be con- 
siderably improved by employing a graded doping profile in the active region. Specifically, 
a linearly graded doping increasing from the cathode toward the anode improves the con- 
version efficiency, the output power, reduces the electric field at the anode, and results in a 
smaller current density compared with a flat profile. A structure with a doping decreasing 
toward the anode is not desirable because it increases the electric field at the anode and 
does not reduce the dead zone. As a result, the device breaks down at lower voltages and 
the performance is degraded. 

A processing technology for GaAs and InP Gunn devices has been developed based on 
the integral heat sink processing technique. An Etch-stop layer between the substrate and 
the epilayers was included in the wafer design in order to completely remove the substrate 
and obtain better uniformity across the chip. InGaAs cap layers were used to reduce the 
contact resistance. A process was developed for plating the top contacts with gold to 
facilitate bonding. 



Third International Symposium on Space Terahertz Technology 



Page 491 



Plated Au 
n-ohmic 



n + InGaAs 




IbP Substrate 



a) Island definition, n-ohmic evaporation, and 
gold plating (25 Jim) 



n-ohmic 
Ni/Ge/Au/Ti/Au 




b) Substrate thinning, etch stop layer removal, 
and n-ohmic evaporation 



Figure 11: Processing sequence for InP Gunn fabrication 



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Third International Symposium on Space Terahertz Technology 



Ti/Au/Ti 




Photoresist 



c) Gold plating of ohmic contacts 




d) Final diodes after annealing and mesa etch 



Figure 11: Cont. Processing sequence for InP Gunn fabrication 



Third International Symposium on Space Terahertz Technology p a g e 493 



References 

[1] W. Fawcett, A. D. Boardman and S. Swain, "Monte Carlo Determination of Electron 
Transport Properties in Gallium Arsenide," J. Phys. Chem. Solids, 30, 1969, pp. 643. 

[2] K. Brennan, K. Hess, J. Y. Tang, and G. J. Iafrate, "Transient Electronic Transport 
in InP Under the Condition of High-Energy Electron Injection," IEEE Trans, on 
Electron Dev., ED-30, 12, pp. 1750-1753, Dec. 1983. 

[3] D. C. Herbert, W. Fawcett, and C. Hilsum, "High Field Transport in Indium Phos- 
phide," J. Phys. C: Solid State Phys., Vol. 9, pp. 3969-3975, 1976. 

[4] G. H. Glover, "Study of Electron energy Relaxation Times in GaAs and InP," J. Appl. 
Phys., 44, No. 3, pp. 1295-1301, March 1973. 

[5] T. J. Maloney, and J. Frey, "Transient and Steady-State Electron Transport Proper- 
ties of GaAs and InP," J. Appl. Phys., 48, No. 2, pp. 781-787, Feb. 1977. 

[6] M. V. Fischetti, "Monte Carlo Simulation of Transport in Technologically Significant 
Semiconductors of the Diamond and Zinc- Blende Structures- Part I: Homogeneous 
Transport," IEEE Trans, on Electron Dev., ED-38, No. 3, pp. 634-649, March 1991. 

[7] R. A. Zettler, and A. M. Cowley, "Batch fabrication of Integral-Heat Sink IMPATT 
Diodes," Electronics Letters, Vol. 5, No. 26, pp. 693-694, Dec. 1969. 

[8] S. Y. Narayan, J. P. Paczkowski, "Integral Heat Sink Transferred Electron Oscilla- 
tors," RCA Review, Vol. 33, pp. 752-765, Dec. 1972. 

[9] A. Paolella, R. L. Ross, and J. Ondria, "Advanced mm- Wave Sources by Automated 
MBE," Microwave Journal, p. 149, April 1986. 

[10] J. M. Woodall, J. L. Freeouf, G. D. Pettit, T. Jackson, and P. Kirchner, "Ohmic Con- 
tacts to n-GaAs using Graded Band Gap Layers of Ga\_ x In x As Grown by Molecular 
Beam Epitaxy," J. Vac. Sci. TechnoL, 19, No. 3, Sept./Oct. 1981, p. 626. 



Page 494 Third International Symposium on Space Terahertz Technology 

A Recent Advances in Superconducting- Mixer 



JJ 3. -27 ? 68 

Stimulations 



S. Withington and P.R. Kennedy 

Cavendish Laboratory, 

University of Cambridge, 

England. 

March 22, 1992 



1 Introduction 

Over the last few years, considerable progress has been made in the development of tech- 
niques for fabricating high-quality superconducting circuits, and this success, together with 
major advances in the theoretical understanding of quantum detection and mixing at mil- 
limetre and submillimetre wavelengths [1], has made the development of CAD techniques 
for superconducting nonlinear circuits an important new enterprise. For example, arrays of 
quasioptical mixers are now being manufactured, where the antennas, matching networks, 
filters and superconducting tunnel junctions are all fabricated by depositing niobium and 
a variety of oxides on a single quartz substrate. There are no adjustable tuning elements 
on these integrated circuits, and therefore, one must be able to predict their electrical 
behaviour precisely. This requirement, together with a general interest in the generic be- 
haviour of devices such as direct detectors and harmonic mixers, has lead us to develop 
a range of CAD tools for simulating the large-signal, small-signal, and noise behaviour of 
superconducting tunnel junction circuits. 



2 Large-signal analysis 

To model the behaviour of a quasiparticle mixer, it is first necessary to simulate the large- 
signal steady-state dynamics of the local-oscillator circuit. Once the large-signal operating 
point is known, it is then possible to perturb, either numerically or analytically, the under- 
lying system of equations to gain information about the linear relationships between signal 
and noise variables. 

The main problem is how does one calculate the periodic current that flows through a 
tunnel junction when a periodic voltage is applied? For semiconductor devices this calcula- 
tion is almost always carried out in the time domain, and fast Fourier transforms are used 
to interface the terminal waveforms to the frequency- domain description of the embedding 
circuit. 

Classical resistive mixer diodes are relatively easy to simulate because the induced 
current is an instantaneous function of the terminal voltage. Quantum mixer diodes on 
the other hand are difficult to simulate because the tunnelling current depends on the 
voltage that was across the junction at very long times in the past. In the time domain 



Third International Symposium on Space Terahertz Technology Page 495 



the current is calculated through an integral which is similar to the convolution integral 
of linear systems theory, and the tunnel junction is characterised by a response function 
■which oscillates at the gap frequency with an envelope which decays inversely with time 
at large times. To evaluate the tunnelling current it is necessary to sample the terminal 
voltage at a rate greater than the gap frequency , and to integrate beyond a limit which is 
inversely related to the voltage width of the dc nonlinearity. Time-domain simulations are 
useful for studying the switching behaviour of tunnel junctions, but they are inappropriate 
for studying the steady-state behaviour of RF circuits. 

When a sinusoidal potential is applied to a superconducting tunnel junction, the wave- 
functions associated with the quasiparticle states on one side of the barrier are coherently 
phase modulated. The spectrum of the phase factor is a comb of delta functions whose 
coefficients are the elements of a Bessel-function sequence. The Bessel functions have the 
same argument, determined by the voltage drive level, and consecutive orders ranging from 
some large positive integer to the same negative integer. The trick is to recognize that when 
a periodic potential is applied, the spectrum of the overall phase factor is the convolution 
of the spectra associated with the individual harmonic contributions. Once the spectrum 
of the overall phase factor is known, it is possible to calculate the harmonic phasors of the 
tunnelling current from the dc I-V curve and its Hilbert transform [2]. 

The above procedure describes a way of calculating the periodic current that flows in a 
tunnel junction when a periodic potential is applied. In a real circuit the tunnel junction 
is embedded in a linear network and the problem of determining the various voltages and 
currents is complex. Applying the method of harmonic balance [3] to a generic circuit 
comprising a tunnel junction and a Thevenin voltage source, leads to a system of coupled 
nonlinear algebraic equations. Mathematically, the problem then consists of finding the 
roots of these equations; electrically, the problem is equivalent to searching for a waveform 
that simultaneously satisfies the circuit equations at every harmonic frequency. In nonlinear 
CAD terminology, the scheme is a frequency- domain spectral-balance method, however, 
unlike other versions, the spectral decomposition is based on device physics, rather than on 
expanding the terminal behaviour in a set of basis functions. 

The set of algebraic equations that results from applying the method of harmonic bal- 
ance to a tunnel-junction circuit must, in general, be solved numerically. By repeatedly 
analyzing, in different ways, a large pseudo-random set of tunnel-junction circuits, we have 
investigated the speeds and stabilities of a range of iterative root-finding techniques. A 
comparison of the techniques is shown in Fig. 1, where we have plotted the percentage 
of circuits that converge, and the mean number of iterations taken, as a function of the 
damping factor. The damping factor is a coefficient between and 1 which determines the 
degree to which the result of an iteration influences the next guess. A small value improves 
stability at the expense of reducing the rate of convergence. The solid and dashed lines in 
Fig. 1 correspond to two different quality characteristics. It should be appreciated that the 
plots represent a total of around 20,000 circuit simulations. 

The tunnel junction is a nonlinear admittance in the sense that it is most easy to 
calculate the current in terms of the terminal voltage. However, fixed-point voltage-update 
methods [4] are inappropriate for analyzing tunnel- junction circuits, especially in a common- 
user environment, because they fail to converge when the large- signal harmonic admittances 
of the tunnel junction are much greater than those of the embedding circuit. This problem 
is clearly demonstrated in Fig. 1 where it is seen that the routine will only converge if 
the system is heavily damped. A slightly more sophisticated way of finding the roots is 



Page 496 



Third International Symposium on Space Terahertz Technology 



-a 
v 
ho 

u 

V 

> 
S 
o 
u 



s 
u 





Fixed Point 


80 


HI 1 1 1 1 1 III 1 ILL 


60 


— — 


40 


^r x — 


20 


r\ -: 







1 1 1 1 1 1 1 rm 1 1 1 1 







Secant 


80 


III 


|... 


|i.i|iLL 


60 






— 


40 






— 


20 




— 








— 





771 


Llll 


inlirr 



.2 .4 .6 .8 



.2 .4 .6 .8 




Harmonic Newton 

iiijmjmjjii 



-rr 1 1 1 1 1 1 1 1 1 1 1 i-r 



.2 .4 .6 .8 



L_l 1 I I I I f 1 I I I I 1 I 1_| 




.2 .4 .6 .8 



Jlllllll II II IIL| 

200 E- 
150 
100 
50 

OhIihIiiiIuj: 



Jll I I II I I III ML 



.2 .4 .6 .8 

Damping factor y 




.2 .4 .6 .8 



Figure 1: Comparison of various techniques for calculating the large-signal quantum be- 
haviour of superconducting tunnel-junction circuits. 



to use a multi-dimensional variant of the secant method [5]. This method is similar to 
the fixed-point method, in the sense that it is only necessary to calculate the tunnelling 
current once per iteration, however, because coarse derivative information is included one 
might expect the routine to behave more reasonably. Somewhat surprisingly, the routine is 
significantly worse despite the additional information. The problem is caused by the fact 
that, effectively, only the terms on the leading diagonal of the Jacobian matrix are included, 
and coupling between harmonics relies on the current calculations. As long as the current 
at a given harmonic is most strongly influenced by the voltage at the same harmonic then 
the routine will work well. In a highly nonlinear tunnel-junction circuit, however, there 
is strong coupling between harmonics and the routine is inadequate. Fig. 1 shows the 
behaviour of a harmonic-Newton [6] [7] scheme where the full Jacobian matrix is used. It is 
possible to calculate the Jacobian matrix analytically, however, we prefer to calculate the 
Jacobian matrix using finite differences. Harmonic Newton results in the least number of 
failures; in fact, it finds a solution for 75 % of the circuits studied, and to a large extent, 
the stability of the method is independent of the quality of the junction being investigated. 
The fact that the convergence parameter has little effect on this fraction, together with the 
almost reciprocal dependence of the mean number of iterations, shows that if it is possible 
for the method to find a solution then it will eventually do so. Reducing the damping factor 



Third International Symposium on Space Terahertz Technology Page 497 



simply reduces the size of the voltage steps taken at each iteration, however, these steps 
are usually in the correct direction. 

We have now performed a very large number of real circuit simulations, and despite 
the fact that 25 % of the randomly generated circuits failed to converge, we have never 
come across a real circuit that has not converged. We have investigated this problem in 
some detail, and we have found that many of the circuits that do not converge are behaving 
in a non-periodic manner. This behaviour usually requires that the embedding circuit 
impedances are very much larger than the normal-state resistance. 

An alternative approach to finding a root in many dimensions is to recast the problem 
into a multidimensional optimization. To do this change, the error function is used to 
construct a scalar quantity that has a global minimum at the required root. One of the 
attractions of optimization is that uninteresting variables, such as the local oscillator drive 
level, can be eliminated from the analysis by making the variable part of the objective 
function. In general, we have found that optimization methods are slow and should not be 
used unless there is a particular reason to do so. Unfortunately, there is insufficient time 
to discuss this more advanced topic in this short paper. 

The results of a typical large-signal analysis are shown in Fig. 2. The sequence of plots 
shows how the pumped dc I-V curve of a typical Nb-AlOx-Nb tunnel junction evolves as 
the wCR product is changed. As the capacitance decreases the subgap current increases, 
and non-classical negative differential resistance is induced on a number of photon steps. 
Notice that large capacitances are required before the characteristic relaxes to its constant 
sinusoidal- voltage-driven form. Also shown, for comparison, is an analysis where the har- 
monic feedback is turned off. Curiously, it seems as if internal harmonic pumping can 
enhance the negative differential resistance induced on high-order photon steps — for this 
reason it is possible, in certain circumstances, for the small-signal behaviour of a mixer to 
be very sensitive to harmonic impedance levels. 



3 Small-signal analysis 



Once the large-signal behaviour of a mixer has been established, it is possible to calcu- 
late the small-signal and noise performance. The admittance and noise-current correlation 
matrices are determined, in the usual way, through quantum-mechanical generalizations of 
commonly used classical concepts. We then use a selection of linear transforms to reduce 
the admittance parameters to two-port impedance and scattering parameters, and the cur- 
rent correlation matrix to a noise-temperature matrix from which the standard two-port 
noise parameters can be deduced [8] [9]. It transpires that the whole scheme, both signal 
and noise, can be very elegantly normalized to the gap voltage and gap current of the 
tunnel barrier. The advantages of our generalized approach are that one does not have to 
specify before hand which ports are to be used for the input and output, and one can easily 
calculate the two-port small-signal and noise parameters which can then be loaded into pro- 
prietary microwave circuit simulators for further analysis. For example, we are interested in 
designing mixers that have the first stage of low-noise IF amplification in the mixer block. 
A further advantage of our scheme is that the noise performance is described in terms of 
correlated travelling noise waves, and this approach is an elegant way of considering a mixer 
as an integral part of a quasioptical system along which noise waves propagate. 



Page 498 



Third International Symposium on Space Terahertz Technology 



IS 



No harmonics. 
Rs = 6.0 , u>CR = 0.0 






2 


J T r 1 1 | 1 


1 1 1 I 1 1 1 1 1 1 


1 1 1 1 






! Rs = 2.0 


uCR =1.0 


/ - 




l.S 


- 




- 





1 






- 







- 




- 




IiimIi 


, , 1 , , , , 1 , , 


, 1.! 



J I I I I I I I I ' I I I I 1 I 

Rs = 2.0 , u/CR = 0.0 




Figure 2: The dc I-V curve of a pumped Nb-AlOx-Nb tunnel junction for different values 
of ujCR product. The first plot does not include internal harmonic pumping. 

4 Mixer simulations 



To date we have studied the large-signal, small-signal, and noise behaviour of mixers by 
adopting the design procedure suggested by Kerr [10]. That is to say, the mixer is operated 
in a double-sideband mode, and the source and load impedances are assumed to be real. The 
ratio of the source and load impedances, which in practice is determined by the geometry 
of the mount, is set at some fixed value. In general, we use a value of unity as a higher 
value tends to degrade the input return loss of the mixer. It is interesting to note, however, 
that it may be possible to choose the ratio so as to minimize the sensitivity of the gain 
to variations in the source resistance. The free parameter, as far as the design process 
is concerned, is the normal-state resistance. Although we assume that the capacitance of 
the tunnel junction is tuned out at the fundamental, we assume that the impedance at 
the harmonics is given by the capacitance of the tunnel junction alone. The effects of 
junction capacitance are considered in companion paper [11], here we simply demonstrate 
the procedure by plotting, in Fig. 3, the transducer gain, noise temperature, input return 
loss, and normalized output impedance of a typical Nb-AlOx-Nb mixer as a function of the 
normalized source resistance; the various curves are for different normalized frequencies ( 
normalized to the gap frequency). It is interesting to note that the overall performance is 



Third International Symposium on Space Terahertz Tech7iology 



Page 499 



s 

•3 

bo 

CQ 
CO 
CO 




-10 



V 



V 

a. 
S 

V 



o 

a 

CO 
co 

CO 




1.5 2 



_ 20 



05 

w 
O 



J 10 - 



s 
a 

0) 



3 
O. 

a 



- 



-10 



_i r t - i [ 1 1 1 i | 1 1 


1 r | f i i r 


" °- 2 _ .101 


__ 


^^0.8 N. 


^s. ^_ 


~i 1 1 1 1 1 1 1 1 1 1 1 


i\ 1 1 1 1 1" 



1.5 




Normalized source resistance 



Figure 3: The transducer gain, noise temperature, input return loss, and normalized output 
impedance of a typical Nb-AlOx-Nb mixer as a function of the normalized source resistance. 



very poor for low values of source resistance, and this is probably the single most important 
reason why mixer performances improve significantly when integrated tuning elements are 
used. 

A useful normalized expression can easily be derived for the source resistance at which 
unity gain, good input match, and minimum noise temperature can be achieved. The 
expression is 



|^ = 0.5Vp° 92 = 78 



f(GHz) 



V„ (mV) 



-0.92 



(1) 



and it applies for frequencies between 0.2 and 0.8 of the gap frequency. The exponent is 
slightly different from that given by Kerr and Pan, because we have taken into account the 
fact, that the optimum bias point does not remain in the middle of the first photon step 
below the gap for frequencies greater than about 0.5 of the gap frequency. 

It is well known that if one plots the conversion gain, at a given frequency, as a function 
of the uiC R product, at some point the conversion gain becomes depressed. This can be 
regarded as the frequency at which harmonic effects become significant; or equivalently, the 
frequency at which the five-port, rather than the three-port model should be used. Using 
the above value for the source resistance, we have investigated this behaviour and generated 



Page 500 



Third International Symposium on Space Terahertz Technology 



wCRn = 0.5 



10000 



a 
■a 

t 
w 



O 



1000 



















































































* \ 










^^^ 






\ 
\ 










_^^~ 




S r — - 




_^00*~ 




^•"^ 




r»* 


^t*~~ 


















^***~ 






^0* *' 












. 


+*** 
















^^ 


^00**^ • 
















^-"^. 


,'' ^^--^ 


^^-^ 














^^^- *■ 


















-"' ' ^s^*Z^" 


















r ^O-* **Z*>* <^ *^ ^""^ 






















































i^"*" ^^ 00r ' ^*** 0r 


















r*^^**^ 


















*" 



















1.0 



2.0 



9.0 



100 



Frequency / (GHz) 



1000 



Figure 4: Critical current density against frequency for different values of u/CR product. 
The dotted line shows the optimum current density as a function of frequency. 



the following expression for the optimum wCR product 



uCR 



\opt 



= Vf° 75 = 61 



V g (mV. 



If (GHz)} 



0.75 



(2) 



and this in turn generates the following expression for the optimum value of the critical 
current density: J c (Acm- 2 ) - OAf(GHz) 175 . We have assumed I c R n = 1.8ml' and a 
specific capacitance of 45fFfim~ 2 . Once again the above requirement is less severe than 
that published by Kerr. Finally, in Fig. 4 we plot the optimum critical current density as 
a function of frequency, and we show lines of constant u)C R product. Above the dotted 
line, harmonic effects are important, whereas below the dotted line harmonic effects can be 
ignored. 



References 



[1] J.R. Tucker and M.J. Feldman, "Quantum detection at millimetre wavelengths," Rev. 
Mod. Phys., vol. 57, pp. 1055-1113, 1985. 



Third International Symposium on Space Terahertz Technology Page 501 



[2] S. Withington and E.L. Kollberg, "Spectral-domain analysis of harmonic effects in 
superconducting quasi-particle mixers,"IEEE Trans. Microwave Theory Tech., vol. 
MTT-37, pp. 231-238, 1989. 

[3] K.S. Kundert and A. Sangiovanni- Vincent elli, "Simulation of nonlinear circuits in the 
frequency domain.'TEEE Trans. Computer-Aided Design, vol. CAD-5, pp. 521-535, 
Oct. 1986. 

(4] R.G. Hicks and P.Kahn, "Numerical analysis of nonlinear solid-state device excitation 
in microwave circuits," IEEE Trans. Microwave Theory Tech., vol. MTT-30, pp. 251- 
259, Mar. 1982. 

[5] C. Camacho-Penalosa, "Numerical steady-state analysis of nonlinear microwave cir- 
cuits with periodic excitation,"IEEE Trans. Microwave Theory Tech., vol. MTT-31, 
pp. 724-730, Sep. 1983. 

(6] C.-Y.E. Tong and R. Blundell, "Simulation of superconducting quasiparticle mixer us- 
ing a five-port model," IEEE Trans. Microwave Theory Tech., vol. MTT-38, pp. 1391- 
1398, Oct. 1990. 

[7] S. Withington and P. Kennedy, "Numerical procedure for simulating the large-signal 
quantum behaviour of superconducting tunnel-junction circuits," Proc. IEE, part G, 
vol. 138, pp. 70-76, Feb. 1991. 

[8] S. Withington, "Scattered noise waves in microwave and millimetre- wave net- 
works, "Microwave Journal, vol. 32, pp. 169-178, 1989. 

[9] L.R. D'Addario, "Noise parameters of SIS mixers," IEEE Trans. Microwave Theory 
Tech., vol. MTT-36, pp. 1196-1206, July 1988. 

[10] A.R. Kerr and S.-K. Pan, "Some recent developments in the design of SIS mixers," 
Infrared and Millimetre Waves, vol. 11, pp. 1169-1187, 1990. 

[11] A.R. Kerr, S.-K. Pan, and S. Withington, "Embedding impedance approximations in 
the analysis of SIS mixers," Proceedings Space Terahertz Conference, 1992. 






Page 502 Third International Symposium on Space Terahertz Technology 

SUBMILLIMETER WAVE DETECTION WITH 
SUPERCONDUCTING TUNNEL DIODES 



Michael J. Wengler M Q 3 - %7 '7 6 9 

University of Rochester 



ABSTRACT 
Superconductor-Insulator-Superconductor (SIS) diodes are the detector elements in the 
most sensitive heterodyne receivers available from 100 to 500 GHz. SIS mixers are the front end 
of radio astronomical systems around the world. SIS mixer technology is being extended to 1 THz 
and higher frequencies for eventual use on spacebome astronomical experiments. 

Here is a short review of submillimeter SIS mixers. The role of impedance matching in the 
proper design of an SIS mixer is described. A variety of methods for achieving good impedance 
match at submillimeter frequencies are presented. The experimental state of the submillimeter SIS 
jnixer art is described and summarized. 



1 . INTRODUCTION 

Twelve years ago, the first descriptions of mixing on a superconducting tunnel diode called 
a Superconductor-Insulator-Superconductor (SIS) were published [1,2]. At about the same time, 
a comprehensive theoretical investigation of the SIS made clear that SIS's respond to photons at 
millimeter and submillimeter wavelengths [3]. The SIS has become the instrument of choice for 
millimeter spectroscopic radio astronomy, finding use on radio telescopes around the world [4-13]. 
Since the discovery of the SIS mixer, the technology for fabricating SIS circuits has improved 
immensely so that nearly ideal niobium devices with sub-micron feature sizes can be fabricated 
[14-20]. Recently, the high performance of SIS mixers has been extended to submillimeter 
wavelength radio astronomy [8, 21-23, 20, 24]. 

This paper is a shortened version of a review published elsewhere [25]. The full review 
includes an in-depth introduction to the field of SIS mixing. An excellent and comprehensive 
earlier review of both theory and experiment is that of Tucker and Feldman [26]. Reviews of the 
state of the art of SIS mixers have appeared regularly [27-30]. Other reviews include both SIS 
mixers and competing receiver technologies used in radio astronomy [31, 32]. 



Third International Symposium on Space Terahertz Technology Page 503 

2 . RF ADMITTANCE OF THE SIS 

The SIS mixer has two important if input admittances. The first one is Ylo» the admittance 
the SIS presents to the LO. The second is Yin, the admittance the SIS mixer presents at the signal 
frequency. Because the signal power must always be much less than the absorbed LO power for a 
mixer to avoid saturation, it is useful to think of Ylo as the "large signal admittance" of the SIS, 
and Yin as the "small signal admittance." 

For a submillimeter SIS mixer, Ylo is generally quite close to Gn, the dc normal state 
admittance of the SIS. Ylo is nearly constant over the range of useful dc and LO biasing 
conditions. Ylo is completely independent of the signal, image, LO, and harmonic source 
admittances to which the SIS is coupled. In fact, Ylo is the admittance presented by the SIS seen 
as a passive absorber of radiation. 

Yin, on the other hand, is a highly variable quantity. It does change with all of the 
parameters mentioned above. It is possible for Yin to have a negative real part, while Ylo mus t 
always have a positive real part. For this reason, Yin m ust be interpreted as the input admittance 
of the SIS viewed as an active device powered by the LO. 
2.1. SIS Parasitic Capacitance 

In submillimeter SIS mixers, a dominant part of the SIS's rf admittance is due to its 
parasitic capacitance. The capacitance per unit area of the SIS junctions used in submillimeter 
mixers is nearly a constant depending only on the materials from which the junctions are made. 
Numbers commonly used in design are 50 fF/(jim) 2 for lead alloy SIS's, and 50 to 70 fF/(|i.m) 2 
for niobium SIS's with aluminum-oxide insulating layers. 

Many mixer designers aim to minimize the SIS capacitance. In fact, many of the best SIS 
mixers have been built with low capacitance junctions. The lowest SIS receiver noise temperature 
at 230 GHz are achieved with 0.25 (|im) 2 .Nb junctions with C = 1 .7 x G^/lnf [13]. Open- 
structure SIS mixers have work fairly well at their low frequency ends where C < 1.0 x GN/27rf 
[8, 33]. 

However, there is some evidence that C = 3 x G^/2Kf provides more benefit than harm in a 
well designed SIS mixer. The reasons for this, and the value of the optimum capacitance are 



Page 504 Third International Symposium on Space Terahertz Technology 

discussed in these references [1 1, 34]. The larger capacitance seems especially important when 
series arrays of SIS's are used [35]. 

Whether or not a non-zero capacitance is helpful becomes a less important question as the 
operation frequency of SIS mixers is raised. All suggestions of a non-zero capacitance still have 
that optimum falling as f" 1 or f" 2 . As f is raised it becomes increasingly difficult to fabricate SIS's 
with C less than any of the proposed optimum values. 
2 .2 . Optimum Signal Source Admittance: Match to Yio 

The signal appears to come from a source admittance Ys. The value of Ys is one of the 

major design available to an SIS mixer designer. For submillimeter SIS mixers, the proper choice 
is Ys = Y* . This choice of Ys minimizes the SIS mixer noise temperature. This result is 

predicted from photodiode mixer theory [25], and also by detailed Tucker theory calculations [36]. 

It is paradoxical that it is not the actual small signal admittance Yin to which the mixer 

structure must supply a match, but the large signal LO admittance, Ylo In fact, a choice of 

Ys = YjL will maximize the SIS mixer's gain instead of its noise. Because IF amplifiers have very 

low noise levels, it is much more important to minimize mixer noise than to maximize mixer gain in 
submillimeter SIS mixers. 

3 . IMPEDANCE MATCHING TECHNIQUES 
There are two tasks which must be accomplished to match the signal source admittance Ys 
to the SIS's input admittance Ylo- Fu " st > the values of Ylo typical of SIS's are much larger than 
the source admittances presented by most antennas and waveguide structures. Second, the 
parasitic capacitance of the SIS must be tuned out. These two tasks can be accomplished in a few 
different ways. In some cases, a single tuning structure can do both of these things 
simultaneously. 
3.1. Integrated tuning 

It is a relatively simple matter to integrate tuning structures with the SIS diode in its 
photolithographic stage of fabrication. There are at least two layers of low loss superconductor 
required in SIS fabrication, which allows for various "two-wire" tuning elements. With the 
addition of a thick (around 2,000 A) insulating layer, stripline tuning structures can be fabricated. 



Third International Symposium on Space Terahertz Technology 



Page 505 



I 



T i' 




a ) 



b ) 



Figure 1 [38]. Two tuning structures integrated with SIS junctions, a) The sickle- 
shaped piece is the bottom layer of superconductor, the straight piece the top. 
Where the two layers overlap, they are actually separated by a thick (~2,000 A) 
layer of insulator, except in the dashed square, where they come much closer to 
form the SIS. b) A tuning structure based entirely on microstrip. 

Direct measurements of the efficacy of tuning elements in the 100-500 GHz range have been made 
using Fourier Transform Spectrometer (FTS) measurements [37]. 

The two tuning structures shown in fig. 1 are compact and simple designs proposed by 
Kerr, Pan, and Feldman [38]. Their primary role is to provide an inductance at the SIS which 
tunes out its parasitic capacitance. The structure shown in b) is particularly appropriate for higher -- 
rf frequencies. In both a) and b) structures are shown which are A/4 in length. Here, A refers to 
the wavelength associated with rf radiation propagating along the integrated tuning structure. The 
structures labeled with length A/4 are designed to present a short circuit to the rf at their left ends, 
but are open circuited at dc and fjp. In a), the tuning structure shown to be / long is essentially a 
single-turn inductor. In b), the tuning structure shown to be / long is a short length of high 
impedance microstrip. Since it is rf-shorted at one end, it presents an inductive susceptance across 
the SIS. The reactances associated with these structures are very small at the IF and so have no 
effect on the IF or dc properties of the mixer. 

Using SIS circuits designed with integrated tuning, it is possible to build SIS mixers with 
no mechanical tuning which have excellent responsivity over an entire waveguide band at 
millimeter wavelengths [11, 39]. These two mixers are quite different in their design, both should 
be reviewed for a good appreciation of the range of integrated tuning circuitry which is possible. 



Page 506 



Third International Symposium on Space Terahertz Technology 





Figure 2 [40]. A spiral antenna SIS mixer with transmission line tuning a) The 
radiauon coupling structure is a 3.4 mm diameter spiral antenna, b) Instead of 

SfXJ S at *? e °T^ lead i ° f the spiral (1) ' a mi <™stripline (3) is formed 
and the SIS (2) is placed at its end. 

3 .2 . Microstrip transmission line transformers 

The inductive structures shown above address only the problem of tuning out SIS parasitic 
capacitance. Using microstrip transmission lines, it is possible to tune out the SIS's capacitance 
and to lower Ylo to a more convenient value. 

The simplest circuit which does this is shown in fig. 2. The large-scale structure shown in 
a) is a spiral dipole antenna. Radiation coupled by this antenna to the antenna leads at its center has 
a source admittance of about (1 14 Q)-l. Rather than placing the SIS at the center of the antenna, 
the SIS is placed at the end of a microstrip transmission line as shown in b). One of the poles of 
the antenna is used as the ground plane for this transmission line. The transmission line has a 
fairly low impedance and a length chosen so that the SIS's high value of Y L0 is transformed down 
to about (1 14 Q)-l where it is connected to the center of the antenna. In fact, the transmission line 
transformer is a little longer than a standard quarter-wave transformer so that it also tunes out the 
SIS's parasitic capacitance. 



Third International Symposium on Space Terahertz Technology Page 507 

This simple technique can be extended in useful ways. The junction need not be placed at 
the end of the transmission line structure [41]. A symmetric structure with junctions and 
transmission lines on both antenna leads adds flexibility to design and will avoid mismatch due to 
the transition from a balanced mode on the antenna to an unbalanced mode on the microstrip [41]. 
A multi-step (Chebyshev) transformer can be used for excellent broadband performance [42]. The 
transformation to low impedance junctions improves the dynamic range of the mixer [41]. A 
tapered transmission line transformer can provide a broadband match [23]. 

4. SIS MIXER RESULTS 
SIS mixers of some variety have been built. Niobium technology is the best current choice 
for submillimeter SIS's. Both waveguide and quasioptical coupling structures are useful at these 
high frequencies. Integrated tuning structures of increasing complexity are being used to improve 
mixer performance. 

4.1. SIS Junction Fabrication 

Many of the best SIS mixers use lead alloy SIS junctions. However, niobium and niobium 
alloy junctions have such great advantages that even the fabricators of lead alloy SIS's are 
developing niobium technology. Niobium junctions have been fabricated and used in high 
performance SIS mixers [15, 18, 19, 43, 13, 23]. Niobium-nitride alloy SIS's [17, 44] are 
interesting as they will operate at higher temperatures, and possibly higher frequencies than pure . 
niobium SIS's. 

4 .2 . Radiation coupling structures 

A sensitive SIS mixer requires an SIS junction which is well coupled to its input radiation. 
The coupling structure chosen effects this in two important ways. 1) The coupling structure 
determines Ys, the source admittance from which the radiation appears to come. 2) The coupling 
structure determines the beam pattern of the SIS mixer. A good beam pattern is essential for high 
efficiency coupling between the mixer and the radiation it is intended to detect. Table 1 shows the 
reported results of a variety of SIS mixers and receivers along with the coupling structure that each 
one uses. 



Page 508 



Third International Symposium on Space Terahertz Technology 



Table 1. Summary of some of the best reported SIS receiver results. The bold 
lines separate the results into frequency ranges. Within a frequency range, the 
results are listed in order of increasing Trecdsb- 



Rf 
(GHz) 


Trec 
DSB 

(K) 


Tmtx 
DSB 

(K) 


Gain 
DSB 
(dB) 


SIS material, size, 
configuration 


Coupling 
(waveguide or 
antenna shape) 


Reference 


230 


48 


38 


-2.6 


0.25 sq. micron Nb 


waveguide 


[13] 


230 


48 






Nb 


waveguide 


[11] 


230 


80 


60 


-2 


PbBi alloy 


waveguide 


[7] 


240 


100 






Nb 


waveguide 


[47] 


228 


114 






PblnAu submicron 


waveguide 


[6] 


230 


116 






PblnAu submicron 


spiral 


[8] 


241 


153 


85 


-7.5 


PblnAu submicron 


waveguide 


[5] 


228 


163 






PblnAu submicron 


waveguide 


[6] 


230 


200 




-8 


PblnAu submicron 


waveguide 


[10] 


220 


250 


25 


-9 


SIN! (NOT an SIS) 


waveguide 


[48] 


345 


150 






PblnAu submicron 


waveguide 


[21] 


345 


200 






Nb 


spiral 


[42] 


345 


215 






PblnAu submicron 


spiral 


[8] 


342 


214 






3 PbBi in series 


quasiopuc 


[24] 


312 


275 




-9.5 


Nb/PblnAu 


waveguide 


[22] 


492 


171 


114 


-9 


0.25 sq. micron Nb 


waveguide 


[20] 


426 


220 






Nb 


spiral 


[42] 


490 


420 


240 


-10 


Nb 


twin slot 


[23] 


525 


470 






PblnAu submicron 


spiral 


[8] 


492 


500 






Nb 


spiral 


[42] 


761 


1100 






PblnAu submicron 


spiral 


[8] 



4.3. Waveguide SIS mixers 

Until 1985, all SIS mixers used waveguide coupling structures. The rf input side of the 
mixer is almost always coupled to a scalar feedhorn [45, 46] which provides a very high quality 
Gaussian beam pattern. The waveguide behind the mixer usually contains a movable back short 
Which can help tune out the parasitic reactances of the SIS and its mounting structure. In some 
designs, a second waveguide tuner can be included to increase the range of tuning which is 
possible. In other designs, no mechanical tuning is necessary. 

A full height rectangular waveguide has dimensions a little smaller than 1/2 x 1/4 Xo where 
Ao is the free-space wavelength at frf. At 500 GHz, this waveguide is quite small, 300 x 150 um. 
It is hard to say what the upper frequency limit is for a waveguide design. SIS's in waveguides 



Third International Symposium on Space Terahertz Technology 



Page 509 



EPUNETtWAVEOUOE 
TTUNgFORMER SECTKM 



FOUTPUT 



fzzzzz&L 



^TTTzm 



CORRUQATED- 
FEEOHORNSECTON 



MICROMETER DRIVE 
(80TPI)- 




TO SECOND MICROMETER ORIVE 



Figure 3 [6]. The two-tuner waveguide design which has been implemented for 
230, 345, and 492 GHz band mixers. 

are the most sensitive available mixer at 500 GHz Walker, 1991 #134]. It seems clear that 
waveguide designs will be attempted to at least 800 GHz. 

The good features of waveguide designs are summarized. They provide excellent control 
over the rf source impedance seen by the SIS, either through fixed or mechanically tuned "" 
structures. They have excellent beam patterns capable of high efficiency coupling to radio 
telescopes and other signal sources. 

4.3. 1 . Ellison's Waveguide Mixer 

Some of the lowest noise SIS receivers for the 200 to 500 GHz frequency range [13, 21, 
20] have been built at the California Institute of Technology using a mixer designed by Ellison [6]. 
This design, shown in fig. 3, includes two movable waveguide shorts, one behind the junction 
(back-short), and one in a waveguide tee in the E-plane wall of the waveguide near the junction 03- 
plane tuner). The SIS is mounted across the E-plane of the waveguide. One side of the SIS is 
shorted directly to the mixer block. The other side is coupled through an rf-choke structure to the 
IF and dc-bias circuits. In later designs, a symmetric mount is designed so that the ground 
connection is made at the end of an rf-choke. 

This design has been verified and improved through the use of microwave scale modeling 
[49]. These models showed that the rf impedance seen by the SIS can be tuned over more than 



Page 510 Third International Symposium on Space Terahertz Technology 

half the area of a Smith chart at almost all frequencies in the waveguide band. This modeling also 
demonstrated the critical importance of the placement of the SIS in the waveguide. The SIS must 
be centered in the waveguide to avoid extra reactance due to coupling to a higher order waveguide 
mode. 

In all of the mixers based on Ellison's design, a great amount of attention has been paid to 
the IF matching circuit This circuit is designed for two criteria: 1) transform the 50 Q IF amplifier 
to a higher impedance, 100 to 200 Q, at the SIS and 2) provide a low impedance at the SIS at 
frequencies outside the desired IF band. The first criterion improves the coupled gain of the SIS 
mixer since its IF output impedance is very high. The second criterion protects the SIS from IF 
saturation [50, 51]. 

4.3.2. Round Waveguide Mixer 

An early attempt at designs that would be appropriate for high frequency scaling is 
Woody' s round waveguide mixer [4]. Mixers based on this design have been built for 115 GHz 
[4], 230 GHz [5], and 345 GHz [22]. This design is relatively easy to scale to high frequencies. 
The round waveguide is fabricated by drilling the proper sized hole in a block. The rf choke is 
built into the mixer block. The SIS junction has no rf choking or any other frequency dependent 
structures on it, so the chip can be made very small. While the sensitivity of the round waveguide 
mixers is good, the greater complexity of the other designs discussed here is clearly justified by 
their lower noise temperatures. 

4.3.3. IRAM Mixer 

SIS mixers are built at the Institut de Radio Astronomie Millimetrique (IRAM) for use on 
European radio telescopes. The IRAM mixer block uses a reduced height rectangular waveguide 
with the SIS junction mounted across its E-plane. The reduced height waveguide provides a lower 
rf input impedance than full height waveguide, and eliminates resonant coupling to higher order 
modes which trouble Ellison's design. There is a single mechanical tuner, a backshort in the 
waveguide behind the SIS. The rf choke is fabricated integrally with the SIS and is symmetric 
around the junction: it is the basis for the rf choke structure now used in Ellison designed mixers. 
IRAM mixers have been built for 100 GHz [19] and 230 GHz [7, 47]. An SIS mixer is now being 



Third International Symposium on Space Terahertz Technology 



Page 511 




Figure 4 [1 1]. The NRAO-401 170-260 GHz mixer. The center channel contains 
microstrip circuitry including the SIS junctions with inductive tuning. The channel 
on the left is a waveguide carrying the signal and LO into the mixer. The top and 
bottom channels on the right are waveguides containing movable shorts for tuning. 
Stripline to waveguide coupling structures are shown in solid black. 

developed for 345 GHz [47]. The simplicity and success of the IRAM design make it a good 
candidate for higher frequency operation. 
4.4. Waveguide-Substrate Mixers 

A number of mixer designs use a mechanically simple waveguide design to couple input 
radiation to a microstrip or coplanar transmission line built on a dielectric substrate. Critical if 
circuit components in these mixers are fabricated photolithographically. It is easier to fabricate 
small components this way than to machine them into a waveguide block. Because of this, the 
approach is very promising for higher frequency mixers. These designs have been fabricated up to 
230 GHz with excellent results. They should perform equally well at higher frequencies in the 
future. A few of these designs are presented below. 
4.4.1. NRAO-401 

The National Radio Astronomy Observatory makes a wide variety of receivers for radio 
astronomy in the US. Their NRAO-401 SIS mixer is shown in fig. 4 [11]. The rf is coupled from 
an input waveguide into stripline circuit through a broadband transition very similar to commercial 
waveguide to coaxial transitions available at much lower frequencies. Two adjustable waveguide 
tuning elements are also coupled to the circuit through similar probes to provide series and parallel 
reactances to the SIS. This mixer's sensitivity in a 230 GHz DSB receiver is the best currendy 



Page 512 Third International Symposium on Space Terahertz Technology 

available. A similar mixer is part of a 40 K DSB receiver at 100 GHz [9]. Both of these mixers 
can be tuned for SSB operation with 20 dB image rejection. 

4.4.2. NRAO Tunerless Mixer 

Another NRAO SIS mixer achieves 40-80 K DSB receiver temperatures over the entire 
WR-10 waveguide band, 75-1 10 GHz, without any mechanical tuning [11]. As in the NRAO- 
401, the waveguide to stripline transition couples radiation from the input waveguide to the 
substrate circuit. The stripline radiation makes a second transition to coplanar waveguide before it 
reaches the SIS and its integrated tuners. 

4.4.3. Yale Waveguide-Microstrip Mixer 

In this design, the input waveguide is coupled to a microstrip circuit with a 4-section 
Chebyshev single ridge transformer [39]. Nearly full WR-10 band coverage is achieved with no 
mechanical tuning. Circuits with microstrip inductive compensation of the SIS capacitance were 
investigated, but the lowest noise temperatures for this mixer were achieved without inductive 
tuning. Because of its simple mechanical design, this mixer is a good candidate for scaling to 
higher frequencies. 
4.5. Quasioptic SIS mixers 

An interesting alternative to waveguide coupling is shown in fig. 5. The SIS is fabricated 
integrally with a planar antenna. The SIS- antenna is placed on a hyperhemispherical lens. The 
lens focuses the input radiation into the center of the SIS-antenna which earns this scheme the 
name "quasioptic." A comprehensive introduction to the properties of antennas on dielectrics is 
given by Rutledge et al. [52]. The superconducting films from which the SIS-antenna is made are 
very good conductors, so the fabricated antennas will have low resistive losses despite the 
submillimeter frequencies involved. This low-loss property may not hold for fjf > 2A/h, about 
700 GHz for niobium technology. 

The planar antenna designs have the following positive features. Many of these mixers can 
be used over a few octaves of frf. The spiral and bowtie planar antennas are frequency independent 
both in their beam partem and in their antenna impedance. The SIS-antennas are fabricated 
photolithographically so no waveguide or feedhorn must be machined. Large substrates can be 
used. If an SIS is to be placed across a waveguide, the SIS and its substrate must be much smaller 



Third International Symposium on Space Terahertz Technology 



Page 513 



cm 




H 



SIS in 
Bowtie 



Quartz 
Hyperhemisphere 



Teflon Lens 




Quartz 
Substrate 



Figure 5 [33]. The bowtie dipole antenna is formed as an extension of the leads of 
the SIS diode. The radiation from this antenna is focussed into a fairly narrow 
beam by a hyperhemispherical lens placed up against the substrate. Further 
focussing is provided by a plastic lens 

than an rf wavelength. An SIS-antenna, however, should actually be fabricated on a substrate 
which is bigger than a wavelength. 

4.5. 1 . Spiral and B owtie Antennas 

The bowtie mixer in fig. 5 was improved on by using a spiral-shaped dipole antenna [8] 
like the one shown in fig. 2a. The mixer with spiral antenna had noise temperatures about half 
those of the bowtie mixer over the same large frequency range. This mixer has been used at the 
Caltech Submillimeter Observatory for radio astronomy at 1 15, 230, 345 and 492 GHz. Its beam 
couples to the telescope about as well as the beam from a waveguide receiver, resolving a difficulty 
with the bowtie design. 

The spiral antenna has a much cleaner beam pattern than the bowtie, with sidelobe and 
pedestal structure at the -20 dB level. The cleaner beam is primarily due to the better radiation 
pattern of the spiral, but it is also helped by the use of a mirror less than 1 mm behind the SIS- 
antenna. The lower noise of the spiral SIS is attributable to its better beam pattern, as is its 
excellent coupling efficiency to the CSO radio telescope. 

The submillimeter spiral SIS mixer has very recently been improved remarkably at the 
expense of its multi-octave bandwidth [42]. The design is similar to that shown in fig. 2b, but the 



Page 514 



Third International Symposium on Space Terahertz Technology 

SIS 
junction 

I 



A/4 
radial 
stubs 



^^"" l ' ' ' ' "i^TTTT 
A'^^ li muni ' - '™ 




4000A SiO 

Insulating 

Film 



Tapered 

Microstrip 

Line 



RF 
choke 



Ground 
Plane 



Slot 
Antennas 

Figure 6. The planar antenna-SIS mixer of Zmuidzinas and LeDuc [23]. The entire 
structure is built on a niobium ground plane which is etched away to create the two 
slot antennas. 

transformer is a more complicated two section Chebyshev design. The result is 75% or better 
coupling between the SIS and the spiral antenna in a one-octave rf band centered at 350 GHz. 
Receiver results using this mixer are included in Table 1. The fall-off in performance at 492 GHz 
is due to the low center frequency of this particular mixer. It should be a simple matter to adjust 
the design so that Trecdsb ^ 250 K is achieved at 492 GHz. 
4.5.2. Twin-slot antenna 

Fig. 6 shows a different planar antenna design used in the mixer developed by Zmuidzinas 
and LeDuc [23]. This receiver is reported in further detail elsewhere in this conference publication. 
This design couples a single SIS junction to two planar slot dipole antennas as shown in fig. 6. 
Unlike the bowtie and spiral, the twin-slot is a resonant antenna, but it is reasonable to expect the rf 
bandwidth of this mixer to be similar to that of waveguide designs: about one-half octave. 

The slot antennas have a low antenna impedance. They are coupled to the SIS through 
tapered transmission lines which reduce the impedance even further. Finally, the radiation 
impedance is reduced even further since the two slots appear in parallel across the SIS. The source 
admittance of radiation presented to the SIS at the design frequency is Y$ = (4 Q)" 1 . As a result, 
an SIS with a large junction area, 2.3 (|im) 2 , can be used. The result is Trecdsb = 420 K at 



Third International Symposium on Space Terahertz Technology 



Page 515 





60- 
50 - 




■ 


Quasioptic 




■ 






o 


Waveguide 












(/) 




o 










c 














o 




o 






■ 




u 

sz 


40- 












CL 




o 


o 




■ " 




CD 














10 














O 


30- 


o 














o 




1 






> 
0) 


?0 - 


B 




■ 






o 

CD 

cr 


10 - 
- 


o 
o 

o 




o 

— ■ 1 "- 


o 

1 ' 1 





200 300 400 500 600 

Frequency (GHz) 



700 



800 



Figure 7. The noise of the receivers summarized in Table 1. Plotted is 
TRECDSB/(hf/2ke). This normalizes the receiver noise to its quantum limit value. 

492 GHz (the conference publication on this receiver should be reviewed for the most up to date 
information. It is reasonable to expect that the noise of this kind of receiver will drop quickly as 
more is learned about its design. 
4.6. Overall performance summary 

A summary of the best reported SIS receiver results is shown in fig. 7 and in Table 1. The 
most common results reported are Trecdsb, so those are tabulated. DSB values of gain or mixer 
temperature are included in the table when they are quoted in the literature. Generally, SSB values 
can be estimated by doubling Trecdsb. doubling Tmixdsb. and subtracting 3 dB from the DSB 
Gain. The receiver noises in fig. 7 are normalized to the quantum limit of Nrec = 1 for an ideal 
photodiode mixer. The noise values achieved are many times this limiting value. However, these 
noise values represent a very great improvement over values available just a few years ago. It is 
reasonable to expect that they will continue to fall in the future. 



Page 516 Third International Symposium on Space Terahertz Technology 

4 . 7. Unusual Mixers for Higher Frequencies 

This section is concluded with the mention of some unusual modes of superconducting 
mixer operation. None of these methods have been developed to the point that they are useful on a 
radio telescope. However, each method addresses problems with submillimeter SIS mixers in 
ways which may be useful in higher frequency superconducting mixers. 

4.7.1. Harmonic mixin g 

A mixer can be designed so that it is sensitive to input signals at fs = n fLO ± fiF where n is 
an integer. If n > 2, the mixer is said to be subharmonically pumped. Belitsky et al. investigated 
the gain of their spiral-transformer mixer with subharmonic pumps [53]. The mixer gain fell off 
slowly as n increased from 1 to 3. The advantages of subharmonic pumping include 1) it is easier 
to get powerful oscillators at lower LO frequencies and 2) the large separation between fLO and fs 
makes it easier to couple both of these to the SIS with high efficiency. 

4.7.2. SIN junction mixer 

If one of the superconducting sides of an SIS is replaced by a non-superconducting metal 
film, the resulting diode is a Superconductor-Insulator-Normal metal junction, or SIN. The SIN 
does not have as sharp a non-linearity in its IV as the SIS. But it also doesn't have any Josephson 
currents, which can be a problem in high frequency SIS mixers. 

An SIN mixer at 230 GHz was built and tested by Blundell and Gundlach [48]. Compared 
to a similar SIS mixer, its gain was about 7 dB lower, but its mixer noise was quite low. 
Considering the very low noise IF amplifiers now available (Tip ~ 3 K at 1 .5 GHz), this is a very 
promising result. While it does not give any advantage over an SIS at 230 GHz, its lack of 
Josephson currents might make the SIN mixer useful at submillimeter wavelengths. Theoretical 
work on the SIN mixer operated both as a fundamental mixer [36, 54] and as a subharmonically 
pumped mixer [55] suggest it should be capable of TrecdsB < 100 K up to 660 GHz. 

4.7.3. Josephson effect mixing 

While the Josephson effect is known to produce excess noise in submillimeter mixers, it is 
also capable of enhancing the gain of these mixers [33, 56]. While the current results for 
Josephson mixing show the noise increasing faster than the gain, the mixers tested are not 



Third International Symposium on Space Terahertz Technology Page 517 

designed to take advantage of the Josephson currents. Much work remains to be done on this type 
of mixer before it is known if it will be useful. 

5 . SUMMARY AND CONCLUSIONS 
SIS mixers are used extensively for 100 to 500 GHz radio astronomical receivers. Many 
useful techniques have been developed for using SIS's at submillimeter wavelengths. The 
waveguide-feedhorn and the planar-antenna-lens (quasioptical) techniques provide excellent 
radiation beam patterns for SIS mixers. Integrated tuning elements including inductive 
components and transmission-line-transformers allow impedance matching of the SIS to both of 
these radiation coupling structures. SIS mixers using these techniques have been reviewed here. 

6. ACKNOWLEDGEMENTS 
The author thanks the many researchers who sent letters and reprints which helped him to 
write this review. The development of this article was supported by National Science Foundation 
Grant ECS-8857868. 

7. REFERENCES 



[1] P. L. Richards, T. M. Shen, R. E. Harris and F. L. Lloyd, "Quasiparticle heterodyne 

mixing in SIS tunnel junctions," Appl. Phys. Lett., vol. 34, 1 March, 1979, pp. 345-347. 

[2] G. J. Dolan, T. G. Phillips and D. P. Woody, "Low-noise 1 15-GHz mixing in 

superconducting oxide-barrier tunnel junctions," Appl. Phys. Lett., vol. 34, 1 March, 
1979, pp. 347-349. 

[3] J. R. Tucker, "Quantum limited detection in tunnel junction mixers," IEEE J. Quantum 
Electron., vol. QE-15, November, 1979, pp. 1234-1258. 

[4] D. P. Woody, R. E. Miller and M. J. Wengler, "85-1 15 GHz receivers for radio 

astronomy," IEEE Trans. Microwaves Theory and Techniques, vol. MTT-33, February, 
1985, pp. 90-95. 

[5] E. C. Sutton, "A superconducting tunnel junction receiver for 230 GHz," IEEE Trans. 
Microwaves Theory and Techniques, vol. MTT-31, July, 1983, pp. 589-592. 

[6] B. N. Ellison and R. E. Miller, "A low noise 230 GHz receiver," Int. J. ofIR and MM 
Waves, vol. 8, 1987, pp. 608-625. 



Page 518 Third International Symposium on Space Terahertz Technology 



[7] R. Blundell, M. Carter and K. H. Gundlach, "A low noise SIS receiver covering the 

frequency range 215-250 GHz," Intl. J. ofIR and MM Waves, vol. 9, April, 1988, pp. 
361-370. 

[8] T. H. Buttgenbach, R. E. Miller, M. J. Wengler, D. M. Watson and T. G. Phillips, "A 
broad-band low-noise SIS receiver for submillimeter astronomy," IEEE Transactions on 
Microwave Theory and Techniques, vol. 36, December, 1988, pp. 1720-1726. 

[9] S.-K. Pan, A. R. Kerr, M. J. Feldman, A. W. Kleinsasser, J. W. Stasiak, R. L. 

Sandstrom and W. J. Gallagher, "An 85-1 16 GHz SIS receiver using inductively shunted 
edge-junctions," IEEE Trans. Microwaves Theory and Techniques, vol. 37, March, 1989, 
pp. 580-592. 

[10] D. P. Woody, C. J. Giovanine and R. E. Miller, "Dual channel 1 15 and 230 GHz SIS 
receivers in operation at the Owens Valley Radio Observatory," IEEE Trans. Magn., vol. 
25, March, 1989, pp. 1366-1370. 

[11] A. R. Kerr and S.-K. Pan, "Some recent developments in the design of SIS mixers," Int'l 
J. ofIR and MM Waves, vol. 11, October, 1990, pp. 1169-1187. 

[12] H. Ogawa, A. Mizuno, H. Hoko, H. Ishikawa and Y. Fukui, "A 1 10 GHz SIS receiver 
for radio astronomy," Intl. J. ofIR and MM Waves, vol. 11, June, 1990, pp. 717-726. 

[13] J. W. Kooi, M. Chan, T. G. Phillips, B. Bumble and H. G. Leduc, "A low noise 230 
GHz heterodyne receiver employing .25," vol. 1991, 

[14] H. G. LeDuc, S. K. Khanna, J. A. Stern and S. Thakoor, "All refractory NbN/MgO/NbN 
tunnel-junctions," IEEE Trans. Magn., vol. 23, March, 1987, pp. 863-865. 

[15] A. W. Lichtenberger, C. P. McClay, R. J. Mattauch, M. J. Feldman, S.-K. Pan and A. R. 
Kerr, "Fabrication of Nb/Al-A^OyNb junctions with extremely low leakage current," 

IEEE Trans. Magn., vol. 25, March, 1989, pp. 1247-1250. 

[16] W. C. Danchi, E. C. Sutton, P. A. Jaminet and R. H. Ono, "Nb edge junction process for 
submillimeter wave SIS mixers," IEEE Trans. Magn., vol. 25, March, 1989, pp. 1064- 
1067. 

[17] J. A. Stern, B. D. Hunt, H. G. LeDuc, A. Judas, W. R. McGrath, S. R. Cypher and S. 

K. Khanna, "NbN/MgO/NbN SIS tunnel junctions for submm wave mixers," IEEE Trans. 
Magn., vol. 25, March, 1989, pp. 1054-1059. 

[18] A. H. Worsham, D. E. Prober, J. H. Kang, J. X. Przybysz and M. J. Rooks, "High 

quality sub-micron Nb Trilayer tunnel junctions for a 100 GHz SIS receiver," IEEE Trans. 
Magn., vol. 27, March, 1991, pp. 3165-3167. 

[19] T. Lehnert, C. Grassl, K. H. Gundlach and J. Blondel, "Nb-Aloxide-Nb junctions for 3- 
mm SIS receivers," ISEC, vol. 1991, 

[20] C. K. Walker, J. W. Kooi, M. Chan, H. G. LeDuc, J. E. Carlstrom and T. G. Phillips, 
"A 492 GHz SIS waveguide receiver for submillimeter astronomy," Intl. J. ofIR and MM 
Waves, vol. submitted, 1991, 



Third International Symposium on Space Terahertz Technology Page 519 



[21] B. N. Ellison, P. L. Schaffer, W. Schaal, D. Vail and R. E. Miller, "A 345 GHz receiver 
for radio astronomy," Int. J. ofIR and MM Waves, vol. 10, 1989, pp. 937-947. 

[22] E. C. Sutton, W. C. Danchi, P. A. Jaminet and R. H. Ono, "A superconducting tunnel 
junction receiver for 345 GHz," Intl. J. ofIR and MM Waves, vol. 11, February, 1990, 
pp. 133-149. 

[23] J. Zmuidzinas and H. G. LeDuc, "Quasi-optical slot antenna SIS mixers," IEEE Trans. 
Microwaves Theory and Techniques, vol. Submitted, 1991, 

[24] H. Rothermel, D. Billon-Pierron and K. H. Gundlach, "An open structure SIS mixer for 
350 GHz," Digest ofthel6th Int. Conf. on Infrared and Millimeter Waves, Lausanne, vol. 
August, 1991, 

[25] M. J. Wengler, "Submillimeter wave detection with superconducting tunnel diodes," Proc. 
IEEE, vol. Accepted, 1992, 

[26] J. R. Tucker and M. J. Feldman, "Quantum detection at millimeter wavelengths," Rev. 
Mod. Phys., vol. 57, 1985, pp. 1055-1113. 

[27] T. G. Phillips, "Submillimeter and far-infrared detectors," Astro. Lett, and 
Communications, vol. 26, 1988, pp. 293-304. 

[28] P. L. Richards and Q. Hu, "Superconducting components for infrared and millimeter-wave 
receivers," Proc. IEEE, vol. 77, August, 1989, pp. 1233-1246. 

[29] R. Blundell and D. Winkler, "The superconductor insulator superconductor mixer receiver 
- a review," in NATO Applied Research Workshop on Superconducting Electronics and 
2nd Workshop on Josephson Devices. Capri, Italy: 1990. 

[30] D. Winkler, Z. Ivanov and T. Claeson, "Superconducting detectors for mm and sub-mm 
waves," in 2nd Nordic Symposium on Superconductivity. R0ros, Norway: World 
Scientific, 1991. 

[31] J. W. Archer, "Low-noise heterodyne receivers for near-millimeter wave radio 
astronomy," Proc. IEEE, vol. 73, 1985, pp. 109-130. 

[32] J. M. Payne, "Millimeter and Submillimeter Wavelength Radio Astronomy," Proc. IEEE, 
vol. 77, 1989, pp. 993-1017. 

[33] M. J. Wengler, D. P. Woody, R. E. Miller and T. G. Phillips, "A low noise receiver for 
millimeter and submillimeter wavelengths," Intl. J. ofIR and MM Waves, vol. 6, August, 
1985, pp. 697-706. 

[34] S. Withington and E. L. Kollberg, "Spectral-domain analysis of harmonic effects in 

superconducting quasiparticle mixers," IEEE Trans. Microwaves Theory and Techniques, 
vol. 37, January, 1989, pp. 231-238. 

[35] D.-G. Crete, W. R. McGrath, P. L. Richards and F. L. Lloyd, "Performance of arrays of 
SIS junctions in heterodyne mixers," IEEE Trans. Microwaves Theory and Techniques, 
vol. MTT-35, April, 1987, pp. 435-440. 

[36] M. J. Wengler and D. P. Woody, "Quantum noise in heterodyne detection," IEEE J. of 
Quantum Electron., vol. QE-23, May, 1987, pp. 613-622. 



Page 520 Third International Symposium on Space Terahertz Technology 



[37] Q. Hu, C. A. Mears, P. L. Richards and F. L. Lloyd, "Measurement of integrated tuning 
elements for SIS mixers with a Fourier transform spectrometer," Intl. J. ofIR and MM 
Waves, vol. 9, April, 1988, pp. 303-320. 

[38] A. R. Kerr, S.-K. Pan and M. J. Feldman, "Integrated tuning elements for SIS mixers," 
Intl. J. ofIR and MM Waves, vol. 9, February, 1988, pp. 203-212. 

[39] D. Winkler, N. G. Ugras, A. H. Worsham, D. E. Prober, N. R. Erickson and P. F. 

Goldsmith, "A full-band waveguide SIS receiver with integrated tuning for 75-110 GHz," 
IEEE Trans. Magn., vol. 27, March, 1991, pp. 2634-2637. 

[40] V. Y. Belitsky, M. A. Tarasov, S. A. Kovtonjuk, L. V. Filippenko and O. V. 

Kaplunenko, "Low noise completely quasioptical SIS receiver for radioastronomy at 1 15 
GHz," 21st European Microwave Conference, Stuttgart, vol. September, 1991, 

[41] J. A. Carpenter, A. D. Smith, E. R. Arambula, L. P. S. Lee , T. Nelson and L. Yujiri, 
"100 GHz SIS mixer with improved rf matching," IEEE Trans. Magn., vol. 1991, 

[42] T. H. Buttgenbach, H. G. LeDuc, P. D. Maker and T. G. Phillips, "A Fixed Tuned 
Broadband Matching Structure For Submillimeter SIS Receivers," IEEE Trans. Appl. 
Super conduct.., vol. submitted, 1992, 

[43] A. B. Ermakov, V. P. Koshelets, I. L. Serpuchenko, L. F. Filippenko, S. V. Shitov and 
A. N. Vystavkin, "SNAP structures with Nb-AlO-Nb junctions for mm wave receivers," 
IEEE Trans. Magn., vol. 25, March, 1989, pp. 1060-1064. 

[44] W. R. McGrath, J. A. Stern, H. H. S. Javadi, S. R. Cypher, B. D. Hunt and H. G. 
LeDuc, "Performance of NbN Superconductive tunnel junctions as SIS mixers at 205 
GHz,"/£££ Trans. Magn., vol. 27, March, 1991, pp. 2650-2653. 

[45] B. M. Thomas, "Design of corrugated conical horns," IEEE Trans. Antennas Propagat., 
vol. AP-26, March, 1978, pp. 367-372. 

[46] P. F. Goldsmith, "Quasi-optical techniques at millimeter and submillimeter wavelengths," 
in Infrared and Millimeter Waves, K. J. Button, Editor. 1982, Academic Press: New York, 
pp. 277-343. 

[47] M. C. Carter, S. Navarro, A. Karpov, D. Billon-Pierron, T. Lehnert and K. H. Gundlach, 
"Receiver development with Nb/Al-Oxide/Nb SIS mixers in the frequency ranges of (201- 
270) and (320-370) GHz," Intl. J. ofIR and MM Waves, vol. in press, 1991, 

[48] R. Blundell and K. H. Gundlach, "A quasiparticle SIN mixer for the 230 GHz frequency 
range," Intl. J. ofIR and MM Waves, vol. 8, December, 1987, pp. 1573-1579. 

[49] T. H. Buttgenbach, T. D. Groesbeck and B. N. Ellison, "A scale mixer model for SIS 
waveguide receivers," Intl. J. ofIR and MM Waves, vol. 11, January, 1990, pp. 1-20. 

[50] M. J. Feldman and S. Rudner, "Mixing with SIS arrays," in Reviews of Infrared and 
Millimeter Waves, K. J. Button, Editor. 1983, Plenum: pp. 47-75. 

[51] A. D. Smith and P. L. Richards, "Analytic solutions to superconductor insulator 

superconductor quantum mixer theory," J. Appl. Phys., vol. 53, May, 1982, pp. 3806- 
3812. 



Third International Symposium on Space Terahertz Technology Page 521 



[52] D. B. Rutledge, D. P. Neikirk and D. P. Kasilingam, "Integrated-circuit antennas," in 

Infrared and Millimeter Waves, K. J. Button, Editor. 1983, Academic Press: New York, 
pp. 1-90. 

[53] V. Y. Belitsky, I. L. Serpuchenko, M. A. Tarasov and A. N. Vystavkin, 

"Subharmonically pumped SIS mixer," in 19th European microwave conference. London, 
UK: 1989. 

[54] C. E. Tong, L. M. Chernin and R. Blundell, "Harmonic mixing in a superconducting 
tunnel junction,"/. Appl. Phys., vol. 68, 15 October, 1990, pp. 4192-4198. 

[55] L. M. Chernin and R. Blundell, "Harmonic mixing in a superconductor-insulator-normal 
metal tunnel junction receiver," /. Appl. Phys., vol. 69, 15 February, 1991, pp. 2682- 
2684. 

[56] M. J. Wengler, N. Dubash, G. Pance and R. E. Miller, "Josephson effect gain and noise 
in SIS mixers," IEEETrans. Microwaves Theory and Techniques, vol. 40, in press, 1992, 



Page 522 Third International Symposium on Space Terahertz Technology 

. trzrft Evaluation of integrated tuning elements with SIS devices. 

N93-27770 






\p 



M.M.T.M. Dierichs 1 , C.E. Honingh 2 , R.A. Panhuyzen 2 , BJ. Feenstra 1 , A. Skalare 2 ^, JJ. 
Wijnbergen 2 , H. v.d. Stadt 2 , Th. de Graauw 2 . 



1: Dept. of Applied Physics and Materials Science Centre, University of Groningen, 
Nijenborgh 4, 9747 AG Groningen, The Netherlands. 

2: Space Research Organization of the Netherlands, Groningen, 
Landleven 12, 9747 AD Groningen, The Netherlands. 

3: Dept, of Applied Electron Physics, Chalmers University of Technology, 
GSteborg, Sweden. 



Abstract. 



The resonance of integrated tuning stubs in combination with SIS detectors is measured 
and modeled. The predicted resonances are compared with measurements of stubs 
integrated with Nb/Al 2 3 /Nb junctions in a log-periodic antenna using a Michelson 
interferometer. Different stub lengths were made on different substrates (on 200 nm 
thick quartz and on a 7 jrni thick silicon membrane) and the results show a fairly good 



Third International Symposium on Space Terahertz Technology Page 523 

^agreement with the model calculations. Quartz substrates showed resonances up to 580 
GHz, silicon membrane stub resonances reach as high as 480 GHz. An observed 
resonance at 560 GHz is probably a substrate effect from the membrane. The gap 
frequency for all the samples is 650 GHz and no resonances are detected above this 
frequency. Up to the maximum detected frequency dispersion is found to be negligible. 



I Introduction. 



SIS mixers with Nb/Al 2 3 /Nb junctions are very sensitive submm detectors. Recent 
progress in SIS mixer development is due to the ability to manufacture smaller junctions 
down to sub-micron dimensions 1,2 . Instead of continuing to put more effort into the 
fabrication of smaller junctions and thus reducing the junction capacitance, it is also 
possible to implement integrated tuning elements, which are fairly easy to fabricate and 
result in a high sensitivity and broad bandwith. It has been shown that junctions with 
integrated tuning used in submm-wave mixers give good results 3 ' 4 . 
The first published stub measurements used the self-pumped steps in the I-V 
characteristic to measure the resonance of the stub 5 . A more accurate and complete 
evaluation can be performed with a wide-band Michelson interferometer as first shown 
by Hu et. al. 6 . 

In this paper we first describe our design criteria for niobium stubs in combination with 
niobium junctions. Next, we describe how they can be analyzed on a log-periodic antenna 
with two 1 Mm 2 junctions in series. Each junction has its own stub. Stubs for 100 GHz 



pa e 524 Third International Symposium on Space Terahertz Technology 



and 350 GHz have been designed. The first type is expected to have multiple resonances 
from which the dispersion in niobium can be calculated. These antennas are made on 
200 /xm thick quartz substrates and on 7 /im silicon membranes. The results are used to 
separate stub and antenna resonances and to estimate the dispersion. 
The organization of this paper is as follows: the theoretical background will be 
introduced in Sec.II, the fabrication results are presented in Sec.III, the experimental 
details are described in Sec.IV, the comparison between theory and experiment is 
discussed in Sec.V, and the conclusion will be drawn in Sec. VI. 



II Model calculation. 



To tune out the geometric capacitance an inductive tuning element was used. An 
example of the devices studied is shown in Fig.l. Two junctions in series, placed in the 
center of a log-periodic antenna, were used. To each junction a stripline type inductor 
is attached. The total arrangement can be modelled with the circuit shown in Fig.2. For 
completeness the connecting strip between the two junctions is included as an inductor 
Lieads- I n practice this inductance is negligible in evaluating the frequency response. 
Using integrated tuning, the junction impedance can not simply be described as a pure 
resistor with a parallel capacitor. Instead, it must be described as a capacitor in parallel 
with a complex admittance with a conductive part (G Q ) and a susceptive part (B Q ). Since 
the experiment works in the small signal limit and the Josephson effect is suppressed by 
a magnetic field, the junction admittance can be described as follows 7,8 : 



Third International Symposium on Space Terahertz Technology Page 525 



G °- ■&5- lz - ir '* i T ) "'-'"i-*?" <l> 



These equations show that the resonant frequency depends on the bias voltage and the 
photon-energy. In these equations is >iS>/e the energy of the photon step, V is the bias 
voltage, and 

where 1,^ is the Kramers-Kronig transform, which can be calculated from the dc-IV 

curve. 

The inductance per unit length of the stub can be calculated as follows 9 : 

L <- &■"*♦ — hr * — h~ ] <« 

tanh(-ri) tanh(-r^) 

where w is the width of the stub, k is the fringing factor 10 , t x , tj and t d are the thicknesses 
of the ground plane, the stub, and the dielectric layer respectively, and X. is the 
penetration depth of the niobium layers. The capacitance per unit length of the stub is 
given by: 

c = jt-» •« 'Jul '^' 

Knowing the capacitance and the inductance of the stub,^the impedance Z„ and the 
phase velocity v follow from the definitions: 



Page 526 Third International Symposium on Space Terahertz Technology 



Zo = 






v = X (7) 



The impedance of a transmission line with small loss and dispersion follows from: 

Z= 2 + 2 (8) 

sin 2 (-^) j.tan(-^l) 

Where a is the loss per unit length and 1 is the length of the stub. The RF coupling 
coefficient C^p defined as the fraction of the available power dissipated in the junction 
is given by: 



Crf= 1 ' 



**-£ 



Ya+Yj 



(9) 



where Y A = 1/R A is the simplified admittance of the antenna (1/120 n), and Yj is the 

admittance of the right hand side of Fig.2. 

If both areas and stub lengths are equal then the resonant frequency can be 

approximated from: 

<•>• C: + i_ + i_ = o 

J <o- x s <■»• X q 

(10) 

In our equations we do not take into account the behaviour of the log-periodic antenna, 

this is rather complex and not known in all detail. Therefore it will be very difficult to 
identify the loss from the observed resonances, because it could be loss in niobium, bad 
coupling to the antenna or a combination of both. 



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Page 527 




*ig.l. Photo of a log-periodic antenna with two junctions in series. Each junction has 
its own stub. 



I ^junctic 



5 



stub ,-J-, * quantum L K quantum 



L leads 







1 stuDJU * quantum L ^ quantum 



Fig.2 Electrical equivalent of two junctions in series with integrated tuning elements. 



page 528 Third International Symposium on Space Terahertz Technology 



III Fabrication of devices. 



The detector is positioned in the center of a broad-banded log-periodic antenna. We 
used 2 junctions in series of each 1 /im 2 and a current density of 12000 A/cm 2 . On top 
of the junctions the wiring layer was defined with a stub for each junction (Fig.l.). The 
dielectric layer between the ground plane (antenna) and the stub was 250 nm thick 
sputtered SiO z . The junctions have been fabricated with the Selective Niobium Over-Etch 
Process (SNOEP) 11 . 

Antennas have been fabricated on 200 /xm thick quartz substrates and on 7 Mm thick 
silicon membranes (Fig.3.). The membranes have been etched in ethylenediamine- 
pyrocatechol-water (EPW) 12 . The junctions on the membranes were fabricated after the 
etching of the membranes. With the obtained thickness, the membrane is transparent 
which simplifies the alignment of the antenna on the membrane. 
Two different stub lengths were fabricated for different purposes. Firstly, short stubs 
(around 120 /ira) were designed to resonate at 350 GHz. A single resonance simplifies 
the comparison with the model and it can easily be implemented in the waveguide mixer 
chip design. Secondly, long stubs (around 500 /im) were designed to have a fundamental 
resonance around 100 GHz and multiple resonances at higher frequencies. From the 
frequencies of the resonances in principle the dispersion and loss in niobium can be 
estimated. 



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Page 529 




Fig.3. Photo of the antenna fabricated on a 7 nm thick silicon membrane. 



IV Measurement set-up 



For measuring the response of the detector, we used a Michelson interferometer with 
a Hg arc lamp as source 13 (Fig.4.). The operating frequency range was determined by a 
50 urn thick kapton film beam-splitter. The mechanical traveling distance was 50 mm 
resulting in a resolution of 4 GHz. Both single sided and double sided interferograms 
were measured. An example of the resulting spectra with multiple resonances is plotted 
in Fig.5. 
The antenna was mounted in a liquid helium dewar with dc-bias connections. Since the 



G-7 



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Third International Symposium on Space Terahertz Technology 




Michelson 
interferometer 



S = source (Hg-arc) 
Ml = mirror 1 

(moving) 
M2 = mirror 2 

(static) 
B = beam splitter 



4.2 K Dewar 



"ig.4. Schematic of the Michelson interferometer. 



Michelson was not under vacuum during the measurement, the water absorption lines 
at 380 GHz, 448 GHz, 557 GHz and 752 GHz were visible when sufficient resolution was 
used. All antennas fabricated on a quartz substrate were glued to a quartz hyper- 
hemispherical lens. The lens optimizes the optical coupling and results in a better 
sensitivity. We have also performed measurements with the log-periodic antenna on a 
thin membrane. In that case no lens was used, resulting in a much lower signal. By using 
longer integration times we could improve the signal to noise ratio to an acceptable 
level. 



Third International Symposium on Space Terahertz Technology 



Page 531 



7 














6 


- 






~ 5 










a 
~ 4 




\ I 






>% 

! » 

*» . 

a 

2 




| 






1 




\ \ A 















( 


> 100 200 300 400 S00 600 700 


800 900 1000 


Frequency (GHz) 




7 ig.5 Spec 


trogram of a device 


with multiple resonances. 







V Results 



A: Short stubs on 200 /im quartz substrates. 



Antennas with stub lenghts around 120 pm were investigated to determine the specific 
capacitance of the junction and the penetration depth of the niobium layers. Resonant 
frequencies were measured from two different batches. The resonant frequency is 
depending on the bias voltage due to the behaviour of the quantum impedance (eq.l and 
eq.2). Calculated and measured results are shown in Fig.6. Best agreement between 



Page 532 



Third International Symposium on Space Terahertz Technology 



theory and experiment was obtained with the assumption of a specific capacitance of 55 
fF/Mm 2 and a penetration depth of 100 nm. These values were further used in 
calculations of the long stubs. Differences between calculations and measurements are 
due to the noise in the spectrogram which complicates the determination of the resonant 
frequencies. No multiple resonance is observed. 















~ theory theory ♦ measure • meainre 






6A1 9A2 6A1 9A2 




AdSJ 










400 


- 






"n 

3 


380 




/ * \ 

its \ 

/ ' * \ 

/ / \ \ 

/ ' * \ 

/ ' * \. 




> 


360 








I 

Li. 


340 
320 




j^^O 








300 




1 1 1 1 1 




0.00 


0.50 1.00 1.50 2.00 2.50 3.00 
Bias Voltage (mV) 


Fie.6 


Calculi 


ited 


and measured resonant frequencies as a function of bias vc 


tltage 



for two different junction batches. 



B; Long stubs on 200 \xm quarz substrates. 



Next, the resonances of a 527 ^m long stub on a quartz substrate were measured. The 
results of the measurements at different bias voltages are plotted in Fig.7 and are 



Third International Symposium on Space Terahertz Technology 



Page 533 



compared with model calculations. At the first two resonances the measurements agree 
fairly well with the model both below as well as above the gap voltage. The two higher 
resonances have a larger frequency shift close to the gap than the model predicts. More 
measurements with different lengths are planned for a more detailed evaluation. Stub 
resonances up to 580 GHz are observed, while no antenna resonances are visible. We 
do not see any dispersion in the resonances of the stubs (Fig.5). 



calc 



calc 



calc 



calc 



a meat • meat + meat ♦ meaf 



500 



410 



«♦♦♦♦♦♦♦♦♦♦ 



^♦♦W'W*'*" 




Fig.7 Calculated and measured resonant frequencies for an antenna with a 527 nm 
long stub on a quartz substrate as a function of bias voltage. 



Page 534 



Third International Symposium on Space Terahertz Technology 



C: Long stubs on 7 pm silicon membranes. 

The resonances of a 527 j*m long stub on a 7 nm thick silicon membrane were measured. 
A double sided interferogram was used to decrease the noise in the spectrogram. The 
resolution (8 GHz) is lower because of the decreased scan length. The effect of the 
quantum susceptance is not clearly observed because of the loss in resolution and signal. 
We measured stub resonances at 110, 200 and 325 GHz. Incidentally resonances at 165, 
310 and 540 GHz occur which are probably substrate resonances. The highest stub 
resonance appears to be at 450 GHz which is lower than the measurements on quartz. 
The lowest three resonances are compared with the model calculations in Fig.8. It is not 
clear why above 330 GHz no well defined resonances occur except for the substrate 
resonance at 540 GHz. 



M 

x 



>> 

O 

a 
e 

3 
9 

o 
u 

a. 


400 
330 
260 
190 
120 




cal — 




• me 


+ 


me 






cal ' cal ^ me 








+ ¥-++*^ 


++ 












• +*"$ 


• 




— 




A J^ 


A* 

1 






i 


+KMA 

i i i i 






SO 

c 


> 


1 


2 3 4 5 
Biaa voltage (mV) 


6 


7 


1 


I 



Fig.8 Calculated and measured resonance frequencies for a 527 urn long stub 
on a silicon membrane as a function of bias voltage. 



Third International Symposium on Space Terahertz Technology Pa i« 535 



VI Conclusions 



The theoretical model and the experimental results for the short stubs lead to a 
penetration depth of 100 nm and a specific capacitance of 55 fF/nm 2 . This is 
independent of the measured batch. For long stubs the model predicts a different 
behaviour at higher resonances than is measured. The measured frequency shift at the 
higher resonances is larger than the model predicts. Both below and above the gap 
voltage the resonances agree fairly well with the model. Resonances up to 580 GHz are 
observed. For antennas fabricated on 7 nm thick silicon membranes resonances up to 480 
GHz are observed. Also possible substrate resonances are measured. No resonances 
above 600 GHz are observed, which is close to the gap frequency of niobium (650 GHz). 



Acknowledgements 



We thank T.M. Klapwijk for his stimulating discussions, H.G. Golstein and G. de Groot 
for their help with the Michelson interferometer, MJ. de Boer for etching the 
membranes, G. de Lange for his support with the model calculation and H.H.A. 
Schaeffer for the mechanical support. This work is supported by the Stichting Technische 
Wetenschappen (STW) and the Stichting voor Fundamenteel Onderzoek der Materie 
(FOM), which are part of the Nederlandse Organisatie voor Wetenschappelijk 



Page 536 Third International Symposium on Space Terahertz Technology 

Onderzoek (NWO). We also acknowledge the financial support of the European Space 
Research (ESA) through contract 7898/88/NL/PB(SC). 



References 



1. W.R. McGrath, H.H.S. Javadi, S.R. Cypher, B. Bumble, B.D. Hunt, and H.G. LeDuc, 
Proceedings of Second International Symposium on Space TeraHertz Technology, 423 
(1991) 

2. J.W. Kooi, M. Chan, T.G. Phillips, B. Bumble, and H.G. LeDuc, Proceedings of 
Second International Symposium on Space TeraHertz Technology, 459 (1991) 

3. A.W. Lichtenberger, D.M. Lea, A.C. Hicks, J.D. Prince, R. Densing, D. Peterson, and 
B.S. Deaver, Proceedings of Second International Symposium on Space TeraHertz 
Technology, 439 (1991) 

4: J. Zmuidzinas, and H.G. LeDuc, Proceedings of Second International Symposium on 
Space TeraHertz Technology, 481 (1991) 

5. A. V. Raisanen, W.R. McGrath, P.L. Richards, and F.L. Lloyd, IEEE Trans Microwave 
Theory Tech. 33, 1495 (1985) 

6. Q. Hu, C.A. Mears, P.L, Richards, and F.L. Lloyd, Int. J. of Infrared and MM Waves, 
9, 303 (1988) 

7. J.R. Tucker, and MJ. Feldman, Rev. Mod. Phys. 57, 1055 (1985) 

8. Q. Hu, C.A. Mears, PL. Richards, and F.L. Lloyd, Phys. Rev. B, 42,10250 (1990) 

9. J.C. Swihart, J. Appl. Phys. 32, 461 (1961) 



Third International Symposium on Space Terahertz Technology Page 537 

10. W.H. Chang, J. Appl. Phys. 50, 8129 (1979) 

1 1. M.M.T.M. Dierichs, R.A. Panhuyzen, C.E. Honingh, M J. de Boer, and T.M. Klapwijk 
(unpublished results) 

12. K.E. Petersen, Proc. of the IEEE 70, 420 (1982) 

13. P.L. Richards, in Spectroscopic Technique for Far Infrared Submillimeter and MiUmeter 
Waves, (North-Holland, Amsterdam, 1967) 



Page 538 Third International Symposium on Space Terahertz Technology 

Syf-33 

/£o6&J SOURCE CONDUCTANCE SCALING FOR HIGH FREQUENCY 
« SUPERCONDUCTING QUASIPARTICLE RECEIVERS 

N93-27771 

Qing Ke 1 and M. J. Feldman 2 

Physics Department 1 and Department of Electrical Engineering 2 
University of Rochester, Rochester, NY 14627 



/ 



ABSTRACT 

It has been suggested that the optimum source conductance G s for the supercon- 
ductor-insulator-superconductor (SIS) quasiparticle mixer should have a 1/f dependence. 
This would imply that the critical current density of SIS junctions used for mixing should 
increase as frequency squared, a stringent constraint on the design of submillimeter SIS 
mixers, rather than in simple proportion to frequency as previously believed. We have 
used Tucker's quantum theory of mixing for extensive numerical calculations to 
determine G s for an optimized SIS receiver. We find that G s is very roughly independent 
of frequency (except for the best junctions at low frequency), and discuss the implications 
our results for the design of submillimeter SIS mixers. 



INTRODUCTION 

Superconductor-insulator-superconductor (SIS) quasiparticle mixers [1] are now 
firmly established as the most sensitive receiving devices in the vicinity of 100 to 200 
GHz. Their behavior is well described by Tucker's quantum theory of mixing [2]. There 
are now many publications which show excellent agreement between the theory's 
predictions of a mixer's conversion properties and experimental results, especially at 100 
GHz, and the theory also appears to be successful in predicting the noise temperature of 
the most sensitive SIS mixers. 

Given the success of the Tucker theory at 100 GHz, it is desirable to know the pre- 
dicted performance of SIS mixers at higher frequencies, where there are fewer 
experimental results but many experiments underway. A large step in this direction was 
taken by Kerr and Pan [3], who developed a "design procedure" for SIS mixers, really a 



Third International Symposium on Space Terahertz Technology Page 539 



set of rules for scaling a successful and reasonably understood low-frequency SIS mixer 
design to higher frequency. Their argument was carried further and ratified in Ref. [4]. 
Kerr and Pan concluded that the critical current density of SIS junctions used for mixing 
should increase as frequency squared, rather than in simple proportion to frequency as 
previously believed. This result presents a stringent constraint on the design of 
submillimeter SIS mixers, implying that high frequency SIS mixers are much more 
difficult to realize than had previously been appreciated. This widely quoted conclusion 
certainly is influencing the design of the current generation of submillimeter SIS mixers. 

Kerr and Pan based their analysis on the "coRnC = 4 rule": the best SIS mixer 
performance appears to be obtained when the characteristic parameter ©RnC is near 4, 
where w is the LO frequency and Rn is the normal state resistance and C the capacitance 
of the SIS junction. As first advanced in Ref. [5] and more recently discussed in Ref. 
[6], all SIS mixer experiments exhibiting infinite available gain have coRnC ^ 4, while 
coRnC < 1 has always resulted in considerable conversion loss. (To our knowledge this 
correlation still holds to date.) Presumably, good mixer conversion requires the reduction 
of harmonic conversion effects by the relatively large capacitance. Indeed, computer 
simulations show that harmonic conversion becomes significant for coRnC < 4 [7]. On 
the other hand, unnecessarily large capacitance entails greater difficulty in tuning and 
narrower bandwidth. 

The damping time RnC of an SIS junction varies in inverse proportion to its critical 
current density, j c - Therefore j c must increase proportional to frequency to maintain a 
constant coRnC, and this alone requires an inconveniently large j c for submillimeter SIS 
mixers. However, Kerr and Pan rightly note that while the coRnC = 4 rule may be valid 
for 100 GHz SIS mixers, there is no reason to expect that the optimum ©RnC is 
independent of frequency. In particular, their calculations indicate that the quantity GsRn. 
the mixer source conductance normalized to Rn, "should have a 1/f dependence for 
mixers in the quantum-limited regime." This immediately implies that j c should increase 
as frequency squared. 

CALCULATIONS 

It is not feasible to optimize an SIS mixer by maximizing the calculated conversion 
gain. There is no unique optimum bias point: the quantum theory of mixing predicts 
infinite gain for high quality SIS junctions over a wide range of parameter values. Such 
high gain is unrealistic and undesirable. Kerr and Pan avoid this difficulty by positing a 
set of requirements, including unity gain and moderately well matched input (VSWR £ 
2), for optimum SIS design. We take a different approach. 



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Third International Symposium on Space Terahertz Technology 



We use the quantum mixer theory for extensive numerical calculations, to determine 
the minimum value of the SSB (single sideband) noise temperature Tr of an SIS receiver , 
subject to reasonable experimental constraints. Thus our calculation involves a trade-off 
between minimizing the mixer noise temperature and maximizing the mixer conversion 
gain, which is mediated by the noise temperature of the IF amplifier Tip. Full details will 
appear elsewhere. For our current purpose we make the following approximations: We 
consider DSB (double sideband) operation in the three-frequency low-IF approximation, 
which should be a fairly good representation of most well-designed experimental mixers. 
We do not include any interference from the Josephson effect, although this is likely to be 
a problem for experiments at the higher frequencies. In addition, we ignore all 
reactances. Taken together, these approximations are equivalent to assuming 1) that the 
geometrical capacitance of the SIS junction is large enough to both short out the LO 
harmonics and their sidebands and to eliminate Josephson interference, 2) that the 
capacitance is itself resonated by a relatively broadband external tuning circuit, so that the 
intrinsic junction nonlinearity is presented with a resistive embedding impedance at all 
relevant frequencies, and 3) that the quantum susceptance has no significant effect. This 
third assumption is controversial. It has recently been argued that the quantum 
susceptance is a central element of the behavior of SIS mixers [8]. Nevertheless, we 
believe that this nonlinear reactance has little effect on the performance of an optimized 
SIS receiver, though it may affect the optimum bias point. This question will be 
addressed in further research. 




Fig. 1. Three synthetic normalized I- V characteristics used for these calculations. 



Third International Symposium on Space Terahertz Technology 



Page 541 



The equations employed in the calculation of Tr are taken from Ref. [1] and will not 
be reproduced here. For convenience we assume zero physical temperature; the only 
serious effect of this is to ignore the thermal noise from the IF termination which is 
reflected from the mixer back into the IF amplifier. For real SIS receivers this can be an 
important contribution to the total noise. We require a reasonable input match: in 
particular we require that both the signal reflection gain and also the signal-to-image 
conversion gain be < 1/4 (which corresponds to VSWR < 3). We find that this constraint 
completely eliminates every instance of high conversion gain. What remains is a distinct 
solution with stable moderate realistic conversion gain and low mixer noise. Moreover 
we find that our quantitative results are extremely insensitive to the level of returned 
signal or image power allowed. These topics are discussed at length in Ref. [9]. 



Frequency (GHz), for V g = 3 mV 
50 100 500 1000 



2 

3 



8. 
E 



o 

Z 

> 

u 







■ 1 1 i 
■ A 






' /' 


1000 


- 


• /' 
'■/' i 






1 ' : 






/•' >* 


500 




r' 






r s 






pi 
hi 


100 




/ ' •' 
/ ' •' 


50 








f" 


t X 




Ji i 






/'/ 






Dull *~S* / 




10 






Medium •• / 


5 


/ / 
Sharp .••"' / 

. . . . i / 


2k •; 

1 

1 



1.05 0.1 0.5 1 2 

Normalized Frequency co/cOg 



Fig. 2. The SSB noise temperature of a DSB SIS receiver optimized at each frequency, 
calculated for the three I-V curves of Fig. 1, Gl= 0.3/Rn, Tip = 3 K, and V g = 3 mV. 



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Third International Symposium on Space Terahertz Technology 



We have performed these calculations for each frequency for a wide range of 
parameters, but only a few of the results can be presented here. The illustrations given in 
this paper use the synthetic SIS junction I-V curves depicted in Fig. 1. The "sharp" curve 
corresponds to the best experimental SIS I-V curves, the "medium" curve corresponds to 
a good quality junction, and the "dull" curve corresponds to a moderate quality junction. 
We normalize voltages to the energy gap voltage V g , conductances to the normal state 
resistance Rn, and frequencies to the energy gap frequency <o g = eV g //?. 



Frequency (GHz), for Vg = 3 mV 
50 100 500 1000 




0.05 0.1 0.5 1 2 

Normalized Frequency co/Wg 



Fig. 3. The normalized source conductance G s Rn required to optimize 
the receiver of Fig. 2, calculated for the three I-V curves of Fig. 1. 



Third International Symposium on Space Terahertz Technology Page 543 



RESULTS 

Figure 2 shows the minimum theoretical SSB noise temperature of a DSB SIS 
receiver with IF load conductance Gl = 0.3/Rn, Tip = 3 K, and V g = 3 mV, for the three 
I-V curves of Fig. 1. Figure 3 shows the optimum value of the normalized source 
conductance G s Rn required to achieve the minimum Tr. At lower frequencies (below the 
vertical rise in each curve) the mixer is biased on higher number photon steps and G s is 
relatively constant as expected for classical behavior. On the first photon step, however, 
the behavior of G s is quite different. At the lowest frequencies on the first step G s is 
strongly dependent on the I-V curve quality; for high quality junctions the optimum G s is 
quite large. As the frequency increases, the optimum G s gradually changes to approach a 
value = 0.7, for all three I-V curves at frequencies near 2co g . 

Figure 3 clearly shows that the optimum G s does not have a 1/f dependence. To 
emphasize this point, in Fig. 4 we plot the quantity G s co vs. co for the data of Fig. 3. The 
1/f dependence predicted by Kerr and Pan [3] would give horizontal lines in Fig. 4, and 
horizontal lines are nowhere seen. Rather, the optimum G s for the sharp curve is given 
by the empirical formula G s = 1/2 + 0.25/co for bias points on the first photon step. This 
behavior is quite widespread. For instance, Fig. 5 shows the the optimum G s computed 
for SIS receivers with various values of Tip, for the sharp I-V curve. The same empirical 
formula also works well when we consider different values of load conductance, I-V 
curves with considerable leakage current, etc. 

In order to better understand the behavior of the optimum G s , in Fig. 6 we compare 
it with all of the important "input" conductances in our calculations. It is seen that even 
though G s is determined by a trade-off between the gain and the shot noise, the optimum 
Gs is quite close to that which minimizes the shot noise, G s hot> Dut far from that which 
maximizes the gain, G s '. This surprising result can be explained by examination of the 
equations of the SIS mixer. On one hand, the dependence of the conversion gain upon 
Gs is given by a simple impedance matching formula which has its minimum at G s = IG S 'I; 
a fairly large mismatch therefore results in only a small decrease in gain. On the other 
hand, the mixer noise is minimized by the exact cancellation of the correlated components 
of the shot noise at the IF and the signal and image frequencies, which occurs at G s = 
Gshot- ^ Gs strays from this value the shot noise grows rapidly. The optimum G s is also 
far from the signal input conductance, Gs, but never more than a factor of three lest the 
signal reflection gain become too large. 



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Third International Symposium on Space Terahertz Technology 



O 



Frequency (GHz), for V g = 3 mV 

500 1000 1500 




0.5 1.0 1.5 

Normalized Frequency co/o) g 



Fig. 4. The data of Fig. 3 are multiplied by w and replotted (in normalized 
units), and are compared to an empirical formula. 



O 



2.0 
1.5 
1.0 
0.5 







Frequency (GHz), for V g = 3 mV 

500 1000 1500 



0.0 



1 ' ' r 



h K 



' 



0.0 




0.5 1.0 1.5 

Normalized Frequency ca/co g 



Fig. 5. The optimum source conductance G s of an SIS receiver whose IF 
amplifier noise temperature Tip = 10 K, 3 K, and K, respectively, using the 
"sharp" I-V curve of Fig. 1, G L = 0.3/R N , and V g = 3 mV. 



Third International Symposium on Space Terahertz Technology 



Page 545 



VI 

u 



"O 

c 
o 
U 















Frequency (GHz), for V g = 3 mV 



500 



1000 




— • -G° 



:."*•"•» 




■ 



1 



0.5 1 1.5 

Normalized Frequency oo/cog 



Fig. 6. The optimum source conductance G s of an SIS receiver using the 
"medium" I-V curve of Fig. 1, G L = 0.3/R N , Ti F = 3 K, and Vg = 3 mV, 
compared to various "input" conductances: Glo and Gs are the input con- 
ductances of the mixer at the LO and the signal frequencies, respectively, 
G S hot is the value of G s which would minimize the shot noise of the mixer, 
and G s ' is the value of G s which would maximize the gain of the mixer. 



Note in Fig. 6 that G s hot> and thus the optimum G s , follows closely the input 
conductance at the LO frequency, Glo- This is exactly as predicted by the simple 
photodiode theory of SIS mixing [10], which reproduces the the equations of the 
quantum theory of mixing in the limit of small LO voltage amplitude (small a). It is 
surprising that G S hot follows Glo so closely for the relatively large a of our simulations. 
In any case this enables us to explain the empirical formula G s = 1/2 + 0.25/co. In the 
limit of small a, Glo is the slope of the chord connecting the photon point LjcCVo " h®/e) 
to the photon point I<jc(Vo + frw/e) on the unpumped dc I-V curve. Therefore, using the 
preferred value for the optimum dc bias voltage Vo = 0.9 for the sharp I-V curve, this 
gives Glo = 1/2 + 0.35/co in the small a limit. G s follows but is slightly less than Glo 
(Fig. 6) and so is very well approximated by the empirical formula. 



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Third International Symposium on Space Terahertz Technology 



DISCUSSION 

The results presented here are for particular parameter values, but they are quite 
general and representative of our more extensive calculations. In disagreement with Ref. 
[3], we find that G s Rn is very roughly independent of frequency (except for the best 
junctions at low frequency). This means that there is no reason to suppose that it is 
advantageous to increase j c as frequency squared in the design of high frequency SIS 
mixers. 

Why do our results differ from Ref. [3]? It is likely that this disagreement arises 
because in Ref. [3] the gain was fixed to unity, whereas we find that the mixer gain of an 
optimized SIS receiver falls off roughly as 1/co for bias points on the first photon step 
(Fig. 7). Note in Fig. 7 that we find conversion gain as high as 8 dB in the vicinity of 
100 GHz, but in our solution the mixer is operating far from instability [9] with low noise 
and quite low returned signal and image power. 



CO 

T3 



O 

c 
o 



> 

c 

o 

U 



10r 



Frequency (GHz), for V g = 3 mV 
50 100 500 1000 




0.05 0.1 0.5 1 2 

Normalized Frequency co/co g 



Fig. 7. The IF conversion gain corresponding to the three curves of Fig. 2. 



Third International Symposium on Space Terahertz Technology Page 547 



Nevertheless, we agree with Ref. [3] that the "uRnC = 4 rule" should be modified 
for submillimeter SIS mixers. Harmonic conversion effects should become much less 
important as the frequency is increased, because the SIS junction presents a much weaker 
nonlinearity for harmonic frequencies above co g , especially so for frequencies above 2co g . 
This implies that the beneficial effects of the capacitance are reduced as the frequency is 
increased, and smaller values of ©RnC can be tolerated. Since it is more difficult to re- 
sonate the capacitance at high frequency, smaller values of coR^C are desirable. 
However, small area and high critical current SIS junctions are difficult to fabricate, and 
usually entail undesirable consequences such as inferior junction quality, poorer yield, 
etc. Therefore, the choice of wRnC for submillimeter SIS mixers will at best be an 
informed compromise. 



Acknowledgment: Parts of this work were performed under funding from the Air Force 
Office of Scientific Research and from the National Science Foundation. 



[1] J.R. Tucker and M.J. Feldman, "Quantum detection at millimeter wavelengths," Rev. 
Mod. Phys., vol. 57, pp. 1055-1 113, Oct. 1985. 

[2] J.R. Tucker, "Quantum limited detection in tunnel junction mixers," IEEE J. Quantum 
Electron., vol. QE-15, pp. 1234-58, Nov. 1979. 

[3] A.R. Kerr and S.-K. Pan, "Some recent developments in the design of SIS mixers," in 
Proceedings of the First International Symposium on Space Terahertz Technology , 1990, 
pp. 363-376; published in Int. J. Infrared Millimeter Waves, vol. 1 1, 1 169 (1990). 

[4] R. Blundell and D. Winkler, "The superconductor-insulator-superconductor mixer receiver 
- a review," in NATO Applied Research Workshop on Superconducting Electronics and 
2nd Workshop on Josephson Devices, 1990, Capri, Italy. 

[5] M.J. Feldman and S. Rudner, "Mixing with SIS arrays," Reviews of Infrared & Millimeter 
Waves, edited by K.J. Button (Plenum, New York), vol.1, pp.47-75, 1983. 

[6] M.J. Feldman, "Theoretical considerations for THz SIS mixers," Int. J. Infrared 
Millimeter Waves, vol. 8, pp. 1287-1292, Oct. 1987. 

[7] S. Withington and E.L. Kollberg, "Spectral-domain analysis of harmonic effects in 
superconducting quasiparticle mixers," IEEE Trans. Microwave Theory Tech., vol. MTT- 
37, pp. 231-238, Jan. 1989. 

[8] C.A. Mears, Qing Hu, and P.L. Richards, "The effect of the quantum susceptance on the 
gain of superconducting quasiparticle mixers," IEEE Trans. Magnetics, vol. MAG-27, pp. 
3384-3387, March 1991. 

[9] Qing Ke and M.J. Feldman, "Reflected power effects and high gain in the quantum theory 
of mixing," submitted to IEEE Trans. Microwave Theory Tech. 

[10] M.J. Wengler, "Submillimeter wave detection with superconducting tunnel diodes," 
submitted to Proc. IEEE; see also M.J. Wengler and D.P. Woody, "Optimizing double- 
sideband SIS quasiparticle mixers," IEEE Trans. Magnetics, vol. MAG-27, pp. 3388- 
3390, March 1991. 



Page 548 Third International Symposium on Space Terahertz Technology 



/Losbo N93-27772 



V 



RESONANT TUNNELING DIODES AS 

SOURCES FOR MILLIMETER AND 

SUBMILLIMETER WAVELENGTHS 

O. Vanbesien, R. Bouregba, P. Mounaix 
and D. Lippens 

Centre Hyperfrequences et Semiconducteurs U.A. CNRS N° 287 
Universite de Lille - 59655 Villeneuve d'Ascq Cedex - France 

L. Palmateer + , J.C. Pernot, G. Beaudin* 
and P. Encrenaz 

Ecole Normale Superieure 24, rue Lhomond 75231 PARIS 
* Observatoire de Meudon 92195 Meudon Principal Cedex 

E. Bockenhoff + + , J. Nagle, P. Bois, F. Che voir 

and B. Vinter 

Laboratoire Central de Recherche Thomson CSF 
Domaine de Corbeville 91404 Orsay Cedex - France. 

ABSTRACT 

High-quality Resonant Tunneling Diodes have been fabricated 
and tested as sources for millimeter and submillimeter 
wavelengths. The devices have shown excellent I-V characteristics 
with peak-to-valley current ratios as high as 6:1 and current 
densities in the range of 50-150 kA/cm2 at 300 K. Used as local 
oscillators, the diodes are capable of state of the art output power 
delivered by AlGaAs-based tunneling devices. As harmonic 
multipliers, a frequency of 320 GHz has been achieved by 
quintupling the fundamental oscillation of a klystron source. 



+ Now at IBM, Yorktown Heights + + at Mercedes Benz, Stuttgart 



Third International Symposium on Space Terahertz Technology Page 549 



1. INTRODUCTION 

Resonant Tunneling Diodes (RTD's) exhibit very strong non linearity with 
short time response which make them attractive in non linear applications for 
millimeter and submillimeter wavelengths [1]. RTD's have already demonstrated 
their potential for a variety of high speed/high frequency applications [2]-[5]. In 
this paper we report on the effort of a group of laboratories in France on these 
novel devices with special emphasis on local oscillators and harmonic multipliers. 
The fabrication procedures in a whisker contacted technology and in a microwave 
compatible technology suitable for monolithic integration are outlined in section 
2. The DC and AC characterizations are reported in section 3 whereas the 
oscillator and multiplier results using the devices are described in section 4. 



2. TECHNOLOGICAL PROCESS 

The two types of epitaxial structures grown by molecular beam epitaxy are 
given in Figure 1(a) and (b). Both samples noted A and B had 17 A - thick AlAs 
barriers and access regions with a stepped doping profile from l-2xl0 1 '7 cm-3 to 2- 
3x1018 cm-3. They differ mainly owing to the strained Gao.9Ino.1As layers so that 
structure B resembles a triple well resonant tunneling structure. By placing a 
GalnAs well just prior the growth of the double barrier heterostructure (DBH) it 
is expected that the peak-to-valley current ratio (PVCR's) should be enhanced 
because the negative differential resistance effect involves the anticrossing of two 
confined states [6] [7]. In addition by placing a GalnAs well rather than a GaAs 
one and by a proper choice of the well width (Lw = 40 A), the ground state can be 
lowered in energy while keeping the excited state practically unchanged. The 
associated benefits are a reduction of the peak voltage and higher PVCR's [8]. 
Also note that structure A and B are grown on n+ and S-I substrates respectively. 



Page 550 



Third International Symposium on Space Terahertz Technology 



GaAs 


2 10 13 cm -3 


490 nm 


GaAs 


2 10 17 cm- 3 


50 nm 


GaAs 


undoDed (UD} 


5 nm 


AlAs 


UD 


1.7 nm 


GaAs 


UD 


4.5 nm 


AlAs 


UD 


1.7 nm 


GaAs 


UD 


5 nm 


GaAs 


2 10 17 cm- 3 


50 nm 


GaAs 


2 10 18 cm -3 


500 nm 



(a) 



n + substrate 



GaAs 


3 10 18 cm" 3 


500 nm 


GaAs 


10 17 cm- 3 


10 nm 


GaAs 


undoped (UD) 


5 nm 


In i Gag As 


UD 


5 nm 


GaAs 


UD 


0.5 nm 


AlAs 


UD 


1.7 nm 


GaAs 


UD 


0.5 nm 


IniGa 9 As 


UD 


4 nm 



Symmetrical layers 



(b) 



Figure 1 .Growth sequence for the epilayer on n + substrate (sample A), (a) 
Sample B grown on semi -insulating subtrate. (b) 



The epilayers on n + substrate were processed using a whisker contacted 
technology including patterning of Ni/GeAu layers into matrix of 3.5 um diameter 
dots on the epitaxial side of the wafer and uniform deposition on the back side 
followed by alloying of these layers to make ohmic contacts. Mesa isolation was 
performed by chlorine ion beam assisted etching as shown in figure 2, using the 
patterned metal as a mask. As last stages some of the samples were thinned to a 
thickness of about 120 urn and polyimide was used to surround the diodes in order 
to aid whisker contact. 



Third International Symposium on Space Terahertz Technology 



Page 551 





Figure 2 : SEM Photos of diodes formed by RIE. 



For the epilayers on S-I substrate, the diodes were fabricated in a 
microwave-compatible two-step mesa technology [9]. In that case, the devices 
were connected to low-loss transmission lines in such a way that they can be 
characterized at the wafer level. Such vertically integrated devices require a 
means of connecting the contact on the top of the mesa to the pad of the 
transmission line. We thus developed two versions : (i) a dielectric assisted cross- 
over and (ii) an air bridge interconnection. A scanning electron micrograph of two 
representative devices are shown in figures 3 and 4. In figure 3 a coplanar probe 
configuration is apparent. Also clearly shown is the deposited strap which crosses 
over the mesa edges covered with Si3 N4 layer appearing in dark. In the second 
Version, the dielectric cross-over is replaced by an air bridge yielding a reduced 
parasitic capacitance. Figure 4 illustrates the technology employed with the 



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Third International Symposium on Space Terahertz Technology 



mushroom shaped metallization which enables one to connect small anode 
fingers. 



AuGeNi Ohmic Contacts 
Overlay Metallization 




Figure 3 :Schematic cross section and SEM photo of the RTD fabricated in a 
planar technology. 




Figure 4 : SEM of a diode fabricated using air bridge techniques. 



Third International Symposium on Space Terahertz Technology 



Page 553 



3. DC AND AC CHARACTERIZATION 

Figure 5a shows a typical current voltage I-V characteristic for a GaAs/AlAs 
device on n+ substrate at 300K. The device exhibits a peak current of ~ 16 mA at 
~ 1.8 V which corresponds to a peak current density of— 160 kA/cm2 for a 3,5 um 
diameter diode. For larger size of the diodes, heating of the samples prevents us 
from achieving these densities. A typical DC characteristic for a device on SI 
substrate is displayed in Figure 5b. The device exhibits excellent characteristics 
with PVCR's as high as 6:1 along with simultaneously peak current density of 
50 kA/cm2 which compare favorably to the best published results [10] [11] Note 
also the high degree of symmetry in the I-V curve which is a good indicator of 
quality interfaces. 



201- 
16 
< 12 
E 8 



§ 
1-4 
u -8 



-12 
-16 
-20 



T-300K 


A 




." 



-2-1 1 

Voltage V 



8 


T-300K 


i 


6 




k i 


< 4 


- 


\ / 


E 




/ / 


2 




/ -L/ 







/ 


,. /-, / 




c -2 


■ r\ / 




^ 


/ / 




w 


/ / 




3-4 


' V 




-6 


V 




-8 




b 

_l L_ 1 1 



-U5 -.8 .8 
Voltage V 



1J6 



Figure 5 .Typical DC characteristics for a GaAs/AlAs device on n+ substrate (a) 
and for a GalnAs/GaAs/AlAs pseudomorphic device on SI substrate (b). 



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Third International Symposium on Space Terahertz Technology 



Following our previous work [12], on-wafer reflection gain measurements 
were performed between 50 MHz and 40 GHz using cascade RF probes and an 
85107 A HP network analyser. Shown in Figure 6 is the one port measurement of 
a vertically integrated sample. The active area is 20 um2. The diode is biased in 
the NDR region. Note that ho de-embedding was used at this stage to correct for 
parasitics. For frequency evaluation, we used the equivalent circuit which 
consists of a single capacitor Cd with a parallel negative resistance Rd- -hese 
intrinsic lumped elements are completed by the parasitic capacitance C p , the 
inductance Lp attributable to the bonding and Rs the overall series resistance. A 
good fit was obtained for Cd = 36 fF, Rd = - 172 Q, R s = 9 Q, L p = 60 pH and 
C p = 13 fF (air-bridge technology). With this set of data derived from experiment, 
the cut off frequency for NDR is in excess of 100 GHz. This frequency is limited by 
the high impedance level needed to satisfy the stability criteria. 



$ 



c 

A 



su z 

REF 2.5 Units 

900.0 iriJnita/ 
30.09 Q -32.191 Q 

V 

MARKER 
40.0, 




START 0.050000000 GHz 
STOP 40.000000000 GHz 



Figuer 6 .One port measurement of the impedance. The bias is adjusted in the NDR 
region to satisfy the stability criteria. 



Third International Symposium on Space Terahertz Technology 



Page 555 



4. OSCILLATOR AND MULTIPLIER RESULTS 

The wafers on n+ substrate were sawed into chips of 100x100 um2 and 
mounted in a test waveguide for measuring the oscillator power at 35 and 110 
GHz. The power levels measured at 300 K with a bolometer were 36 pW at 38.6 
GHz and 12 pW at 110 GHz. Referring to the oscillator results from the published 
literature on AlGaAs based RTD's [13] given in figure 7 the output power are 
state of the art results. 



100- 



\ 



10 






EXPERIMENT 



1s 



0.1 




E ■ Sollneretal 
- A our work 

j i ' mm ' i ' Him 



1 



10 100 

Frequency GHz 



mill 



1000 



Figure 7 : Experimental powers for a 4 pm diameter diode from reference [13] and 
results from the present work for a 3.5 pm diameter diode. 

For harmonic multiplication, the samples were mounted in a commercially 
available multiplier mount. The measurement set up has a quasi-optical scheme 
and was initially developed to study reactive species of astrophysical interest [14]. 
The diodes were driven by a klystron at 64 GHz and, in the output path, high 
filters enables one to spectrally analyze the power delivered by the diode. The 
receiver is a helium cooled InSb detector. Figure 8 shows the power response at 
3rd harmonic (192 GHz) and 5th harmonic (320 GHz). For comparison in terms of 
available power commercial Schottky diodes were also tested under the same 



__ 



Page 556 



Third International Symposium on Space Terahertz Technology 



experimental conditions. It is interesting to note that equivalent performances 
were obtained for both types of devices by increasing the multiplication order to 
frequency quintupling. 



£ 
a 

UJ 

r- 
O 
UJ 

fc3 

Q 
IU 
O 

< 

r- 

o 

> 



500 T 



400 - 



300 - 



200 - 



100 - 



SCHOTTKY fifth hirm. 



OBRTD third harm. 



♦ 
♦ 

♦ 



D8RTD fifth harm. 



••;/ 



-+- 



±^ 



▲ 
▲ 



16 14 12 10 8 6 4 2 
POWER SOURCE ATTENUATION (dB) 



Figure 8 : Measured voltage of InSb detector against input power delivered by a 
klystron at 64 GHz. 



Third International Symposium on Space Terahertz Technology Page 557 



In the multiplier experiment the devices were unbiased and driven in the 
NDR region to take advantage of multiple extrema in the current waveform. This 
requirement can be unfavorable especially for high threshold voltage devices 
when input power is limited [14]. From this viewpoint, it is clear that 
pseudomorphic structures with a buried well may overcome partly this difficulty. 
From Figure 3 it is apparent that a drastic decrease in the peak voltage 0.8 V 
instead of 1.8 V has been achieved by comparing structures A and B. In addition it 
can be noted that the PVCR's were enhanced. This suggests the superiority of 
these new tunneling devices for multiplication in view of the large harmonic 
content in the current and of the reduction of the amount of input power required 
to pump the diode. 



5. CONCLUSION 

High performance resonant tunneling diodes were sucessfully fabricated in 
a whisker contacted and in a planar technology. The RF capabilities of the diodes 
were demonstrated either by direct measurement of their small-signal impedance 
or by using them for oscillator and multiplier. 



ACKNOWLEDGEMENTS 

This work was supported by the Ministere de la Recherche et de la 
Technologic Technical assistance of M. Bogey, J.L. Destombes and A. Lescluse 
with the Laboratoire de Spectroscopic Hertzienne of the Lille University is highly 
appreciated. 



Page 558 Third International Symposium on Space Terahertz Technology 



REFERENCES 

[1] T.C.L.G- Sollner, E.R. Brown and H.Q. Le, "Microwave and Millemeter- 
wave Resonant Tunneling Devices". Physics of Quantum Electron Devices 
edited by F. Capasso Springer- Verlag. 

[2] E.R. Brown, J.R. Soderstorm, CD. Parker, L.J. Mahoney, K.M. Kolvar and 
T.C. Mc uill, "Oscillations up to 712 GHz in InAs/AlSb resonant tunneling 
diodes". Appl. Phys. Lett. 58, pp. 2291-2293, May 1991. 

[3] A. Rydberg and H. Gronquist "Quantum well high efficiency millimeter- 
wave frequency tripler". Electronics Letters, Vol. 25, pp. 348-349, 1989. 

[4] P.D. Batelaan and M.A. Frerking, "A quantum well frequency multiplier 
with millimeter wave output". Proc. 4th Conf. Infrared Physics Zurich, 
pp. 527-529, 1988. 

[5] R. Bouregba, D. Lippens, L. Palmateer, E. Bockenhoff, M. Bogey, J.L. 
Destombes and A. Lecluse, "Frequency multiplication using resonant 
tunneling diode with output at submillimeter wavelengths". Electronics 
Lett. Vol.26, pp. 1804-1905, October 1990. 

[6] D. Thomas, F. Chevoir, E. Barbier, Y. Guldner and J.P. Vieren, "Magneto 
tunneling of charge build up in double barrier diodes". Proc. of 4th 
International Conference on Superlattices Microstructures and 
Microdevices 5, pp. 219-224, 1989. . 

[7] P. Mounaix, O. Vanbesien and D. Lippens, "Effect of cathode spacer layer on 
the current-voltage characteristics of resonant tunneling diodes". Appl. 
Phys. Lett. 57, pp 1517-1519, October 1990. 



Third International Symposium on Space Terahertz Technology Page 559 



[8] T.P.E. Broeckaert, W. Lee and C.G. Fonstad "Pseudomorphic 
Ino.53Gao.47As/AlAs/InAs resonant tunneling diodes with peak-to-valley 
current ratios of 30 at room temperature. 

[9] D. Lippens, E. Barbier and P. Mounaix, "Fabrication of High-Performance 
Al x Gai- x As/In y Gai-yAs/GaAs Resonant Tunneling Diodes using a 
Microwave-compatible Technology. IEEE Electron Dev. Lett. Vol. 12, pp. 
114-116, March 1991. 

[10] R.M. Kapre, A. Madhukar, and S. Guha "Highly strained 
GaAs/InGaAs/AlAs resonant tunneling diodes with simultaneously high 
peak current densities and peak-to-valley current ratios". Appl. Phys. Lett. 
58, pp. 2255-2257, May 1991. 

[11] H. Brugger, U. Meiners, C. Wolk, R. Deufel, A. Morten, M. Rossmanith, 
K.V. Klitzing and R. Sauer "Pseudomorphic Two Dimensional Electron-Gas- 
Emitter Resonant Tunneling Devices" Microelectronics Engineering 
Elsevier, 15, pp. 663-666, 1991. 

[12] P. Mounaix, P. Bedu, D. Lippens and E. Barbier "Measurement of negative 
differential conductance up to 40 GHz for vertically integrated resonant 
tunneling diodes". Electronics Letters Vol. 27, pp. 1358-1359, July 1991. 

[13] E.R. Brown, T.C.L.G. Sollner, C.D. Parker, W.D. Goodhue and C. Chen. 
"Oscillations up to 420 GHz in GaAs/AlAs resonant tunneling diodes" Appl. 
Phys. Lett. 55, pp. 1777-1779, October 1989. 

[14] J.L. Destombes, C. Demuynck and M. Bogey "Millimeter-wave and 
submillimeter-wave spectroscopy of molecular ions". Phil. Trans. R. Soc. A 
324, pp. 147-162,1988. 



Page 560 Third International Symposium on Space Terahertz Technology 

/£*&/ /1 J N93-27773 



- \? 



Simulation of Electron Transport 

in 
Quantum Well Devices 



D. R. Miller, K. K. GuIIapalli, V. R. Reddy, and D. P. Neikirk 

Department of Electrical and Computer Engineering 
Microelectronics Research Center 
The University of Texas at Austin 



1.0 Introduction 

Double barrier resonant tunneling diodes (DBRTD) have received much attention as 
possible terahertz devices. Experimentally, DBRTD 's have shown detection capabilities at 
sub-millimeter wavelengths 1 . When used as oscillators, small amounts of power have also 
been measured in sub-millimeter range 2 . Despite these impressive experimental results, the 
specific of the device physics (i.e., how the electrons propagate through the structure) are 
only qualitatively understood. Therefore, better transport models are warranted if this 
technology is to mature. 

Near the heterostructure double barrier region, it is generally accepted that quantum 
mechanical transport, via tunneling and reflections, dominate the electron dynamics. 
However, most DBRTDs in use today are designed with extended spacer regions. These 
spacer regions serve the function of increasing the real part of the overall device impedance 
while simultaneously reducing the imaginary part, thereby incorporating DBRTDs in 
millimeter wave circuits a far easier task. Since the spacer regions are sufficiendy removed 
from the heterojunctions, electron propagation should be govern by semiclassical and not 
quantum mechanical considerations. Here, semiclassical refers to transport which is 
adequately describe using some form of the semiclassical Boltzmann equation. 



Third International Symposium on Space Terahertz Technology Page 561 



Past simulation models of DBRTD structures have evolved from simple 
Schrodinger equation solutions of a free electron in a double barrier potential to more 
complicated methods involving multiband, multivalley Schrodinger solutions 3 ' 4 or single 
valley kinetic equations that utilizes quantum Wigner functions 5,6 ' 7 . These methods are 
expected to work reasonably well for ideal DBRTD structures with parabolic bands in 
which only quantum mechanical reflections and tunneling are important. However, 
because the Schrodinger or single valley Wigner models do not include realistic phonon 
scattering or band structure effects (i.e., multiple valleys, rton-parabolicity, multiple bands, 
etc.) these models do not adequately address the transport through the semiclassical region. 
Alternatively, the semiclassical Boltzmann equation provides an adequate description of the 
semiclassical region, but fails completely near the DBRTD heterostructure region. 

To model a DBRTD structure with two distinct transport regions, two options are 
available. The first option incorporates a composite scheme by which each region is 
modeled with an equation suitable for that region. The two solutions are then matched at a 
quantum / classical interface to obtain a self consistent solution throughout the device. We 
find combinations of a free particle Schrodinger equation for the quantum region coupled 
with either the drift/diffusion 8 or Monte Carlo 9 formalism for the semiclassical region quite 
useful. However, agreement between experiment and theory is still lacking since the 
simple Schrodinger equation is only an approximate solution of the electron transport 
within the heterostructure region. 

The second option is to model the entire device with one self consistent formalism 
that, in principle, can account for all the important device physics for each region. The 
kinetic equation based upon the Lattice Wigner function is a promising candidate for such a 
task. In this paper, we will use the Lattice Wigner function to explain important transport 
issues associated with DBRTD device behavior. 



2.0 The Lattice Wigner Function 

The lattice Wigner function 10 - 11 we employ is based on the discrete spectrum 
composed of Wannier and Bloch crystal representations, making it different from other 
Wigner function methods. Because of the choice of representations, band structure effects 
are explicitly included in the kinetic equation. Thus, issues such as T to X tunneling, non- 
parabolicity of the conduction bands, or effective mass variations across the heterojunction, 



Page 562 



Third International Symposium on Space Terahertz Technology 



can be examined in detail. Phonon scattering is also included through the standard 
semiclassical Boltzmann collision term. 

In a multiple barrier heterostructure device with multiple non-parabolic conduction 
band valleys and no interband coupling, it is possible to write separate, but coupled, kinetic 
equations for each valley. The Wigner function for the i* valley, f { , is found from the 

solution of 

_ 9f i w (R,k) [ I 3e.(k) Bf^flU) eE Bf^R.k) r afflU) ^ 
3t h 3k 3R h 9k I, 3t J, 



Col 



#Valley»#Bamers 

-X £ Br+Br+B^ 

j=l m=l 



(1) 



where the barrier scattering matrices, Bo, Bi, and B2 are given by 

Bl 2 £ 3R [* h f [V^y** 2 j + V^^K 2 






3R 2 



i Ie ^- k ,r V] 



VN L R 



'Pulie(ni)| ** + ~ j Ypulie(m)l** ~ 



(2) 
(3) 
(4) 



8im A k . 

The indices, j and q, indicate the j th conduction band valley and the q th barrier. The 
coefficients, B^M, are the potential energy terms that account for one heterostructure 
barrier. In equations 1 through 4, n is the band index, k and k ' are crystal momenta, R 
and R' are lattice vectors, Nl is the number of lattice sites, and E is the electric field. 
V'uise is the potential energy diagonal matrix element of the mft 1 pulse function that localizes 

u *i *i 

the barrier. The change in the i" 1 valley effective mass (m A ), velocity (v A ), and the offset 
energy (AEO are given by 



m 



A "^Barrier 

Barrier / 



m 



Bulk 



i _ 1 ae^Ck) 1 ae^Ck) 



V A = 



dk 



3k 



AE i (k) = e^ raier (k)-ef ulk (k) 



(5) 

(6) 
(7) 



Third International Symposium on Space Terahertz Technology Page 563 



where £^ amer (ki) and e n u (kj) are the minimum conduction band energies of the i* valley 

for the barrier and bulk materials, respectively. 

As noted earlier, equation 1 accounts for both effective mass variations across the 
heterostructure interface and intervalley coupling. The intervalley coupling is possible 

since the i m valley distribution function f { is coupled to the j 01 valley distribution function 

w 
f through the barrier terms. Effective mass variation effects are due to the crystal 

momentum dependence of the barrier, and result in the spatial derivatives of the distribution 
function in the B^ q and Bj q terms. Mass variations are also responsible for raising and/or 

lowering the barrier height, as seen in the expression for Bj q . This is because the energy 

band of the bulk material may rise at a different rate than the energy band of the barrier 
material for a given momentum change. For semiconductors with non-parabolic valleys, 
additional barrier scattering terms arise due to the higher derivatives associated with the 
Wigner-Moyal expansion. However, in this paper we will assume that the effective masses 
and the non-parabolicity factors are approximately the same for the bulk and barrier 
material. Therefore, Bi, B2, and all other higher barrier scattering terms will be set to zero. 
Including some sort of phonon collision processes into a Wigner function 
calculation of an DBRTD is not new. Most calculations have approximated the influence 
on carrier transport from phonon collisions using the relaxation time approximation 7 - 12 ' 13 . 
In addition, these calculations have restricted phonon scattering events to one valley. 
However, because of the high electric fields within the DBRTD structure, intervalley 
scattering is required. Thus, we will assume that the total scattering matrix, Sjotal. is given 
by 

S^(k\k) = S^ p (k , ,k) + SiL Mtic (k I ,k) + Si,L vllUey (k , ,k) (9) 

where Spop, S Acoustic, and Simervalley refer to the polar optical, acoustical, and intervalley 
phonon scattering matrices, respectively. Assuming non-degenerate statistics, we can write 
the collision term in equation 1 for the i* valley Wigner function as 

fa&Bdo] = 1 1 j J ^ L(k)k . )f w (R)kl) _ s ^ (k . >k) ^ (R)k J 

V tit J Col W Lpl [ k' J (g) 

where Ny is the total number of valleys. The material parameters as well as the functional 
form for each scattering matrix are identical to those used in Monte Carlo device 
simulations and are given in reference 14. 



Page 564 Third International Symposium on Space Terahertz Technology 



The Wigner distribution function for each valley is defined in the three dimensional 
crystal momentum space. If the structure and the electric field are homogeneous in two of 
the spatial variables (independent of R y and R z ), equation 1 reduces to a four dimensional 
integral/differential equation. Since solutions to the four dimensional problems are 
exceedingly difficult, further simplification is required. We do this by characterizing the 
distribution function that is transverse to the electric fields by a Maxwellian defined at some 
transverse temperature Tt. Furthermore Tt is assumed to vary with longitudinal position. 
Thus, for electric fields in the (100) direction with quantum well barriers grown in the 
(100) plane, the total distribution function is approximated by the product of a transverse 
and a longitudinal distribution functions 

f i w (R x ,k x ,k y ,k z ) = f i R "(R x ,k x )exp(-E;/k b T t (Rj) (9) 

where k x , k y , and k z are defined with respect to the valley minimum. The transverse 
energy for non-parabolic valleys , Et, is assumed to be of the form 

r"H r" 

2m' 2m, 1 

e; = t — \ y ; do) 




where aj is the non-parabolicity factor for the i th valley. Note that although the 
longitudinal and transverse distribution functions are not coupled through the electric field 
or the barrier scattering matrix, they are coupled by the phonon scattering matrix. It is 
because of the this coupling that the transverse temperature significantly impacts the device 
physics, as discussed in the next section. 

3.0 Simulation Results 

The first obvious question that can be addressed with the lattice Wigner function is 
how close to the quantum well an electron is when quantum mechanical reflections and/or 
tunneling affect the electron's behavior. One measure of this distance is obtained by 
comparing the size of the barrier matrix (Bo) with the size of the phonon matrix (STotal) at 
each lattice site. However, such a direct comparison may provide an overestimate of the 
extent of the quantum region. During one phonon scattering interval, an electron may 



Third International Symposium on Space Terahertz Technology 



Page 565 



propagate through many lattice sites due to either a high electric field or a high initial 
velocity. Therefore, the effective barrier scattering strength is an average over all lattice 
sites occupied during the mean free flight time between collisions. In other words, the 
barriers' influence on the electrons is best determined by the matrix 



-I (Dlfflol 

b Tou1 =-— x f ' Br(R x (t),k x (t))dt 

: i ~ r o j=i ° 



(11) 



integrated over the classical trajectory. In equation 1 1, Bo is the matrix defined by equation 
2, and R x (t) and k x (t) are given by the semiclassical equations of motion. 




l/Tn^OOkV/cm) 



l/xj;0kV/cm) 

1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 ■ ■ 1 1 1 1 1 1 1 1 1 1 1 
M 2( 4( 6C 8( 10( li( 14( 
Position (monolayer! 

Figure 1. Effective quantum well scattering strength versus 
position from the center of a 6/19/6 GaAs/AlAs quantum 
well. 



It is difficult to evaluate the above equation for the general case. However, Figure 
1 shows the effective quantum well scattering strengths determined by taking the matrix 
norm of Bjotal for two simplified conditions. The structure is a 6/19/6 GaAs/AlAs 
quantum well. The first solid line assumes the case Of an extremely slow electron in which 
the electron transverses only one lattice site between phonon collisions. The second line 
assumes that the mean free path was 32 monolayers, corresponding to a distance of 90.6 
A. In both cases the electric field was set to zero, which significantly simplified the 
integration along the classical trajectory. Also plotted are the momentum relaxation times 
corresponding to different electric fields. These values represent the size of the phonon 



Page 566 Third International Symposium on Space Terahertz Technology 



scattering matrix (Sjotal) determined using standard Monte Carlo simulations of 
homogeneous material 9 . Thus, when comparing the relative strengths of the quantum well 
and the phonon scattering matrices, one should compare the fast electron case with the high 
field momentum relaxation time since these two conditions usually exist together. 
Similarly, the slow electron case corresponds to the zero electric field relaxation time. 

Two points are worth emphasizing from Figure 1. The first is that slow electrons 
are influenced by the quantum well with greater strength and at greater distances when 
compared to fast electrons. The second point, however, is the one which we want to 
emphasize. DBRTD structures are usually operated at extremely high electric fields (200 
kV/cm) and under ballistic conditions. Therefore, as can be seen from the fast electron 
curve of Figure 1, the region where the barrier potential has the greatest impact on electron 
dynamics is within 65 monolayers of the center of the well. Beyond that region, phonon 
scattering is more important. This fact supports the hypothesis that much of the DBRTD is 
governed by semiclassical considerations for devices that extend 400 to 800 monolayers 
from the quantum well. 

The Wigner function has been used extensively to account for the quantum 
transport within the heterostructure region of a DBRTD device. However, the question 
arises as too its ability to properly account for electron transport within the semiclassical 
region. To show that this is possible, we have used the Lattice Wigner Function to 
calculate the velocity versus electric field characteristics for homogeneous GaAs. For 
homogeneous samples, equation 1 reduces to 

= _^ af^jo_ j_| K | ey ( s ^ (k>k . )f w (R>kl) _ s ^ (k . >k)f w (R>k) j| (12) 

To solve this equation, the first step is to integrate over the transverse crystal momentum 
directions. The resulting one dimensional phonon scattering matrix is then combined with 
the matrix generated by the electric field term to yield the equation 

= T^jJ?-f R '- j - k "» (13) 

where T is the combined phonon/electric field matrix and fj Rx J. kx is the vectorized 
longitudinal distribution function. The valid solution exists when the minimum eigenvalue 
of T is equal to zero 15 . If the minimum eigenvalue is not zero, the electric field and the 
valley transverse temperatures, T t , are adjusted so that homogeneous equation can be 



Third International Symposium on Space Terahertz Technology 



Page 567 



satisfied. In affect, we are using the transverse valley temperatures as fitting parameters to 
achieve a reasonable velocity versus field curve. Once the zero eigenvalue is calculated, the 
eigenvector for that eigenvalue can be determined, from which the velocity is obtained. 
Note that pure state tunneling models using only the Schrodinger equation do not produce 
reasonable velocity-field curves for GaAs. 

The above procedure yields good agreement with the Monte Carlo results, as 
indicated in Figure 2. Here, a velocity field curve generated by equations 12 and 13 is 
compared against a three valley model calculated using standard Monte Carlo techniques. 
Furthermore, the temperatures required to obtain this level of agreement are comparable to 
the transverse temperatures calculated from the homogeneous Monte Carlo simulations. 
However, at low electric fields there is a larger discrepancy between the two alternative 
approaches. This difference is probably due to the fact that the transverse distribution 
function is not precisely Maxwellian. However, this error is fairly small, demonstrating 
that the Lattice Wigner function is capable of simulating both the quantum transport region 
and the classical transport regions of a device. 





4.00e+7 i 




o 


3.50e+7 - 
3.00e+7 ■: 


. Lattice 


b 


2.50e+7 ■: 




<** 


2.00e+7 - 


/ • •^ 


o 

o 


1.50e+7 - 




> 


1.00e+7 ■: 






5.00e+6 " 


/ Monte Carlo 



0.00e+0 



i i i i 



5000 10000 15000 

Electric Field (V/cm) 

Figure 2. Velocity field calculation for a 3 valley spherical 
non-parabolic Wigner function model. Also shown is the 
corresponding result from a Monte Carlo simulation. The 
scattering parameters used for each calculation are identical, 
and are found in reference 14. 



Thus far, we have applied the Lattice Wigner function to determine generic 
properties of bulk materials and of resonant tunneling transport. This formalism's real 



Page 568 



Third International Symposium on Space Terahertz Technology 



usefulness can be demonstrated by applying it to an actual DBRTD device. Shown in 
Figure 3 is a typical GaAs/AlAs DBRTD. The quantum well consists of 6 monolayer 
AlAs barriers separated by 19 monolayer GaAs well. As can be seen in Figure 3, a 
moderately doped, extended spacer region is added to the right hand side of the quantum 
well. As will be shown latter, it is the semiclassical transport through the extended spacer 
layer that can have a serious impact on device behavior. 




t — ' i ' i — « — r— « — r 

-200 -100 100 200 300 400 

Position (monolayers) 

Figure 3. 6/19/6 GaAs/AlAs DBRTD used for the device 
simulations. The x axis origin is defined as the center of the 
quantum well. 



The equilibrium electron concentration, found by integrating the Lattice Wigner 
function over momentum space, is shown in Figure 4. However, under bias conditions, 
obtaining the electron concentration via the Lattice Wigner function is much more 
complicated. The basic problem is in determining the transverse temperature profile of the 
spacer region. We have previously shown using a composite Schrodinger/Monte Carlo 
model 9 , that for typical bias voltages, electrons are quickly scattered into the upper satellite 
values once they emerge from the quantum well. This intervalley scattering between 
equivalent and non-equivalent conduction band valleys quickly heats up the carrier 
distribution functions in the transverse direction. The effect of the carrier heating on the 
charge densities can be considerable. 



Third International Symposium on Space Terahertz Technology 



Page 569 



10 



19. 



10 1S i 



10 



14. 



Electron Density 




'Impurity Density 



-200 -100 100 200 300 400 

Position (monolayers) 

Figure 4. Equilibrium electron concentration for the device 
shown in Figure 3. 



10 



19. 



10 



14 



Non constant T 




T=300 degrees 



-i — | — i — i — i — i ■ i — i— i — r- 
200 -100 100 200 300 400 



Position (monolayers) 

Figure 5. Electron concentrations obtained from the Lattice 
Wigner formalism for two different transverse temperature 
profiles. The dc bias is at 0.7 V. 



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Third International Symposium on Space Terahertz Technology 



Figure 5 illustrates the effect of transverse temperature on the simulated electron 
concentration for this device biased at 0.7 V. The dashed curve shown in this figure was 
generated under the assumption of constant transverse temperature within the spacer 
region. As seen from this curve, the electron density actually decreases past the quantum 
well region, indicating an increase in the overall electron velocities. The increase in the 
electron velocities is a result of a non-physical assumption. Because the transverse 
temperature in this space charge region is kept artificially low, the electron population is 
also artificially cooled, keeping all electrons in the fast T valley. Therefore, unrealistically 
high velocities would be predicted under this constant temperature assumption. This can 
lead to unrealistic high frequency performance predictions, since fast electrons generally 
result in improved frequency response. 

Also shown in Figure 5 are the results of a Lattice Wigner function calculation 
using a transverse temperature profile obtained through the Schrodinger/Monte Carlo 
model. As seen from this figure, the electron concentration is increased significantly over 
the constant temperature model, indicating a slowing down of the electron population 
within the spacer region. The reduced velocity is a result of the increased phonon 
scattering within the T valley, as well as some T to L intervalley transfer. This slowing of 

the electrons can have an significant impact on the device performance, as discussed below. 
8000 




t — ■ — i — ■ — i — ' — r— • — r 
■200 -100 100 200 300 
Position (monolayers) 



400 



Figure 6. Transverse temperature profile for the device 
given in Figure 3. The dc bias is at 2.0 volts. The 
temperature was obtained from a Schrodinger/Monte Carlo 
transport model discussed in reference 9. 



Third International Symposium on Space Terahertz Technology Page 571 



The bias of 0.7 V is well below the voltage at which peak current is expected in this 
device. However, even under these conditions, the Monte Carlo simulated T transverse 

temperature peaks at 2500 Kelvin. At even higher biases, the transverse temperature 
becomes extremely high. Figure 6 shows the transverse temperature profiles extracted 
through a three valley Schrodinger/Monte Carlo model for a device biased at 2.0 V . As 
seen from this figure, the T temperature can exceed 7000 degrees Kelvin. Even the upper 

satellite valleys are exceptionally hot Thus, we would expect that most of the electrons for 
this bias voltage are in the upper satellite valley, which is indeed the case. 

4.0 Impact on High Frequency Behavior 

Much of the discussion concerning the ultimate frequency limitations of DBRTD 
devices has centered on the frequency limitations imposed by the quantum well itself. It 
has been projected that the quantum well is capable of operating at terahertz frequencies 16 . 
However, the preceding discussion illustrates that the semiclassical spacer regions can have 
a dramatic impact on the behavior of DBRTD devices. Because of the efficient phonon 
scattering processes, the actual number of free carriers within the spacer layer is much 
higher than predicted by either a pure state Schrodinger solution, a one valley Wigner 
solution, or a multiple valley constant temperature Lattice Wigner solution. This free 
charge contributes to a positive resistance which is equivalent in every way to the space 
charge resistance found in transit time diodes 17 . The magnitude of the space charge 
resistance is fundamentally determined by the total number of free carriers within the 
region. Therefore, any analysis which does not realistically describe electron densities 
cannot be used to project the high frequency performance of a real device. 

Understanding space charge resistance is important because it is generally felt that 
in order to improve the output power of a DBRTD device, one must dramatically increase 
the current density. In reality, this procedure will be useful only up the point where 
deleterious space charge resistance effects become dominant. This can be seen even under 
static conditions. For example, considered the measured dc-IV curve of an InGaAs/AlAs 
DBRTD structure, shown in Figure 7. The total spacer region for this device was 1250A. 
With available simulation tools, it is possible to model comparable structures with 
artificially increased current densities. Thus, five static simulations where performed using 
a composite quantum injector/drift/diffusion model, as described in reference 18. For each 



Page 572 



Third International Symposium on Space Terahertz Technology 



simulation, the DBRTD spacer regions where kept constant. The only difference between 
each simulation is an assumed increase in the current density of the device, starting with the 
dc-IV curve given in Figure 7. The results of the simulations are shown in Figure 8. As 
the current density is magnified, the presence of the free carriers within the spacer regions 
can cause a major portion of the negative resistance regime to become positive. It is 
obvious that this positive resistance would prevent useful device operation, despite the fact 
that the difference between the peak and valley current is exceptionally high. 



E 
u 

a- 
m 

< 

>. 

°35 

c 
o 

a 

c 
s 



O 




Figure 7. The experimentally measured dc-IV curve for an 
InGaAs/AlAs DBRTD. The structure is similar to Figure 3 
except that the total spacer layer to the right of the quantum 
well is 1250A. 



The current density where space charge resistance becomes important depends on 
two factors. The first is the length of the spacer regions. As the length is decreased, the 
total number of carriers within the region is also reduced. The price paid for this reduction 
is a corresponding decrease in the overall device impedance. Furthermore, it is doubtful 
that the spacer region can be eliminated beyond a certain point, since there is always a 
depletion region formed due to the high electric fields near the quantum well. This is even 
true if the heavily doped contacts are immediately adjacent to the quantum well. 



Third International Symposium on Space Terahertz Technology 



Page 573 



































e" 


2S0-; 














u 














jdftOX 


cr 
















» 


200 - 














< 














MOX 


*-" 
















>. 


150 ■; 














« 












30X 


c 
















9 
















Q 


100 ■; 














** 










20X 




c 
















a> 
















h* 


■ 














3 


50- 






10X 






u 


0-i 


















1 


2 3 




4 




5 6 








Voltage 









Figure 8. The simulated effect of the space charge resistance 
on the dc-/V curve. These curves were generated using the 
Schrodinger/Drift Diffusion composite model in which the 
current given by Figure 7 was scaled upward 10, 20, 30, 
40, and 50 times. 



The second factor dictating when the space charge resistance becomes important is 
determined by the velocity of the carriers. Higher velocities result in lower electron 
concentrations, since total current must be conserved. It is in predicting these 
concentrations and velocities that previous quantum mechanical models have failed, thus 
failing to predict the importance of the spacer regions in overall low and high frequency 
device behavior. Thus, in order to accurately project the ultimate performance of DBRTDs, 
it is critical to use a quantum kinetic formalism such as the Lattice Wigner function. Future 
results using this model should lead to a determination of the behavior of DBRTDs at 
terahertz frequencies. 



This work has been supported by the Texas Advance Research Program. 



Page 574 Third International Symposium on Space Terahertz Technology 



5.0 References 

1 . T. C. L. G. Sollner, W. D. Goodhue, P. E. Tannenwald, C. D. Parker, and D. D. 
Peck, Applied Physics Letters, 43, 588, (1983). 

2. E. R. Brown, J. R. Soderstrom, C. D. Parker, L. J. Mahoney, K. M. Molvar, and 
T. C. McGill, Applied Physics Letters, 58, 2291, (1991). 

3. D. Z. Y. Ting, M. K. Jackson, D. H. Chow, J. R. Soderstrom, D. A. Collins, and 
T. C. McGill, Solid State Electronics, 32, 1513, (1989). 

4. E. T. Yu, M. K. Jackson, and T. C. McGill, Applied Physics Letters, 55, 744, 
(1989). 

5. W. R. Frensley, Physical Review B, 36, 1570, (1987). 

6. K. L. Jensen and F. A. Buot, Journal of Applied Physics, 65, 5248, (1989). 

7. N. C. Kluksdahl, A. M. Kriman, D. K. Ferry, Physical Review B, 39, 7720, 
(1989). 

8. D. R. Miller, V. K. Reddy, R. L. Rogers, and D. P. Neikirk, SPIE Proceedings 
on High-Speed Electronics and Device Scaling, 1288, 167, March 18-19 1990. 

9. K. K. Gullapalli, D. R. Miller, and D. P. Neikirk, 1991 International Electron 
Devices Meeting, Washington D. C, December 8-11, pg. 51 1, 1991. 

10. F. A. Buot, Physical Review A, 33, 2544 (1986). 

11. D. R. Miller and D. P. Neikirk, Applied Physics Letters, 58, 2803, (1991) 

12. K. L. Jensen and F. A. Buot, Journal of Applied Physics, 67, 7602, (1990) 

13. W. Frensley, Proceedings of the International Symposium on Nanostructure 
Physics and Fabrication, College Station, March 13-15, pg. 231, 1989. 

14. C. Jacoboni and P. Lugli, The Monte Carlo Method for Semiconductor Device 
Simulation . Springer- Verlag Wien, New York, 1989. 

15. J. P. Aubert, J. C. Vaissiere, and J. P. Nougier, Journal of Applied Physics, 56, 
1128,(1984) 

16. W. R. Frensley, Applied Physics Letters, 51, 448, 1987. 

17. S. M. Sze, Physics of Semiconductor Devices . 2nd Edition, Chapter 10, John 
Wiley, New York, 1981. 

18. D. R. Miller, V. P. Kesan, R. L Rogers, C. M. Maziar, D. P. Neikirk, The 13th 
International Conference on Infrared and Millimeter Wave, 1988. 

19 E. Wigner, Physical Review, 40, 749, (1932) 



/' 



Third International Symposium on Space Terahertz Technology Page 575 



Parallel Arrays of Josephson Junctions for ^f£>-33 
Submillimeter Local Oscillators i, ~ ,- 

Aleksandar Pance* and Michael J. Wengler p ^ 

Department of Electrical Engineering INJQQ 9 *y ^ ry A 

University of Rochester M & tJ m 7L i i t rt 

Rochester, NY 14627 

Abstract 

In this paper we discuss the influence of the DC biasing circuit on operation of parallel 
biased quasioptical Josephson junction oscillator arrays. Because of nonuniform distribution of 
the DC biasing current along the length of the bias lines, there is a nonuniform distribution of 
magnetic flux in superconducting loops connecting every two junctions of the array. These DC 
self-field effects determine the states of the array. We present analysis and time-domain numerical 
simulations of these states for four biasing configurations. We find conditions for the in-phase 
states with maximum power output. We compare arrays with small and large inductances and 
determine the low inductance limit for nearly-in-phase array operation. We show how arrays can 
be steered in H-plane using the externally applied DC magnetic field. 

~^""~-—-~- Introduction 

The Josephson junction is a natural choice for submillimeter local oscillator since it is a 
"voltage controlled oscillator" with typical voltage scales of mV and an oscillation frequency fj = 
483 GHz per mV of dc bias. The existence of Josephson radiation into the terahertz range has 
been demonstrated at Cornell [1]. A major disadvantage of the Josephson junction is its very low 
output power. With DC voltage bias of 1 mV at 483 GHz, a junction which could accept 100 ^A 
will put out less than 100 nW of RF power. Therefore, practical local oscillators must use arrays 
of many junctions oscillating in phase. Submillimeter Josephson oscillator arrays with usable 
power levels have been made at Stony Brook [2] and NIST [3]. 

We have proposed to build a large 2-D active grid array of parallel biased Josephson 
junctions [4]. In our design, every junction drives a single antenna and the power from the whole 
array is quasioptically combined. By biasing all junctions in parallel, we assure that all of them 



* 1991 Link Energy Foundation Fellow. 



Page 576 Third International Symposium on Space Terahertz Technology 

radiate at exactly the same frequency. For maximum radiated power, all junctions must also be in 
phase. 

The DC biasing circuit of the 2-D quasioptical Josephson array plays a very important role 
in phase-locking of Josephson junctions. In a two-dimensional array the DC biasing current is 
supplied at the ends (Fig. 1). Because of that, the DC current is nonuniformly distributed along 
the length of the biasing line. This current induces the nonuniform DC magnetic flux in 
superconducting loops between every two neighboring junctions. Because of the superconducting 
quantum interference effects [5], these self-induced fluxes determine the phase differences between 
the neighboring junctions, and therefore the states of the array. These effects will be referred to as 
the DC self-field effects. It is clear that, depending on the particular bias circuit, the in-phase state 
can only be a special, rather than common state of the parallel 2-D Josephson junction array. 

If the rows of the 2-D parallel Josephson array are biased independent from each other, the 
DC self-field effects are, to the first order, limited to each row, and the whole 2-D array can be 
looked at as a collection of 1-D parallel arrays. We will therefore investigate these DC self -field 
effects in linear parallel arrays. 

N-junction linear parallel array 

The most general biasing scheme for the linear parallel array is presented in Fig. 2. We use 
the RSJC model of the Josephson junction that consists of ideal Josephson junction, shunt 
resistance and parasitic capacitance (Fig. 2). The ideal Josephson junction is described by 
relationships between its current I, voltage V and phase difference <J) of the superconducting 
quantum mechanical wave function between two sides of the junction 

dd> 2e 
I = Ic sin(<{)), f = f V 

where Ic is the junction critical current. Assuming that all junctions are identical, the circuit from 
Fig. 2 can be described with the following system of equations: 



Third International Symposium on Space Terahertz Technology Page 577 



ii(t) = ^<!>2(*H>i(t))+^-- — - y <Pex 

ij(x) = f(<t> j+ ,(t)-2 ♦ J <tH> j . 1 (t)) + i , j = (2, N-l) 
A. 2 

W = - t<^H»n-iW) + v + T + T 1 *« 

A. 2 2 A, 



(La) 



where <()j is the superconducting wave function phase difference across the j* junction, ij is the 
total current through the j* junction, Yj, yl and Yr are biasing currents and <p e x is the normalized 
externally applied DC magnetic flux, 

ij(T) = p<t)(T) + <|>(t)+sin(<|)(x)) 
Yj=iuj + iDj» 

"/L =1 Lin + ^out ' YR =1 Rin + ^R out 






(Lb) 



with capacitance and inductance parameters P and X, respectively, given by 



.2 



RT , L , ^o ^ h 






*o= 



■j 2tiL 2e 



c 



(Lc) 



where Lj is the zero-bias Josephson junction inductance and Oq = 2.07 X 10* 15 Wb is the flux 
quantum. In equations (La-c), time is given in units of t£, all currents in units of lc and 

normalized junction voltages, that are just time derivatives of junction phases <t>j, in units of IcR. 

In the case of 2-junction array, the in-phase solution has been reported by Ben- Jakob et al 
[6]. Here we present solutions for several N-junction arrays with different biasing configurations: 
the LL ("left"- "left"), LR ("left"-"right"), UD ("up"-"down") and CB ("central bias") biased array 
(Fig. 3). Although the LL and LR bias are directly applicable to parallel biased two-dimensional 
arrays, the other two configurations are used in other array architectures, such as series-parallel 
combinations, etc. 



Page 578 Third International Symposium on Space Terahertz Technology 



In-phase states 

The general solution of eq. (1) for the junction phases <()j is 

<|) j (T) = ajT + f J <x) + <(> j (0) 

CO = (<t>j(T)) 

f/T+T) = f /T), T= — , (ifx))=0 



0) 



(2) 



where "< >" denotes time average, co is the normalized DC voltage across junctions, fj's arc some 
general, periodic functions with zero time-average and <{>j(0) are constants. For the in-phase 
solution, the following condition must hold for every two neighboring junctions 

<t» j+1 (x)-<}) j (x) = 2 7tm j 

where mj must be an integer. Note that mj represents the number of fluxons in the j* loop. 
Condition (3) is fulfilled if 

f j(t) = f (t) 
j+1 CO = 4>j(t + tp 

X > =m i T (4) 

where Tj is the time delay between the phases of the two neighboring junctions. Substituting (3) 
into the system (l.a) and equating all the currents ij leads solutions 



m j = J a+ <P ex 


(LL bias) 


(5.a) 


m j = - y ) a + cp ex 


(LR bias) 


(5.b) 


m j =( Pex 


(UD bias) 


(5.c) 


mj = ([y]-j)a+(p ex 


(CB bias) 


(5.d) 



where "" in eq. 5.d denotes integer division, and parameter 'a' is given by 



Third International Symposium on Space Terahertz Technology Page 579 



X 



J DC 
a = 



2tcN (6) 

where ioc is me tota ^ DC biasing current. In order for mj to be an integer, which is the 
precondition for the in-phase solution, it is necessary that both (p ex and a be integers: 

<Pex= k q> 
Xi DC . 

a= =k 

2tiN (7) 

where k<p and k are integers. The only exception is the LR array with odd number of junctions N, 
where "a" must be an even integer (2 k). The arrays will be in phase for all currents i^ that satisfy 

, 2tcN 
'k= k — ~ 

* (8) 

Note that these in-phase states are achieved without external locking mechanisms. 

Numerical simulations 

System (1) is solved numerically using the 4th order Runge-Kutta method [7]. Figure 4 
shows the I-V and dV/dl-I curves of the 4 junction LL biased array with X,=20 and P=0.5. 
According to eq. (8), the in-phase states appear for current bias ik= 1.256 k (ik'=0.314 k for bias 
current normalized to the number of junctions, N, as in the Figure 4). The in-phase states are 
visible as voltage maximums in the I-V curve and sharp and deep minimums in dV/dl-I curve, for 
k=4 to 7. Similar structure has been observed experimentally by Clarke et al [8]. 

Other states 

The dV/dl-I curve of Figure 4 reveals considerable periodic structure between the in-phase 
states. Under certain conditions, that will be specified below, these "other" states, for current bias 
iDC * ik» correspond to the general solution of eq. (2) that satisfies the following: 

ViW-fjM-e/T) 

<t> j+1 (T) * *f x + T j ) 

1 J Vj (9) 



Page 580 Third International Symposium on Space Terahertz Technology 



where ej(x) is an error term and Hj does not need to be an integer. Furthermore, |0j is found from 
the same equations as mj (5), except that a and cp ex are no more restricted to integer numbers. In 
other words, all states of the parallel array are described with phases at neighboring junctions 
shifted in time by an amount determined by the DC biasing current and external magnetic field (eq. 
5). It is convenient to define the relative normalized time shift 8j between the waveforms of 
functions fj+i and fj 



6 j =Tj mod T = u,j mod 1 



(10) 



where "mod" is the modulus function, so that < 8j < 1. It has been shown by perturbation 
analysis [6] that in the case of 2-junction array solution (9) holds in the neighborhood of the in- 
phase state (ioc = *k + Aioc) and it has been suggested that it will hold for any state between the 
in-phase states, for the case of weak coupling (k »1). 

Figure 5.a shows the circled part of the dV/dl-I curve of Fig. 4. Points labeled "1" and "4" 
correspond to the in-phase states with k=4 and k=5, respectively. The voltage waveforms on 
individual junctions for these two states are shown in Figures 6.a and 7.a, respectively. Point "2" 
of Fig. 5.a correspond to DC biasing current ioc = 5.65, so eq. (5.a) gives Hi=4.5, H2=5 and 
|I3=5.5 for the number of fluxons in each loop. From eq. (10) we find that relative time shifts 
should be 0i=O.5 between the voltages of the junctions 2 and 1, 82=0 for junctions 3 and 2 and 
03=0.5 between junctions 4 and 3. Numerical simulations shown in Figure 6.b confirm this 
prediction. 

Point "3" of Fig. 5.a correspond to ioc = 5.42, and again from equations (5.a) and (10) 
we obtain 9i=0.333, 82=0.666 and 83=0. The voltage (and phase) at junction 2 is time shifted by 
third of a period from that of junction 1 , voltage at junction 3 is shifted by two-thirds from that of 
junction 2, so that it is in phase with junction 1. Finally, junction 4 is in phase with junctions 1 
and 3. This situation is shown in Figure 6.c. All other states can be determined in a similar 
fashion. 

Radiated power 

As a measure of how good an array performs as an oscillator for a particular bias, we 
calculate the available radiation power. We are interested in power array would radiate broadside 
in the far-field. We define m* harmonic power on unit (1Q) resistance as 

p(m) =(£Vj( m )) 2 



Third International Symposium on Space Terahertz Technology Page 581 



where Vj( m ) is m* harmonic voltage on j^ 1 junction. This power is given in units of (Ic R) 2 . We 
assume that the resistance R of the RS JC model (Fig. 2) includes both the radiation resistance and 
losses. The actual radiated power will at best be the power P( m ) on resistance -j. 

Figure 5.b shows the normalized first harmonic power radiated in the broadside direction 
for different states of the array. The maximum power is obtained only in the in-phase states 
(points "1" and "4"). Significant amount of power is also obtained in states where most of the 
array works in phase, as is the case with state "2". 

Array properties 

Several important properties of arrays of Fig. 3 can be derived from equations (5-10): 

1 ) In the absence of external magnetic field: 

- the UD array will be in phase for any DC bias; the in-phase state is a natural one 
for this array. Maximum power will be radiated at every operating frequency (Fig. 
5.b). 

- LR and CB arrays are symmetrical around the middle of the array; mN-j = -mj, 
which means that the junctions j and N-j are always in phase. 

2) The LR array is equivalent to the LL array with equivalent external magnetic field 

N 
<Pex = <Pex - y a. 

3) The larger the inductance parameter X , the more in-phase states will be found in the 
given current bias span (eq. 8), and corresponding DC voltage and operating frequencies 
span. Similarly, the larger the array (N), the more identifiable "other" states will be found 
in the dV/dl-I curve (Fig. 5.a). 

Magnetically steerable array 

When an array is biased in the in-phase state (irx: = ik) the normalized relative "time shift 
between every two neighboring junctions is the same and proportional to the external DC magnetic 
field: 



e j =e = q> ex modl, Vj 



(11) 



Page 582 Third International Symposium on Space Terahertz Technology 



This situation is shown in Figure 7. In Fig. 7.a the LL array is biased at the in-phase state (point 
"4", Fig. 5.a) with no external magnetic field. When an external magnetic field equal to a quarter 
of the flux quantum is applied, the time shift between the voltages of every two neighbors is equal 
to a quarter of period. 

The time shift 0T translates into the linear phase shift in the frequency domain 2tc0. 
Assuming that every junction drives one antenna, the quasioptical Josephson array becomes a 
phased array [9]. The angular position ceo of the main beam in the H-plane far-field radiation 

pattern of the linear Josephson phased array becomes 

,,2710 9 

cos(ao) = -- ="d^o 

2 it , a x 

_d x 

*o (12) 

where dx is the spacing between the antennas and Xo is the free space wavelength . The broadside 
radiation corresponds to oco = 90° (0 = 0). Equations (11,12) suggest that by changing the 
externally applied DC magnetic field cp e x it is possible to steer the Josephson array radiation pattern 
in the H-plane. As stated earlier, for the LL, LR and CB biased array, this is only true if the array 
is biased in the in-phase state. Since the UD array is always in the in-phase state, it can be steered 
using DC magnetic field at any bias. 

Limitations 

The expressions (9,10) derived for the "other" states will hold only in certain range of array 
parameters and bias conditions. We have derived expressions (5) for the in-phase states starting 
from (1) and assuming that all currents ij are equal These expressions always hold for the in- 
phase states. The same expressions (5) are found for "other" states if we solve (1) with an 
assumption that DC currents <ij> are approximately the same. The only part of the DC junction 
current that is different at every junction, due to DC self-field effects, is the supercurrent 
<sin(<j)j)>. This part will be negligible if either the biasing current per junction is -^» 1 or if 

there is non-vanishing capacitance (P >1). 

In our account of DC self-field effects we assumed the noiseless environment and the 
identical junctions. Therefore, the stability of in-phase and "other" states to noise and variations in 
junction and array parameters remains to be further investigated. 



Third International Symposium on Space Terahertz Technology Page 583 



Strongly coupled arrays (X < 1) 

When the inductance parameter X is small, it is evident from eq. (8) that the first in-phase 
state appears for very large DC biasing current, which translates into large DC voltage and 
operating frequency much above the critical frequency co c = (2e,#) Ic R. Depending on the 
capacitance parameter f$ and shunt resistance R, the operating frequency range is at best of the 
order of several co c . Therefore, the arrays with small inductances are operated in "other" states 
throughout the operating range. According to eq. (5), these states should be characterized with 
small time shifts between the junction phases/voltages. This is obvious, because in the limit of X - 
> the whole array operates as a single junction. 

Figure 8.a presents simulations for 4 junction LL biased array with small inductance (X = 
0.628). The normalized harmonic radiated power is shown in the wide range of bias currents. 
The bias points a=0.25 and a=0.5 with no radiated first harmonic power correspond to the states 
where half of the junctions are in-phase and the other half out-of-phase, according to (5.a). As 
seen from the Figure 8.b, the maximum first harmonic power is below that of the UD array, with 
all junctions in-phase. Figure 9 compares the first harmonic power of 4- and 5-junction array with 
Same parameters. The 5-junction array shows additional minimums in radiated power 
corresponding to states a=0.125 k. These minimums occur whenever there is one or more loops 
of array occupied by odd number of half-flux-quantums. 

It is clear that in order for the array with small inductance to approach the performance of 
the UD array in the wide operating range, the condition for the inductance parameter must be 
X«tj. More precisely, if the total time shift across the array is required to be less than a quarter of 



a period, the condition is: 



X iDCmax^j^y ( 13 ) 



where lDCmax is the DC bias at the end of the operating range. So, if we wanted a 4-junction LL 
biased array to approximately match the performance of UD array in Fig. 8.b, the inductance 

parameter should have been X = t™ instead of ■?. Such small inductances are rather unrealistic, 

specially because the X parameter is proportional to the critical current Ic (eq. 1) which should be as 
large as possible for large output power. 

As a final illustration, Figure 10 shows the influence of the Josephson junction capacitance. 



Page 584 Third International Symposium on Space Terahertz Technology 



The capacitance does not influence the occurrence or existence of in-phase and "other" states. 
However, it has a severe impact on radiated power. Even at not very big capacitance (|3 = 3) the 
first harmonic power is decreased at least an order of magnitude and the operating range is reduced 
below 2 cflfc.compared to the case of very small or no capacitance. 

Conclusion 

We have discussed how the DC biasing circuit determines the operation of linear parallel 
quasioptical Josephson junction arrays. We have shown that the maximum radiated power from 
the array can be achieved only at certain operating points, corresponding to the in-phase states. We 
have found that other states can be described by time-shifted phases and voltages of individual 
junctions, where the time-shift is determined from the DC biasing conditions. We have shown 
how the array can be steered from when in the in-phase state by application of DC magnetic field 
perpendicular to the array. 

When the inductance parameter X is large, there will be numerous in-phase bias points in 
the desired operating range. However, the stability of these states to noise and variations in 
junction parameters needs to be further investigated. When the inductance is relatively small, the 
radiated power will continuously change across the wide operating range, with several points 
where almost no power is radiated. 

If one dimensional quasioptical arrays are designed, the UD biased array is a definite 
choice, because it is in the in-phase state at every bias point. The operation of this array need to be 
further analyzed when junction parameters are not identical. The extension of our considerations to 
2-D arrays is straightforward, as long as rows of junctions are separately biased. 

Acknowledgment 

This work is supported by the Air Force Office of Scientific Research grant AFOSR-90- 



0233. 



References 

1. Robertazzi, R.P. and R.A. Buhrman, Josephson terahertz local oscillator. IEEE 
Trans. Magn., 1989. 25: p. 1384-1387. 

2. Wan, K., A.K. Jain and J.E. Lukens, Submillimeter wave generation using 
Josephson junction arrays. Appl. Phys. Lett., 1989. 54: p. 1805-1807. 



Third International Symposium on Space Terahertz Technology Page 585 



3. Benz, S.P. and C.J. Burroughs, Coherent emission from two-dimensional 
Josephson junction arrays. Appl. Phys. Lett., 1991. 58(19): p. 2162-2164. 

4. Wengler, M.J., A. Pance, B. Liu and R.E. Miller, Quasioptical Josephson 
Oscillator. DEEE Trans. Magn., 1991. 27(2): p. 2708-2711. 

5. Van Duzer, T. and C.W. Turner, Principles of Supeconductive Devices and 
Circuits. 1981, New York: Elsevier. 

6. Ben-Jacob, E. and Y. Imry, Dynamics of the DC-SQUID. J. Appl. Phys., 1981. 
52(11): p. 6806-6815. 

7. Press, W.H., B.P. Flannery, S.A. Teukolsky and W.T. Vetterling, Numerical 
Recipes in C: The Art of Scientific Computing. 1988, Cambridge, Mass.: Cambridge 
University Press. 

8. Clarke, J. and T.A. Fulton, Origin of Low-Voltage Structure and Asymmetry in the 
I-V Characteristics of Multiply-Connected Superconducting Junctions. J. Appl. Phys., 
1969. 40(1.1): p. 4470-4476. 

9. Steinberg, B.D., Principles of Aperture and Array System Design. 1976, New 
York: John Wiley & Sons, Inc. 



-. '5-' 



Page 586 



Third International Symposium on Space Terahertz Technology 



Idc 



+ 



Idc 



+ 



X )( X X )( 



Idc 



Idc 



+ 



X X 






X X 



Idc 



X X X X X 



Idc 



Figure 1. Separate row bias for parallel 2-D Josephson junction array. 



Third International Symposium on Space Terahertz Technology 



Page 587 



out 





Figure 2. General biasing scheme for one-dimensional parallel Josephson junction array. 
The RSJC model used is shown below. 



Page 588 



Third International Symposium on Space Terahertz Technology 



+ 



Idc 



Idc XN X 2X lXL-Lbias 



a) 



+ 



L-R bias 




+ 



X X- - 

N N-l 



UIdc 



\ 



72- % 



'2 i 



X CB bias 



^Idc 



b) 



c) 



d) 



Figure 3. Four common biasing configurations of 1-D array. 



Third International Symposium on Space Terahertz Technology 



Page 589 



> 



1- 







k=4 




h 



k=5 



k=4 



1 2 

Voltage [IcR] 



k=5 k=6 




k=7 







1 2 

Current [N Ic] 



Figure 4. Dynamic resistance and I-V curve of 4 junction LL biased array with X = 20 and 

j} = 0.5. The in-phase states are seen as small steps in the I-V curve and sharp minimums 
in dV/dl-I curve , labeled k=4 to k=7. The area inside a circle is shown enlarged in Fig. 5. 



Third International Symposium on Space Terahertz Technology 




5q 

<? 4.5- 

& a'- 

— 3.5^ 

O 

a< 

o 

•^■« 

c 
o 



DC Voltage [Ic R] 



a) 



J-i 



2.5 -i 
2-. 

1.5-i 
1 J 

0.5- 







/ 

UD array 



0.8 



0.9 



| i 1 1 \ m\ 1 1 1 i ■"] r— i— r— i [— 

1 1.1 1.2 1.3 



i i i 



1.4 



Dc Voltage [Ic R] 



b) 



Figure 5. Four junction array, LL bias, X = 20, p = 0.5. a) Enlarged portion of the dV/dl- 
I curve (Fig. 4) with in-phase states labeled "1" and "4" and two "other" states "2" and "3". 
The waveforms of individual junction voltages for these states are shown in Fig. 6 and 7. 
b) First harmonic power that correspond to states in a). The power is maximum in the in- 
phase states and equal to that of the equivalent UD array. 



Third International Symposium on Space Terahertz Technology 



Page 591 




240 

time [Lj/R] 



a) 




240 

time [Lj/R] 



250 



b) 




240 
time [Lj/R] 



c) 



Figure 6. The waveforms of individual junction voltages for the states of Fig. 5. a) In- 
phase state, b) Junctions 1 &4 in phase, junctions 2 & 3 in phase , but out of phase with 
(1&4). No power radiated, c) Junctions 1,3 & 4 in phase, junction 2 "leads" a third of a 
period. Half the maximum power is radiated. 



Page 592 



Third International Symposium on Space Terahertz Technology 



Z.3- 


©A 






2- 








ST 








a 1 - 5 - 








> 








0.5- 


#4 


#2 


(pex=0 


o : 


#3 - 


■-• #1 

1 1 ' 





230 



240 

time [Lj/R] 



250 



a) 




240 

time [Lj/R] 



250 



b) 



Figure 7. The waveforms of individual junction voltages for the in-phase state labeled "4" 
in Fig. 5. a) No DC magnetic field supplied, b) Quarter of the flux quantum in every 
loop supplied by the external magnetic field. Voltage waveforms uniformly shifted by 
quarter of a period. The main beam of the radiation is steered from the broadside direction. 



Third International Symposium on Space Terahertz Technology 



Page 593 



LL bias 



< 



0) 

o 

o 

'2 
o 







0.000001 



I I I I 



0.3 

a [FiO] 



0.6 




12 3 4 5 6 

Vdc [IcR] 

Figure 8. a) Hannonic power radiated by the 4-junction LL-biased array with X = n/5 and 

(} = 0.03. Minimums correspond to odd numbeT of half-flux-quantums in some of the 
loops, b) Comparison between the LL and UD array with same parameters. Varying 
amount of power is radiated in the very broad operating range, but never a maximum 
possible power, as in the case of UD array. 



Page 594 



Third International Symposium on Space Terahertz Technology 



4.5 -r 



cs 


4- 


< 




^^*\ 


. 




3.5 r 


HH 


■ 


"W 


■ 


•— ' 


V 


I* 




<u 


_ 


o 


2.5^ 


D. 


. 




2- 


e 


- 


o 


1.5- 


ed 




XI 


1- 


•*-» 


. 


C/5 


- 




0.5^ 







4 junctions 



5 junctions 



A 

! \ 




o 



i i i i i i i i i i 



Vdc [IcR] 



Figure 9. Comparison between the 4-junction and 5-junction small inductance arrays; \ = 

7i/5, P = 0.03. As the number of junctions increase, more maximums and minimums 
appear throughout the operating range. 



10rr 



< 

u 




beta=3.00 



- •• - beta=0.03 



2 2.5 

DC Voltage [Ic R] 



i i i i i i 

3.5 



Figure 10. Four junction LL biased array, X = jc/10: Influence of the Josephson junction 
capacitance on power output. As the capacitance is increased, the maximum power and the 
operating range rapidly decreases. 



Third International Symposium on Space Terahertz Technology Page 595 



N 9 3 - 2fr ^ 5 



MONOLITHIC MILLIMETER-WAVE DIODE GRID FREQUENCY 

MULTIPLIER ARRAYS 



Hong-Xia L. Liu, X-H. Qin, L. B. Sjogren, W. Wu, E. Chung, 

C. W. Domier, N. C. Luhmann, Jr., 

Center for High Frequency Electronics 

Department of Electrical Engineering 

University of California 

Los Angeles, California 90024-1594 



ABSTRACT 



Monolithic diode frequency multiplier arrays, including barrier-N-N + (BNN) dou- 
bler, multi-quantum-barrier- varactor (MQBV) tripler, Schottky-quantum-barrier-varactor 
(SQBV) tripler, and resonant-tunneling-diode (RTD) tripler arrays, have been suc- 
cessfully fabricated with yields between 85% and 99%. Frequency doubling and/or 
tripling have been observed for all the arrays. Output powers of 2.4-2.6 W (7?=10- 
18%) at 66 GHz with the BNN doubler and 3.8-10 W ( 77=1.7-4%) at 99 GHz with 
the SQBV tripler have been achieved. 



INTRODUCTION 

Quasi-optical spatial power combining techniques have, in recent years, been ex- 
tensively investigated for millimeter and submillimeter power generation [1] [2] [3]. 
Our research efforts have been focused on monolithically integrating thousands of 
solid state devices to generate Watt level harmonics in the millimeter region. In 
addition to arrays of familiar devices such as the BNN and RTD, several new de- 
vice concepts (MQBV and SQBV [4] [5]) have been developed in the course of this 
work which promise to significantly improve the performance of frequency multipliers. 
Arrays of all of these devices have been successfully fabricated and tested. Several 
exciting results have been obtained. Improvements both in device design and in the 
matching system are underway to further optimize array performance. 



Page 596 



Third International Symposium on Space Terahertz Technology 



FABRICATION AND RESULTS 



(a) BNN frequency doubler array 

A four-mask process based on the self-aligned aluminum Schottky diode process 
employed by C. Zah [6] is utilized to fabricate the BNN doubler array. Figure 1 
shows an individual BNN device after fabrication. The array was then mounted on a 
quarter wavelength thick quartz plate. No bias is required due to the adjustment of 
the build-in voltage resulting from the 6 doped layer (see Fig.2). The test system for 
the BNN frequency doubler is shown in Fig.3 except that the output filter was not 
used. Cutoff waveguides for the fundamental have been used to prevent contamination 
of the detected signal due to the pump signal. A variety of tests were employed 
to conclusively verify that the received signal was actually frequency doubled. All 
waveguide components including attenuators have been calibrated using at least two 
methods. The input and receiving horns have also been calibrated and compared 
with the theoretical values. Figure 4 displays the measured RF results. An output 
power of 2.4-2.6 W and a maximum efficiency of 10-18% have been achieved. The 
calculated cutoff frequency based on the measured low frequency parameters of this 
array is 280 GHz, which results in a maximum theoretical conversion efficiency of 
~20%. 



5|im 




Figure 1. The fabricated individual BNN diode. 




f*t t *t t M"t > *y*? w ?*f w W**t^ 




Semi-insulating GaAs substrate 



2000A Al 

; 200A GaAs undoped 

* 50A n GaAs 2x10 cm' 

1200A n GaAs 6xlo"cni 



. it •> 

1.3jim n GaAs 4x10 cm 



Figure 2. The profile of the BNN diode. 



Third International Symposium on Space Terahertz Technology 



Page 597 




2 5 


-e Output Power- — »•-• 


Efficiency 

1-7 


$ 2 


_i i i i 1 i i i i | i i i i | i i i i | i i i 
'- ° o © / 


't> 


■© 


10 


% 1-5 

o 

t 1 
% 

6 °- 5 
o 


ft. > X5 


, 1 . . 


8 g 

6 1 

4 9 

2 w 
n 


5 10 15 20 

Input Power (W) 


25 


30 



Figure 3. The frequency multiplier setup. 



Figure 4. Measured frequency doubling output power 
and efficiency at 66 GHz as the function of 
the input power. 



(b) Frequency tripler arrays 

MQBV, SQBV and RTD arrays have been successfully fabricated using a back- 
to-back processing method [7]. As a result, all these arrays are suitable for odd 
harmonic generation due to the resulting symmetric C-V characteristics. Figure 5 
displays the array layout. The period of each cell is 400 /ttn. There are 2250 devices 
on the MQBV array, 1300 devices on the SQBV array, and 500 devices on the RTD 
array, respectively. These arrays are tested using the system shown in Fig. 3. Since 
the input is identical for both the doubler and tripler arrays, the same input filter 
and tuning slabs have been used. The output filter for the frequency tripler is ah 
inductive metal grid array [8], and quartz tuning slabs with a thickness of a quarter 
wavelength at the tripled output have been used for the output impedence matching. 
Figure 6 shows the measured output power and efficiency of the SQBV array at an 
output frequency of 99 GHz. An output power of 3.8-10 W and an efficiency of 1.7-4% 
have been achieved. Due to excessive pumping in the initial tests, the performance of 
the MQBV array was degraded significantly (f c dropped from 550 GHz to 100 GHz). 
However, an output power of 0.1 W and an efficiency of 0.4% have been obtained for 
the degraded array which is in good agreement with the theoretical prediction (0.5%). 
Finally, a frequency tripling signal has also been observed with the RTD array. Tests 
are underway to measure the output power and efficiency as the function of the input 
power. 



Page 598 



Third International Symposium on Space Terahertz Technology 






< MQBVr 







' 400|i 

I 



if" 

urn 



4 

3.5 

3 






— Output power 


(W) Efficiency (%) 


1.8 
1.6 
1.4 






* 

4 


1 | 1 1 ! 1 | 1 1 1 1 | 1. 1 1 1 


2.5 






/ 


^/ 




2 










1.2 
1 


1.5 












1 






i X 


- 


0.8 


0.5 




I 

/ 






0.6 







50 100 


150 200 250 300 








Input 


power (W) 





Figure 5. The MQBV and SQB V arrays layout. 



Figure 6. Measured frequency tripling output power 
and efficiency at 99 GHz as the function of 
the input power. 



CONCLUSIONS 



BNN, MQBV, SQBV, and RTD frequency multiplier arrays have been successfully 
fabricated with yields between 85 and 99%. All of these arrays have yielded frequency 
multiplication. An output power of 2.4-2.6 W with maximum efficiency of 10-18% 
has been achieved at 66 GHz with the BNN doubler array; an output power of 3.8-10 
W with a maximum efficiency of 1.7-4% has been achieved at 99 GHz with the SQBV 
array. 



ACKNOWLEDGEMENTS 



This work was supported by the US Army Research Office and the US Department 
of Energy. The authors wish to thank Dr. J. Maserjian and P. Smith of Jet Proposion 
Lab. for providing the processing facilities and the MBE wafers. The authors also 
wish to acknowledge the generous assistance of Dr. R. Bhat and Dr. L. Florez 
of Bellcore, Professor M. Spencer of Howard University, and Dr. A. Miura of the 
Yokogawa Electric Corp. in providing MBE and MOCVD wafers for these studies. 



Third International Symposium on Space Terahertz Technology Page 599 



REFERENCE 

1. C.F. Jou, W.W. Lam, H.Z. Chen, K.S. Stolt, N.C. Luhmann,Jr., and D.B. Rutledge, 

"Millimeter Wave Diode-grid Frequency Doubler," IEEE Trans, on Microwave Theory 
and Techniques, 36, No. 11,1 988. 

2. W.W. Lam, C.F. Jou, N.C. Luhmannjr., and D.B. Rutledge, "Millimeter- W ave Diode- 

grid Phase Shifters," IEEE Transactions on Microwave Theory and Techniques 36, No. 
5, pp. 902, 1988. 

3. Z.B. Popovic, R.M. Weikle, M. Kim, K.A. Potter, and D.B. Rutledge, "Bar Grid 

Oscillators," IEEE Transactions on Microwave Theory and Techniques , 38 No. 3, 
pp. 225, 1990. 

4. Hong-Xia L. King, L.B. Sjogren, and N.C. Luhmann,Jr., "New Concepts for High 

Frequency and High Power Frequency Multipliers and Their Impact on Quasi- Optical 
Monolithic Array Design", International Journal of Infared and Millimeter Waves, 
13, pp.251, 1992. 

5. Hong-Xia L. King, N.C. Luhmannjr., X-H. Qin, L.B. Sjogren, W. Wu, D.B. Rutledge, 

J. Maserjian, U. Lieneweg, C. Zah, and R. Bhat, "Millimeter Wave Quasi- optical 
Active Arrays", Proc. and Conference on Space Terahertz Technology, pp. 293-305, 
Feb. 1991. 

6. C. Zah, D.P. Kasilingam, J.S. Smith, D.B. Rutledge, T. Wang, and S.E. Schwartz, "Millimeter- 

wave Monolithic Schottky Diode Imaging Arrays", Int. J. of Infrared and Millimeter 
Waves, 6, pp. 981-997, 1985. 

7. R.J. Hwu, C.F. Jou, N.C. Luhmann, Jr., M. Kim, W.W. Lam, Z.B. Popovic, D.B. 

Rutledge, "Array Concepts for Solid-State and Vacuum Microelectronics Millimeter- 
Wave Generation," IEEE Trans, on Elec. Dev., 36, No. 11, pp. 2645-2650, 1989. ' 

8. Hong-Xia L. Liu, L.B. Sjogren, and N.C. Luhmann,Jr.,"Grid Bandpass Filters for 

Quasi- Optical Frequency Multiplier Array Application", submitted for publication in 
Microwave and Optical Technology Letters, 1992. 




Page 600 Third International Symposium on Space Terahertz Technology 

PLANAR GaAs DIODES FOR THz FREQUENCY MIXING APPLICATIONS 



William L. Bishop, Thomas W. Crowe, Robert J. Mattauch, and Hasan Dossal 

Semiconductor Device Laboratory 
Department of Electrical Engineering _ ^ yy ry (» 

Thornton Hall f| 9 3 " * • f 

University of Virginia 
Charlottesville, VA 22903-2442 



I. Introduction 

For many scientific applications in the terahertz frequency range, heterodyne reception 
is the only technique which exhibits the necessary combination of high spectral resolution, 
large instantaneous bandwidth and excellent sensitivity. A key component in these receivers 
is the non-linear resistive mixer element. In general, the mixer element should have high 
intrinsic speed, the sharpest possible non-linearity, low parasitic element values, low intrinsic 
noise and impedance levels which can be easily matched to the RF circuit. However, no 
single device exhibits all of these properties and some tradeoffs are necessary [1]. 

The GaAs Schottky barrier diode is the most widely used mixer element at 
submillimeter wavelengths. These diodes are commonly used in the temperature range from 
300 K to 10 K and have demonstrated excellent performance from below 100 GHz to over 
3 THz [2,3]. The closest competitor for Schottky diodes is the SIS element which has 
demonstrated record sensitivities at millimeter and long-submillimeter wavelengths [4,5,6]. 
However, SIS devices are not yet competitive at terahertz frequencies and present 
superconductor mixer elements require cryogenic cooling which increases the cost and size 
of the receiver system. 




Third International Symposium on Space Terahertz Technology Page 601 

Schottky barrier diodes for terahertz applications are typically fabricated as a micron 
to sub-micron circular anode metallization on GaAs which is contacted with a sharp wire 
(whisker). This structure has the benefits of the simplicity of the fabrication of the diode chip, 
the minimal shunt capacitance of the whisker contact and the ability of the whisker wire to 
couple energy to the diode. However, whisker-contacted diodes are costly to assemble and 
difficult to qualify for space applications. Also, complex receiver systems which require many 
diodes are difficult to assemble [7,8]. 

The objective of this paper is to discuss the advantages of planar Schottky diodes for 
high frequency receiver applications and to summarize the problems of advancing the planar 
technology to the terahertz frequency range. Section II will discuss the structure, fabrication 
and performance of state-of-the-art planar Schottky diodes. In Section m the problems of 
designing and fabricating planar diodes for terahertz frequency operation are discussed along 
with a number of viable solutions. Section IV summarizes the need for futher research and 
cooperation between diode designers and RF engineers. 

II. Planar Mixer Diodes 

Planar Schottky barrier diodes have been developed by numerous laboratories over the 
past ten years [9,10,11,12]. This effort has resulted in many benefits. Not only has the 
troublesome and somewhat fragile whisker contact been eliminated, but receivers which 
require two or more individual diode chips, such as balanced mixers, are much easier to 
assemble. Single chips with two or more diodes in a fixed configuration, such as an 
antiparallel diode pair, are easy to fabricate and the extension of the diode contact pads to 
form a planar antenna has been demonstrated. Future work should allow additional receiver 
components such as filters, oscillators and amplifiers to be integrated with the diode. 



Page 602 Third International Symposium on Space Terahertz Technology 

The surface channel planar diode, shown in Figs. 1 and 2, has been developed for use 
at both millimeter and submillimeter wavelengths [9,13]. The chip substrate is semi-insulating 
GaAs. The epitaxial GaAs structure consists of a thin n-type layer on top of a thick, heavily 
doped n+ buffer layer. The anode is formed on the n-type GaAs with Si0 2 providing 
passivation and insulation. An ohmic cathode pad is formed on one end of the chip in close 
proximity to the anode. The anode is connected to a bonding pad by means of a narrow 
finger. A trench is formed beneath the finger and completely across the width of the chip to 
isolate the anode contact pad from the cathode. The isolation trench can be etched deeply into 
the semi-insulating substrate and the wall of this trench can be positioned very close to the 
anode. These two features combine with the inherent air-bridge to reduce the shunt 
capacitance between the contact pads and the shunt capacitance from the contact finger to the 
conductive GaAs of the cathode. This structure produces lower shunt capacitance than other 
designs which rely on mesa or proton isolation. 

The major fabrication steps of the surface channel structure are illustrated in Fig. 3. 
Starting with the GaAs wafer (1), a layer of silicon dioxide is deposited using chemical vapor 
deposition from silane and oxygen (2). The ohmic contact region is patterned, the Si0 2 and 
n-GaAs are removed and the ohmic contact metallization is deposited and alloyed (3). An 
opening for the anode is patterned and etched into the Si0 2 , leaving a thin layer of oxide to 
protect the GaAs until the anode metallization can be deposited. The remaining oxide in the 
anode well is removed with buffered hydrofluoric acid and platinum and gold are 
electroplated to form the diode and fill the oxide well (4). A thin layer of chromium and gold 
is deposited over the entire wafer by sputtering. Photoresist is applied and patterned and gold 
is plated into the opening to form the anode contact pad and finger. The resist is removed and 
the sputter deposited gold and chromium surrounding the anode contact pad and finger are 



Third International Symposium on Space Terahertz Technology 



Page 603 



^^^ Ohmic Contact 



Surface Channel 
Air Bridge Finger 

Anode Contact 

Pad 




Anode 
(beneath finger) 



Semi — Insulating 
GaAs Substrate 



Figure 1. Surface Channel Planar Diode Structure 




Figure 2. SEM Photographs of a Surface Channel Planar Diode 



Page 604 



Third International Symposium on Space Terahertz Technology 



1. n / n+ / S.I. GaAs 



n GaAs 



n 


n + 




S.I. 


GaAs 



2. Deposit Silicon Dioxide 




\/////////7////////////////////////A 



n + 



S.I. GaAs 



3. Forn Dhnic Contact 




'////////////////////////, 


D.C. 


n 


n + 


S.I. GaAs 



4, Form Anode 




'/////////////////////Wy 


D.C. 


n 


n + 


S.I. GaAs 



5. Forn Anode Pad/Finger 




Anode 

Pad V y///// //A 
|SiD 2 | 



Dhnic 
Contact 



'///////////////7/////M. 



Anode Pad/Finger 



' r inqs 

7 777? 



n + 



D.C. 



S.I. GaAs 



6. Etch Surface Channel 



Anode 




Dhnic 
Contact 






V7 


Pad 


S.I. 
GaAs 





Anode Pad/Finqer 
V////////777\ 

Air 



n + 



i 



D.C. 



S.I. GaAs 



Figure 3. Surface Channel Diode Fabrication Sequence 



Third International Symposium on Space Terahertz Technology 



Page 605 



etched away (5). Finally, the surface channel is patterned with photoresist and the Si0 2 and 
GaAs are etched to form the isolating trench (6). 

This fabrication sequence offers several advantages compared to other configurations: 
(1) expensive and troublesome proton bombardment is not required, (2) planarization is 
unnecessary, and (3) the wafer surface is nearly flat for the critical steps of anode formation 
and anode-to-finger alignment. 

SEM photographs of two surface channel diode chips are shown in Figs. 4 and 5. The 
SC2T1 single anode chip is about 125 x 375 x 75 microns. This device has a total 
capacitance of about 14 fF, zero-bias junction capacitance of 2.5 fF and series resistance of 
12-15 Q. This gives a figure-of-merit cutoff frequency of 4.2 THz for the junction. The 
SC2T1 has been tested in a room temperature mixer at 345 GHz with a mixer noise 
temperature of 1,370 K (DSB) and a conversion loss of 9.5 dB (SSB) [14]. This is 
comparable to the best whisker-contacted diode results. The SC1T4 chip is an antiparallel 
diode pair for subharmonic pumping. It is only 80 x 180 x 50 microns. These chips have a 




Figure 4. SC2T1 Planar Diode Chip Figure 5. SC1T4 Planar Antiparallel 

Diode Pair 



Page 606 



Third International Symposium on Space Terahertz Technology 



total capacitance of about 16 fF, zero-bias junction capacitance of 3 fF per anode and series 
resistance of 7-9 Q. This diode has been successfully used in a room temperature mixer at 
205 GHz with a mixer noise temperature of 800 K (DSB) and a conversion loss of 4.4 dB 
(DSB) using an LO of approximately 100 GHz [15]. This result is better than has been 
previously reported for antiparallel subharmonic mixers of either planar or whisker-contacted 
design. 

A dual anode planar diode chip for balanced mixer operation is shown in Fig. 6. This 
chip was developed in collaboration with Aerojet General, Electronic Systems Division under 
the direction of Robert Haas. This configuration allows individual DC bias of each diode. 
This device has excellent DC electrical characteristics and is being evaluated in a 100 GHz 
mixer. 





Figure 6. Planar Balanced 
Mixer Diode Chip 




Third International Symposium on Space Terahertz Technology Page 607 

HI. Planar Diodes for THz Frequency Applications 

The surface channel diode structure must be optimized for terahertz operation. These 
improvements reflect the fundamental need to reduce the RsC j0 product, minimize shunt 
capacitance, and to efficiently couple energy into the diode. These optimization issues are 
addressed in the following subsections: 

A. Reduction of Anode Diameter 

Theory and experimental results with whisker-contacted diodes have shown that very 
small anodes combined with higher active layer doping are necessary for good performance 
in the THz range [16]. Whisker-contacted diode chips have been fabricated at UVa with 
anodes as small as 0.25 microns using direct write electron beam lithography and reactive ion 
etching [17]. Planar diodes have been fabricated at UVa with 0.5 micron diameter, anodes 
using optical lithography and reactive ion etching. We are also investigating a novel 
Electroplate Window Shrink (EWS) technique. In this method, circular openings are etched 
through a thin (0. 1 micron) metal layer which overlies silicon dioxide, using UV lithography 
and wet or dry etch methods. Metal is then electroplated onto this thin conductive layer. 
Since the plating proceeds laterally as well as vertically, the diameter of the openings is 
reduced. These reduced-diameter windows are then used as a non-eroding mask to RIE etch 
the silicon dioxide. Etched wells less than 0.2 microns in diameter have been formed in this 
manner. 

It should be realized that the main issue is not just the fundamental task of forming 
small anode wells, but also the problems of uniformity and control of anode size. The UVa 
anode formation process depends on leaving a thin layer of Si0 2 of known thickness in the 
bottom of the anode wells after RIE. This protective layer is removed by etching with 



Page 608 Third International Symposium on Space Terahertz Technology 

■ buffered hydofluoric acid just prior to anode formation. Underetching of this remaining oxide 
results in open circuits or high resistance. Overetching can result in high Cj and in some 
cases, excessive diode noise [18]. Unlike whisker-contacted diode chips which can be etched 
and plated on a chip-by-chip basis, all planar diode anodes on a wafer are formed 
simultaneously. This obviously places very tight limits on dielectric thickness, thickness 
uniformity and etch rate calibration. 

For these reasons, it would be most helpful to have a very thin RE etch stop layer 
to protect the GaAs. This etch stop layer would relax the requirements for oxide thickness 
and uniformity and allow reasonable overetching during RIE without the risk of damage or 
contamination of the junction area. Schemes which utilize mutiple layers of different 
dielectrics could, in principle, satisfy this need. A very thin layer (100-500 A) of silicon 
dioxide could first be deposited onto the GaAs. This would be followed with a thicker layer 
of another dielectric, such as silicon nitride, polyimide or boron-doped silicon dioxide. This 
thick layer could be patterned and selectively etched (possibly with a dielectric or metal 
mask) so that the underlying thin layer of oxide acts as an RIE etch stop. Research in this 
important fabrication area will provide improved control of anode diameter and the reliable 
production of sub-half micron planar diode anodes. 

B. Optimization of Chip Geometry 

The dimensions and layout of planar diode chips must be optimized for terahertz 
frequency applications. The volume of the chip must be reduced to minimize the field 
disturbing effect of high dielectric constant GaAs and to allow the devices to fit into the 
smaller waveguides which are required at higher frequencies. The geometry of the planar 
diode must be improved to minimize shunt capacitance. 



Third International Symposium on Space Terahertz Technology 



Page 609 



Shunt capacitance in the planar diode structure can be separated into two primary 
components: capacitance from the anode contact finger and pad-to-pad capacitance through 
the high dielectric constant substrate. Finger capacitance will be reduced by several means. 
The width of the contact finger can be reduced from the current value of about 2.5 microns 
to 1 micron. Improved mask design, alignment and surface channel etch control will allow 
the surface channel wall to be etched as close as possible to the anode. A thick (1 micron) 
dielectric, perhaps a polyimide, would further reduce finger capacitance. 

Pad-to-pad shunt capacitance can be reduced by decreasing pad area, increasing pad 
separation, increasing the surface channel depth and/or reducing substrate thickness. Our 
present technology produces chips which are 50 microns thick with pads which are 30 x 60 
microns and a surface channel depth of 10 microns. For the lowest possible pad-to-pad 
capacitance, the GaAs substrate can be removed. This has been demonstrated in a procedure 
that replaces the GaAs with quartz, as shown in Fig. 7 [19]. The quartz substrate can be 
permanent or it can be removed once the chip is bonded to a circuit as shown in Fig. 8. 




Figure 7. Surface Channel Diode Chip Figure 8. Surface Channel Diode Chip 
with Quartz Substrate with Quartz Substrate 

Removed After Bonding 



Page siO Third International Symposium on Space Terahertz Technology 

The effect of finger length on planar diode performance is an important issue, 
particularly for waveguide mixers. Longer fingers result in reduced pad-to-pad capacitance 
but increased finger inductance. A new mask set has been fabricated which will provide 
small area, antiparallel planar diodes with finger lengths from 10 microns to 50 microns in 
10 micron steps on the same wafer. This mask was designed in collaboration with Peter 
Seigel of JPL and the devices will be RF tested at JPL in a waveguide mixer at frequencies 
as high as 600 GHz. 

Very short contact fingers are required in integrated antenna designs. Surface channel 
formation is very difficult when the contact finger is under 10 microns in length. Research 
is underway to characterize a combination of chlorine-based reactive ion etching and wet 
chemical etching processes to form the surface channel isolation trench with these short 
contact fingers. The new mask sets for both the small area antiparallel chips and the log 
periodic antenna designs include levels for this new process. 

C. Minimization of Ohmic Contact Resistance 

Ohmic contact resistance contributes to diode series resistance and thus reduces cutoff 
frequency. As contact pad dimensions shrink, ohmic contact resistance increases. This is of 
particular importance in the case of integrated antenna devices where the pad geometry is 
dictated by the antenna design. Specific contact resistance can be improved by using a very 
highly doped buffer layer and through the use of a more advanced ohmic contact technology. 
For example, ohmic contacts to an n 4 " 1 " InGaAs layer are reported to have specific contact 
resistivity as low as 10" 7 Sl-cm 2 , a factor of 50 to 100 better than our present ohmic contacts. 



Third International Symposium on Space Terahertz Technology Page 611 

This would be most beneficial for planar THz antenna structures which require small pad 
geometries near the anode. 

D. Integration of Antenna Structures 

The problem of efficient energy coupling to the planar diode is exacerbated at higher 
frequencies where the wavelength begins to approach the size of the chip. For whisker- 
contacted diodes, the whisker itself is used as the antenna element and mixers with a long 
whisker (4X) positioned parallel to the axis of a corner cube have demonstrated excellent 
performance at frequencies as high as 4 THz [3]. 

Another approach for planar diodes is to integrate an antenna, in the form of a bow-tie 
or log periodic shape onto the chip [20]. The fabrication is straightforward, with the antenna 
being an extension of the anode and cathode pads and the radiation can be coupled to the 
antenna through the substrate (GaAs or quartz). An integrated bowtie antenna-diode is shown 
in Fig. 9. It is 700 x 1000 x 50 microns thick with a 0.5 micron anode and an 8 micron 
finger length. Preliminary RF testing with unoptimized coupling produced video response of 
10 V/W. 

Optimization of the integrated antenna will require close interaction between diode 
designers and RF engineers. As a first step towards this goal, a mask set for the fabrication 
of log periodic antenna-diodes has been designed in cooperation with Gabriel Rebeiz of the 
University of Michigan and devices will be fabricated in the near future. With proper diode 
design and good coupling of energy- to the antenna and the diode, it is hoped that RF 
performance will exceed that of the best whisker-contacted diodes. 



Page 612 



Third International Symposium on Space Terahertz Technology 





Figure 9. Integrated Bowtie Antenna-Diode 



IV. Discussion 

Development of the planar mixer diode was driven by the need for a rugged device 
which is inherently simple and easy to assemble in a mixer. However, the tradeoffs for this 
structural ruggedness and simplicity are a more complex and expensive fabrication procedure, 
and a more complex chip geometry with larger shunt capacitance. The RF circuit must be 
redesigned to efficiently couple energy to the diode. In spite of these changes, planar GaAs 
Schottky barrier diodes have demonstrated performance in the millimeter wavelength range 
equal to or better than that of the best whisker-contacted diodes. 

Successful operation of planar diodes at THz frequencies will require several 
improvements in the diode chip including reduced anode diameter, improved control of anode 
diameter, smaller chip dimensions to reduce shunt capacitance, and reduced ohmic contact 
resistance. These concerns are being addressed through research of novel structures and 
fabrication methods. Successful application of planar diodes in the THz frequency range will 
also require optimization of the embedding circuitry and improved methods of coupling 
energy to the diode. Research is underway to apply novel antenna designs to this problem and 
to begin to test high performance planar diodes in waveguide assemblies and to test integrated 



Third International Symposium on Space Terahertz Technology Page 613 

antennas in open structure mixers. The success of this effort will be hastened by very close 
interaction and cooperation between diode designers and RF engineers. 

Acknowledgements 

The authors wish to express their sincere appreciation to Peter Siegel of the 
Jet Propulsion Laboratory, Israel Galin and Robert Haas of Aerojet General and Gabriel 
Rebeiz of the University of Michigan for many helpful discussions regarding the design of 
planar diodes. This work was supported by the National Science Foundation (ECS-8720850), 
the. U.S. Army and the Jet Propulsion Laboratory (958202). 

References 

[1] T.W. Crowe, R.J. Mattauch, H.P. Roeser, W.L. Bishop, W.C.B. Peatman, "GaAs 
Schottky Diodes for THz Mixing Application," Invited paper accepted for IEEE Proa, 
Special Issue on Terahertz Technology, to appear in 1992. 

[2] C.R. Predmore, A.R. Raisanen, N.R. Erikson, P.F. Goldsmith, and J.L.R. Marrero, "A 
Broad-Band, Ultra-Low-Noise Schottky Diode Mixer Receiver for 80-1 15 GHz," IEEE 
Trans. Microwave Theory Tech., Vol. MTT-32, pp. 498-506, May 1984. 

[3] H.P. Roser, R. Wattenbach, E.J. Durwen, and G.V. Schultz, "A High Resolution 
Spectrometer for 100 urn to 1000 um and Detection of CO (J=7-6), CO (J=6-5) and 
13 CO (J=3-2)," Astron. Astrophys., 165, 287-299, 1986. 

[4] S.K. Pan, A.R. Kerr, M.J. Feldman, A. Kleinsasser, J. Stasiak, R.L. Sandstrom, and 
W.J. Gallagher," An 85-116 GHz SIS Receiver Using Inductively Shunted Edge- 
Junctions," IEEE Trans. Microwave Theory Tech., Vol. MTT-37, pp. 580-592, March 
1989. 

[5] A. W. Lichtenberger, D.M. Lea, and A.C. Hicks, "Nb-based SIS Mixer Elements for 
Millimeter and Submillimeter Wavelengths," 2nd Int'l. Symp. on Space Terahertz 
Tech., pp. 439-458, Feb. 1991. 

[6] J. Zmuidzinas and H.G. LeDuc, "Quasi-Optical Slot Antenna SIS Mixers," 2nd Int'l. 
Symp. on Space Terehertz Tech., pp. 481-490, Feb, 1991. 



Page 614 Third International Symposium on Space Terahertz Technology 



[7] J.W. Waters, "A Proposal of the Earth Observing System, Microwave Limb Sounder," 
Jet Propulsion Laboratory, California Institute of Tech, July 1988. 

[8] M.A. Frerking, "The Submillimeter Mission (SMMM) Heterodyne Instrument," 2nd 
Int'l. Symp. Space Terahertz Tech., pp. 17-31, Feb. 1991. 

[9] W.L. Bishop, K. Mckinney, R.J. Mattauch, T.W. Crowe, and G. Green, "A Novel 
Whiskerless Schottky Diode for Millimeter and Submillimeter Wave Applications," 
Proc. 1987 IEEE MTT-S Int'l Symp, pp.607-610, June 1987. 

[10] J.W. Archer, R.A. Batchelor, and C.J. Smith, "Low-Parasitic, Planar Schottky Diodes 
for Millimeter- Wave Integrated Circuits," IEEE Trans. Microwave Theory Tech., Vol. 
MTT-38, No. 1, pp. 15-25, Jan. 1990. 

[11] N.J. Cronin, and VJ. Law, "Planar Millimeter-Wave Diode Mixer," IEEE Trans, on 
Microwave Theory Tech., Vol. MTT-33, No. 9, pp. 827-830, Sept. 1985. 

[12] J.A. Calviello, S. Nussbaum, and P.R. Bie, "High Performance GaAs Beam-Lead 
Mixer Diodes for Millimeter and Submillimeter Applications," Proc. of Intl. Electron 
Device Meeting, Dec. 7-9, 1981. 

[13] W.L. Bishop, K.A. McLeod, R.J. Mattauch, "Whiskerless Schottky Diode," U.S. 
Patent 5,041,881, Aug. 20, 1991. 

[14] T. Newman, W.L. Bishop, K.T. Ng, and S. Weinreb, "A Novel Planar Diode Mixer 
for Submillimeter- Wave Applications," IEEE Trans. Microwave Theory Tech., Vol. 
39, No. 12, pp. 1964-1971, Dec. 1991. 

[15] P.H. Seigel, R.J. Dengler, I. Mehdi, J.E. Oswald, W.L. Bishop, T.W. Crowe and R.J. 
Mattauch, "Measurements on a 215 GHz Subharmonically Pumped Waveguide Mixer 
Using Planar Back-to-Back Air Bridge Schottky Diodes," submitted for publication 
to IEEE Microwave and Guided Wave Letters, Oct. 1991. 

[16] T.W. Crowe and R.J. Mattauch, "Analysis and Optimization of Millimeter-and- 
Submillimeter- Wavelength Mixer Diodes," IEEE Trans. Microwave Theory Tech., 
Vol. MTT-35, Vol. 2, pp. 159-168, Feb. 1987. 

[17] W.C.B Peatman, P.A.D. Wood, D. Poterfield, T.W. Crowe and M.J. Rooks, "A 
Quarter-Micron GaAs Schottky Barrier Diode with High Video Responsivity at 118 
Microns," submitted to the Appl. Physics Lett., Feb. 1992. 

[18] E.M. Winkler, "A Study of the Effect of Reactive Ion Etching on the Noise 
Characteristics of Schottky Diodes," Master of Science Thesis, University of Virginia, 
Charlottesville, VA, August 1991. 



Third International Symposium on Space Terahertz Technology P a S e 615 



[19] W.L. Bishop, E.R. Meiburg, RJ. Mattauch, T.W. Crowe, and L. Poli, "A Micron- 
Thickness, Planar Schottky Diode Chip For Terahertz Applications with Theoretical 
Minimum Parasitic Capacitance," Proc. 1990 IEEE MTT-S Int'l. Symp., pp. 1 SOS- 
BOS, May, 1990. 

[20] P.H. Siegel, "A Planar Log-Periodic Mixtenna for Millimeter and Submillimeter 
Wavelengths," Proc. 1986 IEEE MTT-S Int'l. Symp., pp. 649-652, 1986. 



Page 616 Third International Symposium on Space Terahertz Technology 



5st-2>2> 

/ 1 Q&bh 



Planar Doped Barrier Subharmonic Mixers* 



T. H. Lee. J. R. East and G. I. Haddad 



/ Center for Space Terahertz Technology 

\ ^ The University of Michigan 

^ Ann Arbor, Michigan 



Abstract 



The fPDB (Planar Doped Barrierf* diode is a device consisting of a p + doping spike 
' between two' intrinsic layers and n + ohmic contacts. This device has the advantages of 
controllable barrier height, diode capacitance and forward to reverse current ratio. A 
symmetrically designed PDB has an anti-symmetric current vs. voltage characteristic and is ideal 
for use as millimeter wave subharmonic mixers. We have fabricated such devices with barrier 
heights of 0.3, 0.5 and 0.7 volts from GaAs and InGaAs using a multijunction honeycomb 
structure with junction diameters between one and ten microns. Initial RF measurements are 
encouraging. The 0.7 volt barrier height 4 micron GaAs devices were tested as subharmonic 
mixers at 202 Ghz with an IF frequency of 1 GHz and had 18 dB of conversion loss. The 
estimated mismatch loss was 7 dB and was due to higher diode capacitance. The LO frequency 
was 100.5 GHz and the pump power was 8 mW. 



* This work was supported by NASA under Grant No. NAG W- 13 34 



Third International Symposium on Space Terahertz Technology Page 617 

I. Introduction 

Planar Doped Barrier devices were first proposed in 1980 by Malik etal[l]. The structure 
can be understood as a planar doped p* spike sandwiched between two lightly doped regions and 
heavily doped n-type ohmic contacts. Such an n*-i-p+-i-n + device has a triangular potential barrier 
which is adjustable by the parameters of epi-layer growth such as sheet charge doping density 
and intrinsic layer dimensions. The devices are unipolar and the charge transport over the 
potential barrier can be modeled by the thermionic emission theory. The current vs. voltage 
characteristic is similar to that of Schottky diodes. Accordingly PDB's can be used as a Schottky 
barrier diode replacement with the additional advantage of barrier height control. 

The major applications of PDB devices are as mixers and detectors. If the device structure 
is designed symmetrically with the p + doping spike in the middle of the intrinsic region, the PDB 
diode has an anti-symmetrical I-V characteristics and is ideal for millimeter-wave subharmonic 
mixer applications. A summary of the theory and design techniques for symmetric PDB diodes 
as subharmonic mixers is given by Lee et al[2]. Several subharmonic PDB diodes operating at 
microwave frequencies have been reported[3,4,5]. Giittich et al. measured a D-band silicon PDB 
subharmonic mixer with a minimum conversion loss of 10.8 dB[6], The structure can also be 
designed to have lightly doped regions of different thickness. This will provide a current vs. 
voltage characteristic that is useful in low barrier detectors applications. Dale et a/[7,8]. reported 
a PDB single-ended mixer had a noise figure of approximately 6 dB at 9.4 GHz and required 
only 280 |nW of local oscillator power. When used as video detectors, Anand et al[9]. showed 
zero bias PDBs for low level detection had a burnout limit comparable to high barrier Schottky 
diodes and were less sensitive to the electrostatic discharge. 

PDB devices suffer from several problems. The space charge resistance is relatively high. 
By making the /-layer as short as possible we are able to reduce the space charge resistance. 
However, the /-layer width has to be larger than the Debye length of n + region to avoid the 
charge redistribution at i-n* interface. For a subharmonic PDB diode the ideality factor is at least 



Page si 8 Third International Symposium on Space Terahertz Technology 

two. The conversion ability is thus degraded, A careful design is necessary to obtain an optimum 
performance. 

In this paper we present a group of MBE grown GaAs wafers with a 250A i-layer 
thickness and the p + spike doping densities of 1.5, 2.0 and 2.5xl0 12 cm" 2 for mixer operation. The 
barrier height was designed for 0.3, 0.5 and 0.7 volts respectively. The impurity concentration 
of the intrinsic region is nominally less than 10 14 cm' 3 . Also presented are a group of InGaAs 
wafers grown via gas MBE at the University of Michigan. These wafers are of the same 
specifications as the GaAs ones except with longer n + layers. A preliminary RF result is reported. 

II. Fabrication of Whisker-contact Subharmonic PDB Diodes 

Two material systems have been chosen: GaAs and InGaAs. The use of GaAs has some 
advantages. GaAs is a more mature fabrication technology and tight parameter control of MBE 
grown GaAs is relatively easy. InGaAs, however, has a higher electron mobility and much lower 
contact resistivity than GaAs. For terahertz operation InGaAs might be a better material system. 
In this paper, we present diodes from both material systems. 

A typical wafer structure for the subharmonic GaAs PDB is shown in Fig. 1. An optimum 
structure for small diameter whisker-contact PDB diodes with low series resistance and parasitic 
capacitance is achieved by completely removing the substrate and forming mesas. An etch stop 
layer of AlGaAs between the substrate and epi-layer was included for this substrate removal. A 
good selectivity between the substrate and etch stop layer is possible. In the InGaAs system 
hydrochloric acid was used to remove the InP substrate without affecting the InGaAs epi-layer. 

The fabrication of whisker-contact PDB diodes consists of seven major steps as shown 
in Fig. 2: 

(1) diode definition and metallization, 

(2) mesa etch, 



1 



Third International Symposium on Space Terahertz Technology Page 619 

(3) Si0 2 passivation, 

(4) opening contact holes, 

(5) front side protection, 

(6) backside thinning, 

(7) final metallization and plating. 

More details on the process steps are given in the next section. 

The first step is diode definition and metallization. An image reversal photolithography 
process was characterized to attain desired diode patterns for ohmic metal lift-off. A positive 
photoresist was spun uniformly on the wafers, and then soft-baked at 105°C and UV exposed in 
the conventional way. Next a reversal bake and flood exposure altered the polarity of the 
solubility of photoresist in the images and fields, so that a negative image could be obtained after 
development. The exposure and bake parameters are optimized to obtain an undercut profile 
desirable for the lift-off. 

Metallization was then performed by depositing layers of thin film metals on the 
photoresist patterned samples. Ni/Ge/Au/Ti/Au were evaporated in sequence. The metal covered 
wafers were soaked in the acetone for lift-off. The resulting dot-like patterns also served as a 
self-aligned mask for mesa etch. 

The next step is the mesa etch. To avoid serious undercut we used a dry plasma etch 
instead of a wet chemical etch. Mesa etching was accomplished by a reactive-ion etching system. 
The GaAs wafers were etched in a low pressure chamber filled with BC1 3 and Ar gaseous 
plasmas of 11:9 ratio to obtain highly anisotropic sidewalls, while InGaAs wafers were etched 
by the mix of methane, hydrogen, and argon. The effective area of diodes was determined by the 
area of the p* doping spike. With plasma etching the precise control of device area is possible. 
The desired etch depth is into the first few hundred A of the bottom n + layer. Sometimes a 
subsequent slow wet etch is used to remove a thin damaged layer. 



eY 



Page 620 Third International Symposium on Space Terahertz Technology 

The third step is Si0 2 deposition. After the wafers were mesa etched, a dielectric film was 
deposited for step coverage. A plasma-enhanced chemical vapor deposition system was used. A 
low temperature process was adopted to avoid heating the devices. A silane/oxygen plasma at 
200°C for 90 minutes was used for optimum step coverage. This low temperature silicon dioxide 
layer was amorphous with low dielectric strength. A dielectric covered sample is shown in Fig. 
3. 

The fourth step is opening contact holes. At this stage, wafers were covered by PECVD 
Si0 2 everywhere. The mesa mask was used to open a hole where previously ohmic metals were 
deposited. The image reversal process as described in the first step with a careful alignment was 
required. The silicon dioxide on the top of ohmic metals was RIE-etched using gaseous plasmas 
of CHF 3 and CF 4 . It was difficult to tell, under the microscope, whether ohmic metal has been 
reached. By probing the adjacent diodes and measuring the electrical properties, we made certain 
the dielectric had been removed completely and could m